Solid state monolithic variable phase shifter with ... - Springer Link

International Journal o f Infrared and Millimeter Waves, VoL 7, No. 5, 1986

SOLID STATE MONOLITHIC VARIABLE PHASE SHIFTER WITH OPERATION INTO THE MILLIMETER WAVE WAVELENGTH REGIME Clifford M. Krowne and Robert E. Neidert Electronics T e c h n o l o g y Division N a v a l Research L a b o r a t o r y Washington, D . C . 20375-5000

Received February 5, 1986

ABSTRACT - Information is provided on the theory and design, fabrication, and experimental results for a phase shifter designed to operate in the millimeter wavelength region. The device was fabricated in a manner that makes it compatible with present GaAs monolithic microwave circuit technology. Continuously variable phase shift is obtained by varying the bias voltage from -5.0 to +0.65 V on a Schottky microstrip line. Experimental phase shift and loss data are provided for two different width (w) Schottky lines, w = 1.5 and 7.3 ~m, for frequencies 2-18 GHz.

I. INTRODUCTION Monolithic technology development in the microwave and millimeter wave frequency regimes has put a burden on the device and circuit designers to increase component compatibility with solid state processing methods, including reduction of real estate area utilized, decreased cost per device, and increased yield. The phase shifter is no exception to this trend. Presently, several elements like GaAs MESFET's, lumped capacitors and inductors, diodes and different electrical length (and impedance) transmission lines are used to realize phase shifters with 3 to 5 bit capability, each bit being, for example, 22.5 ~ or 45 ~ . Depending u p o n the frequency region of operation, the number of elements, or their

715 0195-9271/86/0500-0715505.00/0 9 1986 Plenum Publishing Coqaoratioa

716

sizes, may present unacceptable levels of excessive use of GaAs real estate area.

Krowne and Neidert

complexity or

It is possible and maybe highly desirable to put all of the phase shifting functions into a single elemental device, which satisfies the criteria of circuit simplicity and small size. As demonstrated in this paper such a device is realizable using relatively familiar GaAs fabrication techniques. The variable phase nature of the device is obtained by applying a bias voltage to a Schottky junction. Practical implications for a single distributed device performing the phase shifting functions are significant. Consider phase arrays employing electronic steering through phase shifter control. These arrays can have on the order of 104 radiators and their associated phase shifters and amplifiers. Reduction of phase shifter size or complexity can have a huge effect on array cost, reliability, size, and weight. II. THEORY AND DESIGN The device DI consists of a microstrip Schottky transmission line of width w = 1.5 ~m and length s = 1600 um on GaAs [I]. Cross-sectionally, the line sits on top of the depletion zone it forms in an epitaxial n layer (N D = i017/cc, b = 2 ~m); under this layer is an n+ epitaxial l~yer which acts as a ground plane (N D' = 2 x I018/cc, b' = 2 ~m). This multi-layered structure is a mesa residing on top of a semi-insulating GaAs substrate (0 = 7 x 107 ~ 9 cm), 500 ~m thick. The microstrip line transitions off of the mesa by a steep angle, dropping about 4 ~m, down to the Sl GaAs substrate and then it transitions to the contact pads (see Fig. I). Device design was based on choosing the lowest-order mode propagation constant through numerical calculations based on a parallel-plate model and a microstrip model [2]. These models are approximations since both assume planar layers, whereas an actual Schottky line has only a finite-sized depletion layer with its boundary curvilinear. Single mode operation is expected up to about 150 GHz, in part because the n layer epitaxial thickness has been chosen to exclude the infinite number of higher order surface wave modes. Impedance Zc of the lowest order mode may be estimated by using an analytical approximation to the parallel-plate model [3~. Schottky bias voltage V b change from +0.35 to -3.5 V at I0 GHz will change Z c from 28 to 49 ~. Figure 2 shows the metallization scheme

717

Monolithic Variable Phase Shifter

I_

Schottky Microstrip

4/~m

Line on Mesa 1.5 p.m

60 p~m

7I

I Contact Pad

60 ,u,m

t ORDINARY MICROSTRIP LINES

SLOW WAVE--.---MICROSTRIP LINE

Edge of 4/~m high mesa Figure i.

Edge of mesa/contact pad region.

OHMIC CONTA~

OHMIC CONTACT-~

__ I

I~ CONTACT i

I

~,

I

I

|

[Ti=

~ i,oooo / et - 1000 A

Ti = 300

I

"u = ~ooo ~ I

I

I

,u = ~oo ~,

|

Ni= I 0 0 ~

~4.2~m-~-IPt = 1000~14-=4.2 --"1 300~ I

N~ = 3oo

Ge = 4(X)~

GaAslayers : ~ 500/.Lm

Ti = 300 Au = 3000 ~ GROUNDPLANEMETALj Figure 2.

Metallization scheme.

employed in device fabrication. The ohmic metal Sequence Ni/Ge/Au/Ni, which is put down first, is alloyed to the GaAs surface. Penetration of the ohmic metal into the n-

718

Krowne and Neidert

GaAs epitaxial layer is on the order of 1/4 Bm. contact metal sequence Ti/Pt/An follows next.

Schottky

III. EXPERIMENTAL PROCEDURE Here the experimental measurement procedure used to determine the phase shifter device s-parameters is described for frequencies equal to or in excess of 2 GHz employing an automatic network analyzer (ANA). Figure 3 shows the device under t e s t surrounded by two bfas "T"'s, coaxial-to-microstrip transition fixtures, microstrip transmission lines on duroid, and bond wires. It is only necessary to apply bias voltage V b through one bias "T" so that the Schottky microstrip line of the phase shifter has bias. The two microstrip transmission lines on duroid, bond wires attaching these lines to the contact pads on the phase shifter chip (see Fig. 2), and phase shifter chip all are attached to a single carrier structure providing a continuous ground plane and mechanical support. For low VSWR networks on either side of the device chip and bond wires, where these networks are referred to as A and B in Fig. 3, the s-parameter matrix of the device (including bond wires and gap discontinuities) is given by

[4] SpSB =

ST

(1)

S21R

S T is the s-parameter matrix determined by using an ANA on the entire network shown in Fig. 3, and S21R the forward s-parameter of the reduced network which has the chip and bond wires removed. This technique assumes net-

!

," I I I t I

-

i

~

NETWORK

" I I I I

DC _

_

SENSING I

A

!

I

II I | I MICROSTRIP I

'.

i

i

I

MICROSTRIPI

BONO WIRE

BOND WIRE

t

NETWORK

B

DC n

~

SENSING

-- I I I I ! I

I I

Figure 3. Block diagram of the total network to be measured by an A N A including bias "T"'s, coaxial-to-microstrip connectors (CM), microstrip lines on duroid, bond wires, and the phase shifter chip.

Monolithic Variable Phase Shifter

719

works A and B are nearly identical, reciprocal, and symmetrical, which is the situation because they have been arranged to be composed of sub-networks matched to 50 ~ on their input and output ports. $21R is not found directly by an ANA measurement. Instead, the bond wires and chip are replaced by a 50 ~ microstrip line of length s and this is combined with the other two microstrip lines to form a single continuous line on duroid. This reference line is measured with an ANA yielding SR'. Assuming the line length s (electrical length in radlans is @t) is lossless, $21R = $21R' e -JOt

.

(2)

Placing (2) into (I) gives the s-parameters of the device combined with the two bond wires. Neglecting the electrical length of the bond wires (each about 200 um or less, with an electrical length 8b in degrees), results in an error 2Gb/@ of less than 2% in phase angle @ over the 2-18 GHz measurement range. Here @ is the electrical length of the chip itself. Thus the s-parameters of the chip Sps = Sps B given by (i). The maximum difference between the electrical length of the slow wave Schottky line @sw and @ is 4% (2@p/O) due to the spacing s (or Op in degrees) between the middle of a pad and the mesa edge. Both 85 and Gp have no effect on the differential phase shift obtained by varying bias voltage. From the above discussion, @ = - ZS21PS B

(3)

and G = Gsw" Power lost by passing through the slow wave Schottky line is calculated as L = lOs

[ ' S21pSR'2] i - ISIIPSBJ Z

(4)

where in deriving (4) it is assumed that no power is lost in networks A and B, bond wires, or pad-to-mesa distances. Measurements from 2-18 GHz were done using an HP 8743B reflection-transmisslon test set, lip 8350A sweep oscillator, and HP I1590A bias "T"'s. Measurements from 0.i to 2 GHz used an HP 8745A s-parameter test set and an HP 8350 sweep oscillator. A vector voltmeter was used for

720

Krowne and Neidert

frequencies from 40 MHz to i00 MHz. 2 GHz were reported elsewhere [I].

Measurements

below

IV. PERFORMANCE characteristics at room temperature (nominally about 20~ of the device are pad-to-pad resistance Rpp = 51.5 • 1.5 ~, Schottky diode capacitance C at bias V b = O, C(O) = 3.3 • 0.i pF, and shown in Fig. 4 is a DC

Ib(mA) 9 ;8

-7 -6 -5 -4

-2 -1 -5.0

-4.0

-3.0

-2.0

-1.0

1.0 0 .-1 .-2 ,-3 -4 -5 "-6 -7

Figure

4.

Schottky diode I/V curve.

Vb(V)

Monolithic Variable Phase Shifter

721

typical I/V curve. Rpp and C values are based on 46 devices with w = 1.6 • 0.I ~m. The solid circles in Fig. 4 are data points. RF measurements were done from 40 MHz to 18 GHz and indicate microstrip operation above 2 GHz. Below 2 GHz, the two large ohmic contacts used to electrically access the virtual n+ ground plane begin to appear electrically close [i] to the line, making the device appear to be a hybrid of microstrip and coplanar structures. Above 2 GHz, the phase shift O from pad-to-pad is linear when -3.5 < V b < 0.50 V. Figure 5 provides @ (degrees) versus frequency f (GHz) curves parameterized in terms of the Schottky bias voltage V b for device DI. In this figure, the solid circles are data points and the continuous curves approximations to the @ behavior. Notice that the V b = -5.0 V curve is very close to the V b = -3.5 V curve, suggesting that the greatest phase change is in the -3.5 < V b < 0.50 V range. Voltages greater than V b = 0.50 V could lead to damage of the phase shifter device, and so were generally avoided, although some measurements on DI

700

I

I

I

I

l

T

I

I

[

I

V b - 0.5 V |

I

600

500

==

4O0

==

300

V

200

100

0 0

I

I

I

L

I

1

I

I

2

4

6

8

10

12

14

16

[ | 1~

f (GHz)

Figure 5. Electrical length or phase sh~ft @ (degrees) versus frequency (GHz) for device chip DI. Schottky line width w = 1.5 ~m.

722

Krowne and Neidert

were performed up to 0.65 V of bias. The V h dependence of 8 may be understood by examining the approximate expression eeff = eo er ~'~-u bl

(5)

eef f is the effective dielectric constant of the slow wave phase shifter structure, e o the free-space permittivity, e r the relative dielectric constant of GaAs, bef f the effective ground plane depth below the Schottky line due to the n+ epitaxial layer, and b I the bias dependent depletion layer depth, bef f is a function of f, bef f = beff(f) , but for any given f, the beff/b I ratio will increase as V b increases since b I decreases. Depletion depth is given by bl =

i~ 2e~ D

(~B - Vb)

(6)

where ~B is the barrier height of the Sehottky metalsemiconductor interface and q the electron charge. The V b term must be replaced by V b + kTe/q, where T e is the electron temperature, if V b >> kTe/q is violated. T e = TL, the lattice temperature, if deviations from equilibrium are not too extreme. By (6), a V b increase leads to a reduction in b I value, thereby enlarging eel f and increasing the slowing of the propagating electromagnetic wave. Phase shift 8 as a consequence increases because c

'

(7)

w i t h m = 2~f and c the f r e e space v e l o c i t y of l i g h t . It is a p p a r e n t from ( 7 ) t h a t the d i f f e r e n t i a l phase s h i f t AO, due to two b i a s v o l t a g e s being a p p l i e d , decreases as f decreases. Insertion loss L(dB) versus f(GHz) for device DI is given in Fig. 6 for curves parameterized in terms of V b. Loss variation with f has an approximate /f behavior indicating ohmic loss through a metallic conduction mechanism. Consider the lowest order mode (with the greatest electromagnetic slowing) which has an attenuation constant dependent on f quadratically in the design range of parameters for an equivalent parallel-plate layered guiding structure with a depletion zone and a n-GaAs semi-

723

Monolithic Variable Phase Shifter

!!!V

25

V

20

v

m

15

10

2

I 4

I 6

A 8

] 10

L 12

I 14

I 16

I 4~8

f (GHz)

Figure 6. Phase shifter insertion frequency (GHz) for device chip DI.

loss

L (dB)

versus

conductor layer. Since ohmic loss in the parallel-plate guiding structure is about 0.25 dB/mm when V b = 0.0 V and f = I0 GHz, it is likely that most of the loss arises from the Schottky microstrip line and the n+ virtual ground plane. Using (5) in a microstrlp computer program for a loss estimate allows the Fig. 6 results to be reproduced, with the finding that most of the loss contribution comes from the Scottky microstrip line (about 90%). Furthermore, m o s t of the Schottky llne loss is due to the high fields at its line edges (edge condition), not due to an insufficiently small ~/t s ratio where ~ is the skin depth and t s the Schottky line thickness (here t s = 0.63 ~m). The physical basis for the downward shift of the curves is the increasing depletion thickness b I which allows a greater proportion of the electromagnetic wave to propagate in a very low loss region away from the Schottky conductor edges. Using an effective dielectric concept in (5), this means that the decreasing V b increases b I < b causing eef f to be reduced. Also as f + 18 GHz, bef f § b.

724

Krowne and Neidert

Tables I and 2 summarize the information in Figs. 5 through 8 at three bias voltages and three frequency points. Also provided in Table i is Ae = e(0.50) 0(-3.5), the differential phase shift obtained between the bias voltages V b = 0.50 V and -3.50 V. At 18 GHz and V b = 0.0 V, the G/%, AO/s O/L, and AG/L ratios for device DI are respectively 320~ 150~ 27~ and 13~ At 2 GHz and V b = 0.0 V for device DI, these ratios are respectively 56~ 30~ 15~ and 8~ For another device DPLI to be elaborated on below at 18 GHz and V b = 0.0 V, the ratios are respectively 366~ 138~ mm, 39~ and 15~ And at 2 GHz with V b = 0.0 V,

Table I.

Phase Shift @ (degrees) [VD(V), f(GHz)]

f

2

i0

18

104.8

362.7

632.1

89.7

310.1

515.9

56.7

235.2

392.3

48.1

127.5

239.8

102.5

380.5

666.1

o.o

95.7

340.6

586.1

-3.5

74.0

263.7

445.5

28.5

116.8

0.50

0.0

0

A

i

-3.5

0.50

A@

r-q

0-4

>,

~

220.6 J

AG

725

Monolithic Variable Phase Shifter

Table 2.

Insertion

Loss L (dB)

2

I0

18

0.50

9.87

18.20

24.40

0.0

5.93

14.00

18.94

-3.5

4.11

9.88

12.63

0.50

5.71

12.16

17.53

0.0

3.58

9.42

15.10

-3.5

2.04

7.05

10.66

r-4

~q

>

[D

the ratios are respectively 60~ 18~ 27~ and 8~ The phase shift per dB of loss and the differential phase shift per dB of loss as evidenced by the previous ratio values show that device DPLI has better performance than device DI. Because the predominant contribution to L appears to arise from the microstrip conductor, another device (DPLI), with similar dimensions to device DI, was electroplated with Au to roughly twice its thickness t s. w increased by a factor of 4 from the original 1.7 ~m value to approximately 7.3 um. Its @ and L behavior is given in Figs. 7 and 8. Although device DPLI has similar behavior to D1 as regards @, its L is significantly reduced by 24 dB over the 2-18 GHz band (also see Tables 1 and 2). If one desires Z e to he unchanged, i.e., w unchanged, then a new metallization process could be utilized to only thicken the microstrip. However, only increasing t s will have a much smaller effect on L than widening the Schottky line for these small widths w at room temperature.

726

Krowne a n d Neidert

1000

I

I

I

I

I

I

I

I

900

800

700

V - 0.!

-ii

60O

== soo r 4OO

300

200

loo

o 0

I

I

I

I

I

I

I

I

I

2

4

6

8

10

12

14

16

18

f (GHz)

Figure 7. Phase shift line wldth w = 7.3 ~m.

8 for

device DPLi with

Schottky

o o

m

eL

,

l

>f

I

>,f %

I

1

I

I>

(@P) I

0

I ao

J_.

!~

w

0

,@

N

CO

{If ,,.C:

.r,.l ,..C: t~

.,.J

.,,~

4-I

o

0

0 q-I

(I,)

I>

tj .,-I

728

Krowne and Neidert

V. CONCLUSION Slow wave variable phase shifting using a single elemental solid state device has been shown to be practicable in a monolithic format in t h e microwave frequency spectrum, and appears feasible at higher frequencies also. Device insertion loss can be compensated for in circuit applications by pre-or post-amplifier networks, or where this approach is unacceptable, by reduction of microstrip conductor loss using a lower loss metallization scheme of elements and thickness, or lower characteristic impedance Z c. ACKNOWLEDGMENT The authors thank Dr. Richard Gold of the AdamsRussell Semiconductor Center, Burlington, MA for his technical expertise and interest in this work. REFERENCES [I] R . E . Neidert and C . M . Krowne, "Voltage Variable Microwave Phase Shifter," Electronics Letters, Vol. 21, pp. 636-638, July 1985. [2] C. M. Ktowne, "Slow-Wave Propagation in Two Types of Cylindrical Waveguides Loaded with a Semiconductor," IEEE Trans. Microwave Th. Tech., Vol. MTT-33, pp. 335-339, Apr. 1985. [3] H. Hasegawa, M. Furukawa, and H. Yanai, "Properties of Microstrip Line on Si-SiO 2 System," IEEE Trans. Microwave Th. Tech., Vol. MTT-19, pp. 869-881, Nov. 1971. [4] R . E . Neidert, "High Accuracy Microwave S-Parameter Measurements on Solid State Devices," Naval Research Laboratory Report No. 4015, p. 13, June 22, 1979.