A compact radiation system for L band magnetically

A compact radiation system for L band magnetically insulated transmission line oscillator Dong Wang, Fen Qin, Mei-you Shi, Dai-Bing Chen, Jie Wen et al. Citation: AIP Advances 3, 052128 (2013); doi: 10.1063/1.4808020 View online: http://dx.doi.org/10.1063/1.4808020 View Table of Contents: http://aipadvances.aip.org/resource/1/AAIDBI/v3/i5 Published by the AIP Publishing LLC.

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AIP ADVANCES 3, 052128 (2013)

A compact radiation system for L band magnetically insulated transmission line oscillator Dong Wang,a Fen Qin, Mei-you Shi, Dai-Bing Chen, Jie Wen, Xiao Jin, and Sha Xu Laboratory of High Power Microwave Technology, Institute of Applied Electronics, China Academy of Engineering Physics, Mianyang, 621900, China (Received 19 November 2012; accepted 15 May 2013; published online 22 May 2013)

A compact radiation system for L band magnetically insulated transmission line oscillator (MILO) is present. It consists of a coaxial TEM-TE11 mode converter and a coaxial multimode horn antenna. Some metal plates are inserted into the coaxial waveguide to divide it into several sectoral waveguides with different phase propagation constants to compose a coaxial TE11 mode. Then some of the sectoral waveguides are folded up to reduce the length of the mode converter. Multimode horn antenna is adopted to enhance aperture efficiency and power handling capability of the radiation antenna. A radiation system for a 1.58 GHz MILO device is designed with a length of 33 cm and an aperture radius of 20 cm. The gain is 15.8 dBi and aperture efficiency is 86.8 %. Both low power and high power microwave experiment is carried out to test the radiation character. The experiment test results agree well with C 2013 Author(s). All article content, except where otherwise computer simulation.  noted, is licensed under a Creative Commons Attribution 3.0 Unported License. [http://dx.doi.org/10.1063/1.4808020]

I. INTRODUCTION

Many high power microwave (HPM) sources, such as magnetically insulated transmission line oscillator (MILO), relativistic backward wave oscillator (RBWO) and relativistic klystron amplifier (RKA), generate azimuthally symmetric output microwave modes (typically TEM mode or TM01 mode). Thses modes have symmetric field distribution on the antenna aperture if radiated directly, resulting in a divergent energy distribution in the far-field region, and this is a unfavarable effect for applications in most cases. So mode converters are usually adopted to tailor the spatial distribution of electromagnetic energy to meet application requirements. However, the introduction of a mode converter also results in an increase of dimensions and complexity of a HPM source. And progress in fielding HPM sources on mobile platforms requires developing more compact mode converters. TE11 waveguide mode is appropriate for radiation because its radiation pattern makes the illumination of microwave power more effective.1, 2 The majority of TM01 to TE11 mode converting methodes are generally realized with a serpentine type of circular waveguide.2–4 For example, a dual-bend mode converter is designed to effectively convert TM01 to TE11 mode for a high power transmission system.2 It has high power-handling capability, but the input and output ports are not aligned on the same axis, which will affect the compactness of the system. To realize a linear structure, a wide band TM01 to TE11 mode converter is designed by inserting a rectangular waveguide among circular waveguides.4 This structure is more complicated and discontinous, and power-handling capability is the limitation for high power transmission. Another type of linear structure mode converter is designed by inserting several metal plates into a coaxial waveguide.5 In this case, a coaxial waveguide is divided into several sector waveguides

a Electronic mail: [email protected]

2158-3226/2013/3(5)/052128/8

3, 052128-1

 C Author(s) 2013

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(a)

(b)

(c)

FIG. 1. Structure of the compact radiation system: (a) cross-section at the entrance of the mode converter; (b) vertical-section of the compact radiation system; (c) the upper folded sectoral waveguide and lower sectoral waveguides.

with different phase velocities for microwave. Because the phase velocity difference is relatively small in vacuum transmission line, it is difficult to fulfill highly efficient transformation in a short length. The authors suggest an improved structure in,6 where four folded waveguides are used as a substitute for the sectoral waveguides so that the required length is halved and the transformation is accomplished in the transverse direction. In this paper, we present a much compact HPM radiation system for an L band MILO. It consists of a coaxial TEM-TE11 mode converter and a coaxial multimode horn antenna. The former converts a TEM wave into a coaxial TE11 mode, and the later radiates it directly. The proposed mode converter integrates the advantages of both5 and.6 It is much shorter than that in,5 and has a smaller radius than that in.6 As we known, the conical horn antenna suffers from phase differences in the field distribution of the aperture, resulting in an aperture efficiency of about only 50%. Sometimes dielectric lenscorrected horn has been used to get better aperture efficiency. But when it is applied in L band application, it’s bulky and cumbersome. In order to get a compact and high efficiency radiation system, a coaxial multimode horn antenna is designed in this paper. The multimode horn antenna consists of an inner cone, a three ladder horn and a dielectric window. The structure parameters are optimized to achieve high aperture efficiency and power handling capability within a compact volume. II. DESIGN OF RADIATION SYSTEM A. Structure of the antenna

The structure of the compact radiation system is illustrated in Figure 1. As described in,5 a coaxial waveguide is firstly divided into four sectoral waveguides by four inserted metal plates. Then the upper two merge into one and this waveguide is folded up to increase the path length that microwave is transmitted through. The lower two sectoral waveguides are not changed in the mode converter region. Therefore, microwave propagates in two kinds of TE11 sectoral waveguides with different phase velocities and different path length. When the two kinds of TE11 sectoral waveguide modes satisfy the equation of β1 L 1 − β2 L 2 = π

(1)

The upper and lower half TE11 sectoral waveguide modes will meet at the end of the mode converter with a phase difference of π , and finally we could get a coaxial TE11 waveguide mode. β 1 and β 2 are the propagation constants of the upper and lower sectoral waveguide modes, respectively, and L1 and L2 are the path length of the upper and lower sectoral waveguides. As shown in

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FIG. 2. Calculated s parameter of the reflected and transmitted modes.

Figure 1, path L1 is longer than L2 ; and β 1 is larger than β 2 , so the length of coaxial mode converter could be much shorter than that in5 where L1 equals L2 . This mode converter is also more compact than that in6 because β 1 equals β 2 in.6 Additionally, the separation angle of the four metal plates could be adjusted so the value of β 1 and β 2 could be changed to further reduce the length of the mode converter. A metal plate is added on the upper half at the end of mode converter. It serves mainly as a matching plate. The inserted metal plates produce some reflections when they transferring an input TEM waveguide mode into four TE11 sectoral waveguide modes. In this paper, the reflection is compensated by adjusting the length of the matching plate, the separation angle of the metal plates, and the bending angle of the folded waveguide. Therefore, conventional matching transformer is not needed and the length of mode converter is reduced moreover. Multimode horn antenna consists of an inner cone, a three ladder horn and a dielectric window. The cone serves as a transition for the inner conductor. Besides, it changes the field distribution of the radiation horn and results in better radiation directivity.5 The ladder structure is adopted to adjust the amplitude and phase distribution of the field of the aperture. By carefully selecting structure parameters, the antenna could eliminate reflection by itself. B. The coaxial mode converter

According to equation (1), the mode converter can be easily designed. The inner and outer radii of the coaxial waveguide are 6.8 cm and 10.6 cm, respectively. We first choose a separation angle of the metal plates; for example, 90◦ in the lower half, the propagation constants β 1 and β 2 could be calculated. The width of the bending waveguide could be selected according to the need of the power handling capability. Setting a value for L2 , then the path length L1 is obtained. The commercial software cst microwave studio is adopted to optimize the parameter structures after the model is built up. In this paper, optimum length of the mode converter is 8.0 cm at 1.58 GHz. Optimum separation angle between the metal plates in the lower half is 85◦ , and the maximum radius

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FIG. 3. Electric fields in the multimode horn.

of the mode converter is 16.5 cm. The calculated s parameter of the reflected and transmitted modes is shown in Figure 2. There are four modes (TEM, TE11 , TE21 , TE31 ) reflected to the input port and four modes (TEM, TE11 , TE21 ,TE31 ) transmitted to the output port. Around the center frequency, the input microwave is mainly converted into output TE11 mode; other reflected and transmitted modes are very small. C. The coaxial multimode horn antenna

The coaxial multimode horn antenna is designed by inserting a cone inside a three ladder multimode horn. The thickness of the window is designed to be about half of wavelength. The directivity of the horn is affected by all the structure parameters, such as the radius of the aperture,

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FIG. 4. Electric fields on the surface of the dielectric window.

the radii and lengths of each ladder, the height of the cone, the end radius of the cone. It is optimized with a total length of 16 cm by cst. The optimum height of the cone is 7 cm, end radius of the cone is 1.5 cm, and the aperture radius is 20 cm. With these parameters, we get a directivity of 15.8 dBi, corresponding with an aperture efficiency of 86.8%. Figure 3 shows the electric field distribution in the multimode horn. It is clear that field phase difference in the aperture is much small. And microwave propagates through the antenna like a plane-wave. This is the reason for high aperture efficiency. Moreover, for a conventional conical horn with the same directivity and aperture radius, its length has to be 52.5 cm, which is three times longer than the length of the coaxial multimode antenna. Figure 4 shows the electric fields on the surface of the dielectric window. It is shown that the field distribution is relatively uniform compared with a TE11 mode field distribution and the maximum field strength is 73 V/m when a signal with 1W power is input. This means that the antenna is capable of radiating GW level microwave signal.

III. RESULTS AND DISCUSSION

By assembling the coaxial mode converter and the coaxial multimode horn antenna together, we will get the whole structure of the radiation system. The length of the connection section between

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FIG. 5. Reflection coefficients of the antenna.

FIG. 6. Radiation patterns of the antenna at 1.58 GHz.

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FIG. 7. Schematic of the far-field measurement system.

FIG. 8. Radiation microwave signal: 50 ns/ div.

them is optimized to minimize the reflection of the whole structure. Finally, the total length of the radiation system is 33 cm, and the maximum radius is 20 cm. Fig. 5 shows the simulated results of the antenna reflection coefficient. Near the center frequency, the reflection is lower than −20 dB for each mode. The bandwidth of the radiation system is about 50 MHz (lower than −15 dB); it is suitable for application with a narrow band microwave source. A. Low power microwave test

To test its radiation characteristics, the device of a TEM mode excitation for feeding a TEM wave from a coaxial cable is connected to the input of the proposed radiation system. The radiation patterns of the antenna at 1.58 GHz are shown in Fig. 6. The antenna gain is about 15.4 dBi while

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FIG. 9. Power density distribution at H_plane.

the simulation result is 15.8 dBi. The 3 dB beamwidths are 27.1◦ in E-plane and 31.1◦ in H-plane. The sidelobes of the radiation patterns are both less than −15 dB. B. High power microwave experiment

The radiation system is connected to a MILO device and the radiation character is tested in high power microwave experiment. The device is driven by a set of power adjustment equipment in our laboratory. The entire apparatus is pumped down to approximately a pressure of 2 × 10−2 Pa. The measurement system is shown in Fig. 7. To avoid breakdown, some S-band (2.6–3.95 GHz) open waveguides (BJ-32) are used to be microwave signal receivers. Typical results of the detected microwave signal are shown in Figure 8. Microwave frequency is 1.58 GHz, FWHM is over 40 ns. The far-field power density distribution as well as the simulated radiation pattern is shown in Figure 9. The measured pattern agrees well with the simulated result, indicates that the radiation system works well with HPM devices. IV. CONCLUSIONS

An L band compact radiation system consists of a coaxial TEM-TE11 mode converter and a coaxial multimode horn antenna is investigated. It has low reflection coefficient and the bandwidth is about 50 MHz. The total length of the radiation system is only 33 cm, and aperture radius is 20 cm. It produces a gain of 15.8 dBi with an aperture efficiency of 86.8%. Hence, it is suitable for compact narrow band high power microwave apparatus. 1 S.

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