A Highly Area-Efficient Controller for Capacitive Touch ... - IEEE Xplore

T.-H. Hwang et al.: A Highly Area-Efficient Controller for Capacitive Touch Screen Panel Systems

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A Highly Area-Efficient Controller for Capacitive Touch Screen Panel Systems Tong-Hun Hwang, Student Member, IEEE, Wen-Hai Cui, Ik-Seok Yang, Student Member, IEEE, and Oh-Kyong Kwon, Member, IEEE

Abstract - In this paper, a highly area-efficient controller

for capacitive touch screen panels (TSPs) is proposed. The proposed controller uses a 10-bit successive approximation register analog-to-digital converter (SAR ADC) with an adder to compensate for the capacitance variation in the TSP and for the offset voltage variation in the charge amplifier of the sensing circuit. By using the proposed compensation method, the area of the controller can be reduced by 90.3% of the area of the conventional controllers. The measurement results showed that the signal-to-noise ratio (SNR) of the controller increases from 12.5 to 21.3 dB after compensation. Also, its spatial jitter decreases from ±1.5 to ±0.46 mm, which is 7% of the sensor pitch of 8 mm. 1 Index Terms: Projected capacitive touch screen panel, compensation technique, mutual capacitance, successive approximation register analog-to-digital converter

I. INTRODUCTION The demand for touch screen panel (TSP) applications is continuously increasing in home, industry, and mobile applications. Consumers require user-friendly electronic systems with a comfortable and instinctive input method, such as the smart multifunctional displays. Many TSPs have been studied and commercialized to satisfy these demands. Currently, three types of TSPs can be identified based on the location of the touch sensors: the add-on type, the on-cell type, and the in-cell type. The touch sensors of the add-on-type TSP, the most commercial type, found in small to large panels, such as mobile phones, laptop PCs, and TVs, are separated from the display panel. The manufacture of such TSP type is very mature, and many sensing methods for it have been reported [1]-[6]. In the case of the on-cell type, the touch sensors are located between the polarizer and the indium tin oxide (ITO) electrode in the display panel. It is applied to the capacitive sensing method for small-panel applications, 1 This work was sponsored by ETRI SoC Industry Promotion Center,under its Human Resource Development Project for IT-SoC Architects. Tong-Hun Hwang, Wen-Hai Cui, Ik-Seok Yang, and Oh-Kyong Kwon are with the Department of Electronics Engineering of Hanyang University in Seoul, Korea (e-mail: [email protected]). Contributed Paper Manuscript received April 15, 2010 Current version published 06 29 2010; Electronic version published 07 06 2010.

such as mobile phones and mobile devices [7]. The touch sensors of the in-cell-type TSPs are located in the pixels. The in-cell-type TSPs are currently being intensively studied so that they can be applied to the optical, resistive, and capacitive sensing methods. This TSP type has not yet been commercialized, however, due to a number of processing issues [8]-[10]. Many sensing methods for add-on-type TSP applications, such as the resistive [1], capacitive [2]-[4], acoustic-wave [5], and infrared methods [6], have been commercialized. Among these, the capacitive method is increasingly being used due to its sensitivity, durability, and ability to recognize multi-touch. Most sensing methods, except for the projected capacitive method, cannot be applied to multi-touch events and gestures because of the shielding and averaging effects [2]-[4]. In the case of the projected capacitive TSP, a compensation circuit is needed to reduce the sensing errors caused by process variations, such as the capacitance variation in the TSP and the offset variation in the charge amplifier. The previously reported compensation methods utilize the gain-controllable charge amplifier, which uses a programmable capacitor array (PCA), to compensate for the sensing capacitance, and the reference-voltage-controllable ADC, which uses offset DAC to compensate for the offset of the charge amplifiers [4]. The gain-controllable charge amplifiers require a large area, however, because their numbers are equal to the number of column lines of the TSP. This characteristic forces a trade-off between the accuracy and the chip area. In conclusion, the previously reported compensation method has low efficiency in terms of area for large TSPs and high-accuracy compensation systems. In this paper, a compensation method is proposed to reduce the sensing errors caused by process variations, such as the capacitance variation and the offset variation in the charge amplifier of the projected capacitive TSP. The proposed compensation method uses 10-bit SAR ADC with an adder, which includes a C-2C DAC. The area of the proposed controller is reduced by eliminating the PCAs and the offset DAC. Furthermore, the proposed compensation method is free to trade-off between accuracy and area.

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IEEE Transactions on Consumer Electronics, Vol. 56, No. 2, May 2010

(a) (a)

(b)

(b) Fig. 1. (a) Structure of the projected capacitive TSP. (b) Top view of the diamond-patterned TSP.

II. OVERVIEW OF THE TSP SYSTEM A. Structure of the Projected Capacitive TSP Fig. 1 shows the structure of the projected capacitive TSP and its top view, which has diamond-type patterns. Three transparent layers are shown: the top plate, the dielectric layer, and the bottom plate. The sensing lines are patterned on the top plate, and the driving lines are patterned on the bottom plate. The sensing and driving lines are patterned perpendicularly and are separated by the dielectric layer. The sensing lines are located on the top plate so it is closer to the touch object and farther from the display panel. As a result, the sensitivity to a touch object is increased, and the immunity to the noise from the signals of the display panel is improved. At each cross-point of the sensing and driving lines, mutual capacitance is generated due to their electric field, as shown in Fig. 2. Fig. 2(a) shows the model of the cross-sectional view of the projected capacitive TSP along the cross-sectional line x1x1’ as shown in Fig. 1(b), and Fig. 2(b) shows the contour of the electric potential extracted from the Raphael [11] simulation result. Fig. 2(c) and Fig. 2(d) show the model and the contour of the electric potential for the crosssectional line x2-x2’ as shown in Fig. 1(b).

(c)

(d) Fig. 2. The cross-sectional model of the projected capacitive TSP (a) at region A in Fig. 1. (b) Contour of the electric potential for the crosssection of the TSP. (c) Cross-sectional model of the TSP at region B in Fig. 1. (d) Contour of the electric potential for the cross-section of the TSP.

When a finger or other conductive object touches the panel, the electric fields are shunt to the ground through the objects. The electric field appearing at a touched point is different


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from those appearing at the untouched points. The mutual capacitance of the unit length at region A as shown in Fig. 1(b), CA, which is formed between a sensing line and a driving line, is calculated as wε 2πε (1) + CA = s log( 4 s / h ) , where w is the width of the ITO electrode, ε the permittivity of the dielectric material, s the thickness of the dielectric layer, and h the height of the sensing line. In (1), the first term, wε/s, is the capacitance formed at the cross-point, and the second term, 2wε/log(4s/h), is the capacitance formed in the fringing field [12]. The mutual capacitance of the unit length at region B as shown in Fig. 1(b), CB, can be calculated as the capacitance in the fringing area as CB =

2πε log( 4 s / h ) .

(2)

Therefore, the whole sensing capacitance is calculated as C sense = k

αβε s

+k

8πλε , log( 4 s / h )

Fig. 3. Block diagram of the overall system of the TSP.

(3)

where α and β are the length of the overlapped region between the driving and sensing lines, respectively, and λ is the length of the diamond-patterned ITO shown in Fig. 1(b). When the TSP is untouched, the coefficient of the sensing capacitance, k, is 1, and when it is touched, k becomes d/(d+s) where d is the thickness of the top plate. Therefore, k is reduced because the charge between the conductors is distributed. The equation is verified by comparing it to the Raphael simulation data and the measured data with the TSP, whose α, β, h, s, d, and λ are 1.0 mm, 1.0 mm, 1.0 µm, 254.0 µm, 711.2 µm, and 4.6 mm, respectively. Csense became 1.47, 1.5, and 1.55 pF when the equation, Raphael simulation data, and measured data were used, respectively. When the TSP is touched, the fringing capacitance is reduced because the sensing object attracts the electric flux from the driving lines to the fringing area of the sensing lines. Csence , which was touched, became 1.08, 1.15, and 1.19 when the equation, Raphael simulation data, and measured data were used, respectively. By measuring the difference in their Csense , the touch position became evident. B. Controller for the Projected Capacitive TSP Fig. 3 shows the overall system of the TSP sensing system [2], [3]. In Fig. 3, Column 1 to Column M show the sensing lines, and Row 1 to Row N show the driving lines. Csense,11 to Csense,NM represent the sensing capacitance between the sensing and driving lines. CFB,1 to CFB,M represent the feedback capacitors of the charge

amplifier, and the switches from SRST,1 to SRST,M represent the reset switches. An excitation driver is connected to the driving lines of the TSP that can drive the driving lines, which are a large capacitance load. The charge amplifiers are connected to the sensing lines of the TSP. The charge amplifiers perform tasks such as converting the charge to voltage, amplifying the charge at the input node of the operation amplifier, and rejecting the stray capacitance present at the sensing lines. When amplifying the charge to voltage, the ratio is determined by the C sense /CFB ratio shown in Fig. 3. The multiplexer (MUX) consists of switches, which connect the output of the charge amplifier to the input of the ADC sequentially. ADC converts the output voltage of the charge amplifiers to a digital value. After receiving the digital data from ADC, the host process memorizes such data, performs image filters, and determines the touch events and touch points. The operating principle of this system is as follows. First, the reset switches from S RST,1 to S RST,M are closed at the same time, and the voltage of the sensing lines are held as the reference voltage, VREF, because the charge amplifiers become unity-gain buffers. After the reset switches open again, the first driving line is excited, and the charge is induced at all the sensing lines by the boosting effect of the sensing capacitances, Csense,NM. The induced charges in the sensing lines are converted to voltages by the charge amplifiers, and these voltages are simultaneously sampled and sequentially sent to the ADC by the MUX. The different charges boosted by the different sensing capacitances result in different output voltages of the charge amplifiers, thus resulting in different outputs of the digital values of the ADC. When the data conversion of the first driving line is completed, the next driving line is excited. This process is repeated until the last driving line is excited, and its data

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conversion is completed. The N×M-pattern image is then formed. By subjecting the N×M-pattern image to image processing, the touch position can be detected. In practice, process variations such as the capacitance variation in the TSP and the offset variation in the charge amplifier cause sensing errors. Due to these errors, the wrong touch position is determined, and wrong touch events are recognized. By performing compensation, the sensing errors can be reduced, and the touch event and position can be more accurately determined. III. COMPENSATION CIRCUIT

Fig. 4. Schematic diagram of the gain-controllable charge amplifier for the previously reported compensation method.

(a)

(b) Fig. 5. (a) Schematic diagram of the adder-embedded SAR ADC for the proposed compensation method, and (b) its modified switch used in the C-2C DAC of the SAR ADC.

A. The Previously Reported Compensation Method Fig. 4 shows the schematic diagram of the circuits that were used in a previously reported compensation method [4]. The previously reported circuit uses gain-controllable charge amplifiers to compensate for the capacitance variation in the TSP. Using a PCA, the circuit can change CFB and can control the gain of the charge amplifier, Csense/CFB. The reference-voltage-controllable ADC compensates for the offset variation in the charge amplifier. By using the offset DAC, the reference voltage of the ADC can be made programmable. The control data are memorized in the registers and are read when needed. The previously reported compensation method requires a large area because the charge amplifier with a large PCA needs as many column lines as in the TSP. Furthermore, to improve the accuracy of the compensation, the areas of the PCAs and offset DAC are doubled when the bits are increased by one bit. The previously reported compensation method is not very efficient in terms of area when the TSP is larger, and the compensation system needs greater accuracy. Therefore, a high degree of accuracy and area efficiency are required to improve the compensation method. B. The Proposed Compensation Method The use of an adder-embedded 10-bit SAR ADC is proposed to compensate for the capacitance variation in the TSP and for the offset variation in the charge amplifier. Fig. 5(a) shows the schematic diagram of the circuits that are used in the proposed compensation method. The 10-bit SAR ADC consists of a comparator a SAR logic, and a C-2C DAC with a 10-bit adder. Node A is the negative input node of the comparator, as shown in Fig. 5(a). The C-2C DAC performs add operation due to the modified switches, as shown in Fig. 5(b). In the switch of the n-th bit, S n is controlled by the data of COMPn when the sampling phase and S n are controlled by the data of b n or Reg_b n during the conversion phase, as shown in Fig. 6. The register for the embedded adder memorizes the initial data of the TSP converted by the SAR ADC when all the points in the TSP are untouched. The initial data of


the TSP includes the information regarding the capacitance variation in the TSP and the offset variation in the charge amplifier. Using the initial data, compensation data are generated and added to the measured data when recursive operations are performed for a touch-sensing event. The operation principle of the proposed compensation method is as follows. In the initialization period, which is performed in an untouched state during the display turnon time, the initial data of the capacitance of the TSP is extracted. At that time, the adder-embedded SAR ADC is operated as a normal SAR ADC. In the sampling phase, all the switches, from MSB, b 0, to LSB, b 9, are initially connected to the ground, GND. In the conversion phase, the switches are sequentially connected to the reference voltage, VREF, from b 0 to b 9. Then the sampling node is boosted by the capacitor, and the output of the comparator is determined by the sampling node and the reference voltage of the comparator. If a sampling node is higher than VREF, digital data “0” is stored at Reg_bn, and switch b n is connected to GND. If a sampling node is lower than VREF, digital data “1” is stored at b n, and switch b n is connected to V REF. At the end of this operation, the voltage of node A, VIN, is the same as VREF, and the n-bit digital output is satisfied, as follows: 1 1 1 ⎛1 ⎞ VREF − VIN = VREF ⎜ 0 b0 + 1 b1 + L+ N −1 bN −1 ⎟ 3 2 2 ⎝2 ⎠.

=

]

1 1 1 ⎛ 1 ⎞ V REF ⎜ 0 b '0 + 1 b '1 + L + N −1 b ' N −1 ⎟ , 3 2 2 2 ⎝ ⎠

(a)

(b)

(4)

In (4), the right-side term presents the voltage boosted by the C-2C DAC, and b n presents the converted digital data. The SAR ADC then sends the 10-bit digital data to the host processor, which makes the 10-bit compensation data, COMP n, write to memory until the compensation data for all the points in the TSP are memorized. In the touch-sensing phase, the recursive touch sensing and real-time compensation are operated by the adderembedded SAR ADC. In the sampling phase, some switches are connected to GND, while the other switches are connected to VREF by a COMP n combination, which is transferred from the host processor. In the conversion phase, the operation of the adder-embedded SAR ADC is the same as that of the normal SAR ADC. At that time, (5) is obtained from (4) by substituting “VIN-VCOMP” for VIN. V REF − [V IN − VCOMP

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(5)

(c) Fig. 6. The waveform of node A, which is the negative input node of the comparator in the SAR ADC (a) during the initialization period and (b) during the touch-sensing phase with compensation. (c) Timing diagram of the proposed compensation method.

where VCOMP is the boosted voltage converted by the digital data, COMPn, for compensation in the sampling phase, and b'n is the compensated digital data. Equation (6) is obtained from (5). VREF − VIN 1 1 1 ⎛ 1 ⎞ = VREF ⎜ 0 b' 0 + 1 b'1 + L + N −1 b' N −1 ⎟ 3 2 2 ⎝2 ⎠

(6) .

1 1 1 ⎛ 1 ⎞ − VREF ⎜ 0 COMP0 + 1 COMP1 + L + N −1 COMPN −1 ⎟ 3 2 2 ⎝2 ⎠

Therefore, the compensated digital data is extracted as (7) from (4) and (6).

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Fig. 8. The TSP system, which consists of the touch sensor, the controller, and the FPGA board. TABLE I SUMMARY OF SPECIFICATIONS OF THE PROPOSED TSP SYSTEM Fig. 7. Schematic diagram of the adder-embedded SAR ADC for the proposed compensation method.

b ' n = b n + COMP n .

(7)

For example, Fig. 6(a) shows the input node of the comparator in the SAR ADC operating normally for the initial data of the sensing capacitance in the TSP, and its digital data is “0100010011.” Fig. 6(b) shows the input node of the comparator in the SAR ADC when the compensated data is “0110010010,” by adding the compensation data “0001111111.” Fig. 6(c) shows a timing diagram of the SAR ADC, where Reg_b n is the register for the converted data, b n. The 10-bit compensated digital data is sent to the memory in the host processor, and this operation is implemented recursively until the data for all the points in the TSP have been converted and compensated. Next, the host processor performs image signal processing, including image filters and interpolation, and determines the x-y position by using the data in the memory, even if a multi-object is touched. Fig. 7 shows a flowchart of the TSP system with the proposed compensation method. By using the proposed 10-bit SAR ADC, the area efficiency can be increased because the gain-controllable charge amplifiers, including the PCAs, whose number is the same as that of the column lines, are replaced by the charge amplifiers with single capacitors. In [4], for instance, the gain-controllable charge amplifiers use 5-bit PCAs from 2 to 32 pF. In the proposed controller, the charge amplifier uses 2.5 pF. Furthermore, the offset DAC for the control of the reference voltage of the ADC is not required, and the C2C DAC in the SAR ADC makes it more efficient in

TSP

Controller

ADC

Processor

Properties

Specifications

Sensor type

Projected capacitive

ITO pattern type

Diamond

Pattern pitch

8 mm

Spatial resolution

9×7

Size

3.5-inch

Measuring type

Mutual capacitance

Type

SAR ADC

Resolution

10-bit

Sampling rate

100k sample/s

FPGA

Altera Cyclone II

terms of area. Finally, the area of the proposed controller is reduced by 90.3% compared to the previously reported controller. If the column lines increase, the area efficiency will further increase. IV. EXPERIMENT Fig. 8 shows the measurement system for the projected capacitive TSP. The TSP system consists of a readout board with the proposed circuit, an FPGA board, and the projected capacitive TSP. The structure of the TSP includes 9×7 ITO diamond-type electrodes, and the pitch of the ITO lines is 8 mm. The specifications of the TSP system are shown in TABLE I. The information regarding the sensing capacitance of the TSP was stored in the memory of the FPGA board. As the measurement results of this system, the sensing capacitances of the TSP were 1.55 and 1.19 pF when it was touched and


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compensated for by the adder, which was embedded in the SAR ADC. Fig. 9(a) and Fig. 9(b) show the 3D images of the sensing voltage before and after the compensation, respectively. The touch point is represented at (163, 193). The touch point was detected by extracting the minimum capacitance data and the Lagrange interpolation [13] with its adjacent data. After compensation, the affect of the process variations and electrical noise of the system can be decreased. Also, the accurate touch point is determined because interpolation is conducted using the compensated data. The measured performance of this system is summarized in TABLE II. When determining the touch point, the host process performs image processing with the ADC output data. These ADC output data are determined by the output voltages of the charge amplifiers, which convert the difference of the charge on the sensing capacitance in the TSP when it is touched and untouched. The output voltages of the charge amplifiers, however, have the capacitance information with the process variations and noise. In the proposed compensation method, the affect of the process variations and electrical noises can be reduced, and the signal-to-noise ratio (SNR) increases. After compensation, the sensing voltage error due to the process variations and electrical noise was reduced from 57.7 to 21.0 mV because the DAC in the SAR ADC adjusts the input voltage of the ADC using compensation data. The difference in the sensing voltage was 121.6 mV when the TSP was touched and untouched. After compensation, the SNR increased from 12.5 to 21.3 dB. The touch position of the 3.5-inch TSP was determined by the sensing capacitance data of the TSP and its spatial jitter was reduced from ±1.5 to ±0.46 mm. The proposed compensation algorithm can reduce the spatial jitter less than 7% of the pitch of the ITO row and column lines.

(a)

(b) Fig. 9. The 3D image made based on the measurement results of the capacitance when the TSP was touched at (163, 193) as the x-y coordinate for the application of the qVGA (320×240) resolution (a) before compensation and (b) after compensation. TABLE II SUMMARY OF PERFORMANCE OF THE PROPOSED CONTROLLER

Property

Specification

SNR (dB)

21.3

Spatial jitter (mm)

±0.46

Power (mW)

4.3

untouched, respectively, while the stray capacitance of the sensing line was 15.6 pF. Fig. 9 shows three-dimensional (3D) images using the measured sensing voltage of the TSP, which consists of ninerow lines from R1 to R9 and seven-column lines from C1 to C7. The x-y coordinate is the pixel position in the qVGA (320×240) displays. The z-axis is the output voltage of the charge amplifier when it is sampling the sensing capacitance of the TSP. The compensation is performed when the capacitance information was converted to digital data by the 10-bit C-2C SAR ADC in the readout IC. The process variations and noise were

V. CONCLUSION A highly area-efficient controller for a capacitive TSP system is proposed to compensate for the process variations. The proposed controller uses a 10-bit SAR ADC, including a C-2C DAC with an adder circuit, and can compensate for the process variations of the capacitance of the TSP and of the offset voltage of the charge amplifier in a sensing circuit. As the implementation results, the area was reduced by 90.3% compared with the previously reported controller. By increasing the efficiency in terms of area, the cost of making one chip controller for TSPs can become lower than when the previously reported controllers are used. Therefore, the proposed controller is suitable for low-cost and high-performance TSP systems. REFERENCES [1] [2] [3] [4]

R.N. Aguilar and G.C.M. Meijer, “Fast interface electronics for a resistive touch screen,” in Proc. IEEE Sensors, vol. 2, pp.1360–1363, 2002. S. P. Hotelling, J. A. Strickon, and B. Q. Huppi, “Multipoint touch screen,” U.S Patent 10/840,862. May. 11, 2006. S. P. Hotelling and B. R. Land, “Double-sided touch-sensitive panel with shield and drive combined layer,” U.S. Patent 11/650,182. Jul. 3, 2008. S. P. Hotelling, C. H. Krah, and B. Q. Huppi, “Multipoint touch surface controller,” U.S. Patent 11/381,313. Nov. 8, 2007.

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BIOGRAPHIES Tong-Hun Hwang (S’09) received his B.S. degree in Electronics Engineering from Hanyang University in Seoul, Korea in 2008. He is currently pursuing an M.S. degree at the same university. He has been engaged in research for liquid crystal displays. His research interests include the touch screen panel sensing system and transparent displays.

Wen-Hai Cui received his B.S. degree in Electronics Engineering from Yanbian University in Jilin Division, China in 2004. He is currently pursuing an M.S. degree at Hanyang University in Seoul, Korea. He has been engaged in research on the sensing mechanism of touch screen panels. His research interests also include the panel driving circuit of liquid crystal displays.

Ik-Seok Yang (S’07) received his B.S. and M.S. degrees in Electronics Engineering from Hayang University in Seoul, Korea in 1996 and 1998, respectively. From 1998 to 2006, he was with Magnachip Semiconductor Inc., in Seoul, Korea, where he was involved in the development of process integration and driving circuit design for flat panel display applications. He is currently pursuing a Ph. D. degree at Hanyang University in Seoul, Korea. He has been engaged in research on sensing methodologies of touch screen panel systems. His research interests also include the driving method and circuit of flat panel display applications. Oh-Kyong Kwon (S’83-M’88) received his B.S. degree in Electronics Engineering from Hanyang University in Seoul, Korea in 1978, and his M.S. and Ph.D. degrees in Electrical Engineering from Stanford University in 1986 and 1988, respectively. From 1980 to 1983, he was with LG Electronics, Inc. in Seoul, Korea, where he was involved in the development of telecommunications products, including the G-3 fax system and the PCM system. From 1987 to 1992, he was with the Semiconductor Process and Design Center of Texas Instruments, Inc. in Dallas, Texas, where he was engaged in the development of multi-chip module (MCM) technologies and smart power integrated circuit technologies for automotive and flat panel display applications. In 1992, he joined Hanyang University in Seoul, Korea as an assistant professor in the Department of Electronic Engineering, and he is now a professor in the Division of Electrical and Computer Engineering of Hanyang University. Dr. Kwon has been the Dean of the College of Engineering of Hanyang University since 2008. He was an IEEE IEDM subcommittee member on solid state devices from 1997 to 1998, the technical program chairman of the 1999 IEEE International Conference on VLSI and CAD, and a workshop co-chairman in the 2000 and 2001 Asia-Pacific Workshop on Fundamentals and Applications of Advanced Semiconductor Devices (AWAD). Dr. Kwon is now the chairman of IEEE EDS’s Korea Chapter. He was the program manager of the Korean TFT-LCD Research and Development Program from 1993 to 1997 and of the Korean Flat Panel Research and Development Program from 1998 to 2001. He was the technical program chairman of the International SoC (System-on-a-Chip) Conference 2004 and of the International Meeting on Information Displays/International Display Manufacturing Conference 2006. He is currently the Vice-President of the Institute of Electronics Engineers of Korea and the President of the Korea Information Display Society, was the executive chairman of the International Meeting on Information Displays 2007, and the technical program committee member of the Society for Information Displays, the International Solid-State-Circuit Conference, and the International Display Workshop. His research interests include interconnect and electrical noise modeling for high-speed system-level integration, wafer-scale chip-sized packages, smart power integrated circuit technologies, mixed-mode signal circuit design, and the driving methods and circuits for flat panel displays. He has authored and co-authored over 184 international journal and conference papers and 97 U.S. patents.