A Power and Data Link for a Wireless Implanted Neural Recording System Alexander Rush, Student Member, IEEE, EMBS, Philip R. Troyk, Senior Member, IEEE, EMBS
Abstract—A wireless cortical neural recording system with a miniature implanted package is needed in a variety of neuroscience and biomedical applications. Toward that end we have developed a transcutaneous two-way communication and power system for wireless neural recording. Wireless powering and forward data transmission (into the body) at 1.25Mbps is achieved using an FSK modulated Class E converter. The reverse telemetry (out of the body) carrier frequency is generated using an Integer-N PLL providing the necessary wide-band data link to support simultaneous reverse telemetry from multiple implanted devices on separate channels. Each channel is designed to support reverse telemetry with a data rate in excess of 3Mbps, which is sufficient for our goal of streaming 16 channels of raw neural data. We plan to incorporate this implantable power and telemetry system in a 1cm diameter single-site cortical neural recording implant.
Sending raw neural data from the implant allows one to flexibly change the spike sorting/spike detection algorithms in extracorporeal-based software, but comes at the expense of high data rate requirements. To send raw neural data from 16 channels, assuming an ADC resolution of 8bits/sample and a sampling rate of 20kSamples/s, a reverse telemetry data rate of at least 2.56Mbps would be required. To avoid the bandwidth limitations of an LSK system, it is necessary to have another link for reverse telemetry. This second link can use radiated emissions, optical coupling, or inductive coupling to send data out of the body. For our design, we chose to use an inductive link for reverse telemetry.
I. INTRODUCTION any neuroscience researchers as well as emerging prosthesis designs remain limited by the unavailability of wideband transcutaneous wireless neural recording. In the case of neuroscience research on animal subjects, an advantage of wireless neural recording is to remove the effect of tethering on the animal’s behavior. For neural prosthesis design, wireless recording has the advantage of reducing the risk of infection as well as device breakage.
M
A wireless neural recording system requires power and forward data to be transferred to the implant, and neural recording data to be transmitted from the implant. Both power and forward data can be implemented on a single inductive link, with forward data encoded as modulation of the power carrier. Reverse telemetry can be achieved on the same link by a method called load shift keying (LSK), but the data rate is generally limited to a fraction of the power carrier frequency, which is typically in the low MHz range. Furthermore, if multiple implants are powered by the external power coil, an external controller must timedivision multiplex (TDM) the reverse telemetry from the implants.
Manuscript received February 15, 2012. This work was supported by private donations to the Illinois Institute of Technology. A. Rush is with the Illinois institute of Technology, Chicago, IL 60616 USA (e-mail:
[email protected]). P. R. Troyk, is with Illinois institute of Technology, Chicago, IL 60616 USA (e-mail:
[email protected]).
Figure 1. (a) Schematic of bidirectional data transfer system (b) Physical diagram of dual inductive link coils modified from [1]
Ideally the implant would have a single-site, or “button” geometry [2], which would simplify implantation, and prevent potential complications resulting from tethering between multiple sections of the implant. For all wireless neural recording implants with multiple sections of which the authors are aware, one section resides on the skull or in a bone-seat and is tethered to the microelectrode array inserted in the cortex. An increased foreign body response has been observed in brain tissue to implants tethered to the skull [3,4]. Another possible complication is that the tethering
connection between multiple sections of the implant can fail due to wire breakage or deinsulation [5,6]. To achieve a single-site geometry for a dual inductive link, the power and data link must operate in the same volume-space. This necessitates the consideration of magnetic interactions between the power and data coils, because destructive paths of the reverse telemetry signal (out of phase with the constructive paths) can greatly reduce the amplitude of the signal received by the external data coil. This approach was reported by [7] for the design of a dual inductive link for power and forward data transmission for a retinal prosthesis. Design of both the power and data inductive links can be facilitated with the help of an analytic model of the inductive link electrical and performance parameters in terms of the link physical parameters [8-10]. This allows the physical parameters to be iterated on a computer rather than on the bench to find the optimal design within the physical restriction imposed. An analytic model of the link was used here to find the data coil radius which maximizes the effective coupling coefficient between the data coils, taking into account the contributions of the constructive and destructive reverse telemetry coupling paths between the data coils. It is highly beneficial to lock the reverse telemetry carrier to a multiple of the power carrier frequency using a phaselocked loop (PLL). This provides a convenient method for supporting simultaneous reverse telemetry from multiple implants powered by the same magnetic field. One can simply assign a different frequency division ratio to each implant. This method can also simplify demodulation of the reverse telemetry, because one can derive the reverse telemetry carrier from the power carrier frequency. Therefore, we have incorporated an Integer-N PLL into our integrated circuit design, which can generate outputs of 50, 60, 70, 80, 90, and 100MHz, from the 5MHz power carrier. The PLL cell design consumes less than 1.3mW below 100MHz, uses self-biasing techniques for supply rejection, and has dimensions of 350um x 680um. Fig. 1 shows a schematic of our power and bidirectional data transfer system. In this paper we present, our dual inductive link design methodology, implant and external circuitry design, as well as simulation and measurement results. Portions of this work have been previously presented in conference form [1,11,12]. II. SYSTEM IMPLEMENTATION The physical arrangement of the coils is illustrated in Fig. 1(b). For a typical implanted device, Coil 1 (L1) would be the external power coil, Coil 2 (L2) would be the implanted power coil, Coil 3 (L3) would be one of the external differential data coils, and Coil 4 (L4) would be the implanted data coil and is concentric to Coil 2. Power
transfer to the implant is achieved by generating a large AC current in Coil 1 using a Class E converter. AC current is induced in Coil 2, which is proportional to the coupling coefficient between the external and implanted power coils, k12. The resulting AC voltage is rectified to supply the application-specific integrated circuit (ASIC) with power and is also used to generate a reference clock for the ASIC. Forward data transfer is achieved by FSK modulation of the 5MHz power carrier at a data rate of 1.25Mbps in order to send control data to the ASIC. In the ASIC circuitry, the reference clock, derived from the 5MHz power carrier, is multiplied up by an integer-N PLL to generate a reverse telemetry carrier between 50MHz and 100MHz. The reverse telemetry is either amplitudeshift keying (ASK) or binary phase-shift keying (BPSK) modulated. On-chip driver circuitry induces current in Coil 4 to generate the reverse telemetry signal. According to simulation in PSpice A/D via OrCAD Capture CIS (Cadence Design Systems, San Jose, CA), with a power supply of 3V, the driver circuitry can drive 2.5mA peak-topeak current in Coil 4. Data is received by one of the two external differential data coils, Coil 3. A differential coil configuration is used to cancel both the large power signal at its fundamental frequency and harmonics generated by the Class E converter that fall within the frequency range of the reverse telemetry. III. ANALYTIC MODEL In order to avoid time-consuming design iterations on the bench, the dual coil link for power and reverse telemetry, illustrated in Fig. 1a,b, can be optimized with an algorithm which iterates the modifiable link parameters and chooses an appropriate combination of physical parameters which are associated with the best performance, as predicted by an analytic model of the link. This algorithm uses an expression similar to that presented in [7] for a dual coil system to provide power and forward telemetry to a retinal prosthesis. The expression was adapted for a dual coil system to provide power and reverse telemetry. The variables used for the electrical parameters of the link are the same as illustrated in Fig. 1a. The derivation is similar to [7], and space does not allow it to be included here. The assumptions critical to the derivation of the simplified equation for data magnitude, (1), are high quality factor coils and that the power coils are effectively short-circuited at the data carrier frequency (e.g., by a parallel capacitance). This expression for the magnitude of the reverse telemetry signal received by the external data coil, V3, shown in Fig. 1a is
V3 I 4 jdatakeff L3 L4
k eff k34 k 24k 23 k14k13 k14k12k 23 k 24k12k13
(1)
where I4 is the current induced in the implant data coil by
the coil driver circuitry, data is the angular frequency of the
reverse telemetry carrier, and L3, L4, k12, k13, etc. are as indicated in Fig. 1a. V3 and I4 are in phasor notation, so the ‘j’ in (1) indicates that the steady state sinusoidal voltage on Coil 3 leads the inverse of a sinusoidal current in Coil 4 by 90 degrees. According to (1), the data link can be optimized by maximizing the effective coupling coefficient, keff. By analyzing the dual coil link with this equation for k eff, we found that the optimal ratio of implanted power and data coil radii for our design was close to 0.8. To raise confidence in this idealized expression for the effective coupling coefficient between the data coils given in (1) as a performance metric, we compared values of V 3 simulated in PSpice A/D via OrCAD Capture CIS (Cadence Design Systems, San Jose, CA) including non-ideal, parasitic coil parameters (effective series resistance and selfcapacitance) to values of V3 calculated using the idealized equation (1) for ten values of implanted data coil radius, fixing all other physical parameters. The non-ideal coil parameters used for simulation, were calculated using our analytic model for these parameters, presented in [1], which space does not allow to be included here. The physical parameters which were assumed for the data presented in Fig. 2, while the radius of the implanted data coil was varied, are summarized in Table I. The current in the data coil was modeled as a sinusoid with a peak amplitude of 1mA, and coil separation was set to 1cm.
chose 8mm as the optimal diameter for Coil 4 for the 10mm diameter of Coil 2 assumed. In order to test the analytic model, we measured V3 as a function of separation, and compared the measured values to the values calculated with (1) and the equations for self and mutual inductance as a function of the link physical parameters. Again, the parameters listed in Table I were assumed, and Coil 4 was made with a diameter of 8mm. As shown in Fig. 3, the measured values closely match the calculated values. These measurements were made with a test board designed to minimize parasitics, and a custom XYZ positioning system, which has been fabricated for testing inductive link systems in our laboratory. This XYZ positioning system consists of three manual linear actuators fastened together. Each linear positioner has a millimeter scale for accurate measurement.
Figure 3. Comparison of measured values for reverse telemetry data signal amplitude and the values calculated using (1) and the equations for self and mutual inductance as a function of the link physical parameters.
More details on the analytic model of the dual inductive link, such as coil self- and mutual-inductance, selfcapacitance and effective series resistance (ESR) calculations are given in [1]. TABLE I.
PHYSICAL PARAMETERS ASSUMED FOR DESIGN
Coil 1
Coil 2
Coil 3 0.42 Length Length Length mm Length 5 Radius 3cm Radius of Long mm Side N.A.Length Insulation Insulation 5 Litz of Short Thickness Thickness μm Wire Side 9 mm
Figure 2. Comparison of simulated values for reverse telemetry data signal amplitude using non-ideal coil parameters (effective series resistance and self-capacitance) and the values calculated using the equation (1), which was derived assuming nearly ideal coils [1].
As shown in Fig. 2, the simulated values for V3 using nonideal coil parameters (effective series resistance and selfcapacitance) closely match the values calculated for V3 using (1), which was derived assuming nearly ideal coils (high Q, negligible self-capacitance). Based upon these results, we
Wire Wire 25 N.A. Diameter Diameter μm Turns Per Layer
3.5
# of Layers
2
Turns Per 12 Layer # of Layers
3
Trace Width Turns Per Layer # of Layers
Coil 4 0.043 mm
Length
0.42 mm
19 mm
Radius
Varied, See Fig. 2
15 Insulation mm Thickness
5 μm
0.51 Wire mm Diameter
25 μm
1
Turns Per Layer
12
1
# of Layers
1
TABLE II. Electrical Parameter L1 L2 L3 L4
INDUCTANCE VALUES [1] Theoretical — 31.7 μH 0.0573μH 2.76uH
Measured 4.62μH 32.4μH 0.055μH 2.98uH
IV. COIL FABRICATION AND MEASUREMENT The data coil and power coil were wound upon a customfabricated coil form using 50 American wire gauge (AWG) gold wire and subsequently wire bonded to a printed circuit board (PCB) for testing of electrical parameters and interfacing with the implant circuitry. Under the assumption that inductances and coupling coefficients are primarily determined by coil geometry and spacing, these parameters were measured at 1MHz with a 1260 Impedance/GainPhase Analyzer (Solartron Analytical, Farnborough, UK). The measured and theoretical values of the coil inductance are given Table II. The inductance of the external power coil, L1, was measured from an existing Class E inductor in use.
We have explored two different approaches to reduce harmonics in the external power coil. One method is to place a low-pass filter in the series tank circuit as illustrated in Fig. 5a. The other approach, illustrated in Fig. 5b, is to place a notch filter in the series tank of the Class E converter to attenuate the harmonic distortion at the reverse telemetry carrier frequency. Due to the small ratio between the reverse telemetry carrier frequency and the power carrier frequency, the corner frequency of the low-pass filter could not be brought low enough to attenuate the harmonic distortion significantly without disrupting the operation of the series resonant tank of the Class E converter. Therefore, we chose to use the notch filter method. Using this approach, the 12 th harmonic (60MHz) which coincides with the reverse telemetry carrier, was attenuated by 15dB.
V. CLASS E CONVERTER The magnetic field for inductive powering was generated by a Class-E converter transmitter operating at 5MHz. The transmitter coil carried a peak current of 0.65A, had a radius of 3cm and 8 turns of 2MHz litz wire (New England Wire Corporation, Lisbon, NH). Due to the large size of the power signal compared to the reverse telemetry signal, even small amounts of harmonic distortion, occurring at integer multiples of the power carrier frequency, can obscure the reverse telemetry signal. Another source of interference can be the transmitter gate drive, which can couple to the external data coil from the gate-drain capacitance of the Class-E field-effect transistor (FET). Harmonic distortion resulting from normal operation of the Class-E converter and from the gate drive signal is illustrated in Fig. 4.
Figure 4. Class E harmonic interference during normal operation and from coupling of the gate drive signal into the series LC branch of the Class E converter [11].
Figure 5. Methods of filtering Class E harmonics from the series LC branch of the Class E tank circuit (a) Low-pass filter in the series tank circuit (b) Notch-filter in series tank circuit [11].
VI. DIFFERENTIAL ANTENNA The external data receiver chosen was a pair of “bucked” coils connected in parallel and anti-phase. In other words the inner leads were connected together and outer leads were connected together and grounded. This has the effect of canceling both distant sources of RF magnetic interference as well as nulling the 5MHz power carrier provided that the bucked coils are carefully aligned with Coil 1. The receiver coils could have been connected in series, in what is known as a “figure-8” configuration. However, we found that this made our receiver front-end susceptible to noise and feedback. Therefore, we used the parallel coil configuration. However, this required that we place a high-pass filter in series with each of the bucked coils to minimize induced power-carrier current which would have loaded the transmitter and reduced the powering magnetic field at the implant. A photograph of the differential reverse telemetry receiver antenna is shown in Fig. 6. The detection of the reverse telemetry data signal is maximal at the center of either of the bucked coils and very small at the shared edge of the bucked coils. The cancellation of harmonics generated by the Class E converter, which fall within the bandwidth of the reverse telemetry signal, is illustrated in Fig. 7.
technology that integrates bipolar junction transistors and complementary metal-oxide-semiconductor transistors). The PLL has a programmable output frequency to allow multiple implanted devices to send reverse telemetry from roughly the same physical location. Specifically the frequency divider is designed to synthesize voltagecontrolled oscillator (VCO) outputs of 50, 60, 70, 80, 90 and 100MHz depending on the value of a 4-bit control word. Producing these frequencies required division-by-two followed by division by 5, 6, 7, 8, 9, or 10.
Figure 6. Photograph of differential reverse telemetry receiver.
Figure 7. Illustration of harmonic interference nulling by the differential reverse telemetry receiver [11].
Figure 9. PLL (and ASK/BPSK Transmitter in lower right) integrated circuit (IC) Die Photograph. The total die area was 1600umx1450um including the pads and ring. The PLL dimensions were 350um x 680um [12].
VII. IMPLANT CIRCUITRY An application specific integrated circuit (ASIC) was designed to implement the circuit portion of the wireless power and data system, which, for an implanted device, would be located inside the body. As shown in Figs. 8 & 9 the ASIC contains a fully integrated rectifier, a PLL, modulators (ASK and BPSK), and reverse telemetry drivers. The external circuitry for wireless powering and two-way communication is also presented. The integrated circuit was fabricated in the X-FAB (Lubbock, TX) 800nm BiCMOS process [13] (BiCMOS is a term for a semiconductor
Self-bias techniques and a differential buffer were used to improve the power supply rejection of the VCO as described in [14]. A more complete description of the PLL circuitry and its measured performance is given in [12]. The 5MHz reference clock is recovered from the magnetic power field carrier by the on-chip FSK demodulator circuitry which also demodulates commands sent to the RF telemetry ASIC by the transmitter via FSK modulation of the power carrier at a data rate of 1.25Mbps as described in [15].
Figure 8. Schematic of Telemetry IC Multi-Channel Wireless Neural Recording System [12]
TABLE III WIRELESS NEURAL RECORDING SYSTEMS AND CHARACTERISTICS OF THEIR POWER AND DATA SYSTEMS Implantability (According to authors’ stated plans)
Power Source
Data Transmission Method
Raw Neural Recording or Spike Detection
Data Rate
Data Link Energy per bit
WINeR System, Georgia Institute of Technology [16]
No
Inductive
Radiated, ISM band at 915MHz
Raw Neural Recording
58709kSps
607pJ/b
Brown University System [2]
Yes, Two Island (Two-Site) Geometry
Inductive
Optical
Raw Neural Recording
N/A
N/A
INI System, U of Utah and Stanford [17]
Yes, Button (SingleSite) Geometry
Inductive
Radiated, ISM Band 902-928MHz
Spike Detection
157kbps
3185pJ/b
Hermes System, U of Utah and Stanford [18]
No
Battery
Radiated, 3.7~4.1GHz
Raw Data
24Mbps
1250pJ/b
UCSC System [19]
Yes, At Least Two Site Geometry
Battery
Radiated, Impulse radio based UWB, 4GHz
Raw Neural Recording
90Mbps
17.78pJ/b
University of Michigan, Ann Arbor System [20]
Yes, Two Island Geometry
Inductive
Coil Antenna, 70200MHz
Spike Detection
2Mbps
N/A
This Work
Yes, Button Geometry
Inductive
Coil Antenna, 50100MHz
Raw Neural Recording
3Mbps
1962pJ/b
Wireless Neural Recording System
TABLE III
In simulation, the PLL power consumption was below 1.3mW for frequencies below 100MHz with a 3V supply. The power consumption of the fabricated ASIC was measured to be 4.2mW operating at 48MHz with a 2.7V supply, and the measured power consumption closely matched OrCAD simulations. Simulations predict that the PLL consumes only 14% of the total power, while the reverse telemetry driver consumes 74% of the total power. The reverse telemetry driver circuitry included on the chip is not optimized for low power because it has redundant modulation and driver circuitry that was used for evaluation purposes.
and data link with a distance of 20mm and a data rate of 3Mbps is given in Fig. 10.
VIII. DEMODULATION OF THE REVERSE TELEMETRY Although synchronous demodulation might be easily achieved, since the outward RF signal carrier is phase locked to the RF power carrier, for initial testing the reverse telemetry signal was asynchronously demodulated. This is achieved using a bandpass filter centered at the reverse telemetry carrier frequency, followed by a logarithmic amplifier with a large dynamic range. To facilitate digital algorithm testing, the finite impulse response (FIR) filter and data synchronization system was implemented on the Cyclone III DSP Development Kit (Altera, San Jose, CA). With a coil separation of 20mm and a data rate of 3Mbps, the bit error rate (BER) was measured to be 2.03e-4. Measured data of the dual inductive power
Figure 10. Demonstration of ASK transmission and demodulation with a coil separation of 20mm and a data rate of 3Mbps. The top trace is the modulating data signal to the implant circuitry for reverse telemetry. The second trace from the top is the log amp output. The third trace from the top is the digitally filtered (and inverted) log amp output. The bottom trace is the demodulated reverse telemetry data.
We believe 20mm is a separation distance that is adequate for most cortical neural-recording implants based on a conservative estimate of the separation imposed by anatomy for an adult male human, taking into account the thickness of the scalp (≈8mm maximum [21]) and skull (≈11mm [22]), as well as the dura and subdural space [23].
Having shown that the wireless power and telemetry system was capable of delivering the required data rate and coil separation, the wireless power and telemetry circuitry was then demonstrated in the context of a prototype 4 channel wireless neural recording system. The overall architecture of a prototype wireless neural recording array (WNRA) circuitry is shown in Fig. 11.
Figure 12. Simplified block diagram of the DSP filtering and data synchronization system used to demodulate reverse telemetry from the prototype four channel WNRA circuitry.
Figure 11. Overall architecture of our prototype wireless neural recording array (WNRA) circuitry. This ASIC includes neural recording amplifiers and a specialized voltage regulator to reduce power supply noise and assure reliable operation even with expected variations in the powering magnetic field strength. Coil A, B are the inputs connected to the leads of the implant power coil, and Coil C, D are the outputs connected to the leads of the implant data coil. ModSel is used to choose between BPSK and ASK modulation. Vdd is the shunt regulated power supply and input to a low dropout voltage regulator, the output of which is VLV.
The purpose of this prototype WNRA circuitry was to evaluate the wireless power and reverse telemetry link in the context of a wireless neural recording system with an architecture representative of systems into which the power and data link will ultimately be incorporated. The prototype WNRA circuitry used available amplifiers and ADC circuitry. No attempt was made to minimize the noise on the amplifiers, nor were comprehensive noise measurements made. A simplified illustration of the DSP filtering and data synchronization system used to demodulate the reverse telemetry from the prototype four channel wireless neural recording circuitry is given in Fig. 12. This system was implemented on an FPGA so changes could be made quickly by reprogramming, which is preferable to a digital ASIC or analog system, which would require new hardware to be ordered. The Cyclone III DSP Development Board and Cyclone III Data Conversion HSMC (Altera Corporation, San Jose, CA) were used for rapid prototyping purposes. However, ultimately only an ADC, the FPGA, and a flash memory would be required, or the design could be converted to an equivalent ASIC or discrete circuit design.
Although the prototype WNRA circuitry is still under evaluation, a set of preliminary measurements were made. The gain and high pass filter corner of the amplifiers were set to their lowest possible values. The gain was measured to be 62.3 and the bandwidth was measured to be 0.7Hz to 27kHz. The waveforms were digitized by the ADC on the prototype WNRA circuitry at a rate of 20kSps (the potential for aliasing should be fixed in future versions of the chip with an adjustable low pass corner) and with a resolution of 8bits/Sample. A 10mV, 3kHz sinusoidal input was presented to the four electrode inputs of the prototype four channel wireless neural recording array circuitry. Reverse telemetry data was transmitted with a coil separation of 15mm, a carrier frequency of 40MHz, and a data rate of 1.25Mbps. The reverse telemetry was successfully demodulated, and the amplitude of the wirelessly transmitted and decoded amplifier outputs were found to be within the expected range. IX. DISCUSSION We have shown how a dual inductive link for transcutaneous wireless power and two-way communication can be optimized using an analytic model of the inductive link in terms of its physical parameters, avoiding timeconsuming design iterations on the bench. No publication existed previously for the optimization of reverse telemetry achieved using a dual inductive link with coaxial implanted coils. The coaxial coil arrangement is attractive, because the data coil fits within the area of the power coil, and, therefore, does not increase the total area of the implant. Also, in contrast to the approach where the data coils are made to be orthogonal to the power coils the thickness of the implant is not increased by the presence of the data coils. Also, a greater coupling coefficient is achieved with the coaxial dual inductive link method than with the orthogonal dual inductive link method. Our aim was to incorporate the power and two-way communication system into an implant with a diameter of 1cm. A small implant diameter allows more implants in a given area of cortex, all of which could be powered by a
single external power coil. Also, implantation may be achieved more easily with an implant having a smaller footprint. The 1cm diameter restriction on coil diameter presents a challenge in delivering sufficient power to the implant and reverse telemetry amplitude outside the body, because, with the coil separation imposed by anatomy (mainly skull and scalp) a small implant diameter results in small coupling coefficients between the external and implant coils. Therefore, the analytic model for a dual inductive link was used to achieve a working power and reverse telemetry system having implant coil diameters of no more than 1cm. Techniques for reducing and mitigating harmonic interference from the power link onto the data link were also presented, without which, the reverse telemetry signal would be obscured. Particularly important was the use of differential coils for detecting the reverse telemetry signal from the implant data coil, while rejecting harmonic interference from the external power coil. Historically, differential coils, have been used to reject the large power carrier rather than to reduce harmonic interference from the power link onto a separate data link operating in the same space. For instance, differential coils have been used in a dual inductive link for delivering power and forward telemetry to an implant. In this case the purpose of the differential coil was primarily to reduce the size of the filters that would to reject the power carrier in the implant circuitry, where size is strictly limited by anatomy. An end-to-end demonstration of a prototype wireless neural recording array (WNRA) circuitry has shown that the wireless power and reverse telemetry link is functional in the context of a representative wireless neural recording system. Although the preliminary measurements of the 4-channel prototype WNRA circuitry presented here would be considered rough for the purpose of demonstrating a complete neural recording system, the purpose of the end-toend demonstration was to demonstrate the wireless power and telemetry link which has been a historic problem. In continuing work the amplifier and ADC need to be optimized for noise and accurate gain control. However, there are numerous examples in the literature of low noise, low power amplifiers which have been incorporated into neural recording circuitry, so we do not anticipate any obstacles in designing amplifiers and ADC circuitry appropriate for a first generation wireless neural recording system [2,16-20]. Details regarding the layout, materials, and characteristics of the electrode array, which will be used with a first generation recording system utilizing the wireless power and two-way communication system presented here is also beyond the scope of this paper. However, we anticipate that the neural recording system will be designed to work with a variety of electrode types. Table III summarizes several of the most developed wireless neural recording systems for comparison and some essential characteristics of their power and data systems. One strength of the wireless neural recording system presented here is the use of a single site, “button”, geometry,
which would allow for implantation of the entire recording system beneath the dura. A single site implant does not require connections between multiple sections of the implant, which simplifies implantation and eliminates any potential damage which may result from tethering and the relative motion of the skull and brain. The INI system, summarized in Table III, also uses a button geometry. However, in contrast to the INI system, which sends spike detected data, the wireless neural recording system presented here is intended for sending raw neural data on all channels simultaneously, which requires a much higher data rate. The system presented here is also designed to allow multiple implants located in the magnetic field of a single Class E powering coil to send reverse telemetry simultaneously on separate channels (e.g. 50, 60, 70, …, 100 MHz). X. CONCLUSION The system presented here is an important step for providing two-way communication and wireless power in a wideband transcutaneous neural recording system. The dual inductive link was optimized using an analytic model of the link in terms of its physical parameters. Novel methods were used to reduce interference between the power and reverse telemetry link, including filtering of harmonics from the Class-E converter tank, and use of a differential reverse telemetry receiving coil to cancel transmitter harmonic and far-field interference. The wireless power and data system was demonstrated in the laboratory by fabricating an inductive link and pairing with an ASIC. Operation is presented for a separation of 20mm and a data rate of 3Mbps, which we believe would be sufficient for a neural prosthesis utilizing cortical neural recordings. REFERENCES [1]
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Alexander D. Rush (M’09) received the B.S. degree in electrical engineering from University of Illinois at Urbana-Champaign in 2006 and the Ph.D. degree in biomedical engineering from Illinois Institute of Technology in 2012. He is currently developing new hardware for neural recording and stimulation at Plexon Inc in Dallas, TX. His research interests include analog, digital and mixed signal circuit design, neuroprosthetic devices, wireless power and telemetry for implantable electronics, and optogenetics.
Philip R. Troyk (M’83–SM’91) received the B.S. degree in electrical engineering from the University of Illinois at Urbana-Champaign, in 1974, and the M.S. and Ph.D. degrees in bioengineering from the University of Illinois, Chicago, in 1980 and 1983, respectively. He was on the staff of Northrop Corporation, Rolling Meadows, IL, from 1973 to 1981. In 1983, he joined the faculty of the Illinois Institute of Technology, where he is currently Associate Dean of the Armour College of Engineering, Associate Professor of Biomedical Engineering, and Director of the Laboratory of Neural Prosthetic Research. He is also president of Sigenics, Inc, a company involved with design of ASICs for medical use. At IIT he leads a team for development of an intracortical visual prosthesis, and directs IIT’s contribution towards development of the IMES for prosthesis control. His broader interests include development of central and peripheral neural prostheses, the design and packaging of electronic assemblies for implantation in the human body, and polymeric protection of thin film devices operating in high humidity environments.
