A Relaxation Oscillator-Based Transformer Ratio Arm ... - IEEE Xplore

3414

IEEE TRANSACTIONS ON INSTRUMENTATION AND MEASUREMENT, VOL. 64, NO. 12, DECEMBER 2015

A Relaxation Oscillator-Based Transformer Ratio Arm Bridge Circuit for Capacitive Humidity Sensor Tarikul Islam, Shakeb A. Khan, Md. Firoz A. Khan, and Subhas Chandra Mukhopadhyay, Fellow, IEEE Abstract— A simple signal conditioning circuit using a transformer ratio arm (TRA) bridge for converting capacitance change into frequency for capacitive sensors is presented. The circuit employs a relaxation oscillator in which the output frequency is linearly related to the capacitive unbalance of a TRA bridge. The design, analysis, and experimental results of the circuit and its application to a thin-film-based humidity sensor are reported. The experimental results confirm the theoretical value predicted. The circuit which offers the minimum parasitic earth capacitance effect has the potential for accurately monitoring measurement parameters, particularly ppm-level humidity. The simulation results for the effect of parasitic earth capacitances and ambient temperature on the output frequency have also been discussed. The pulse wave output of the circuit is interfaced with microcontroller for direct moisture display in ppm. The frequency sensitivity and nonlinearity of the sensor for the 0–110-ppm moisture range are found to be 10.94 Hz/ppm and 1.2%, respectively. Index Terms— Capacitance to frequency conversion, capacitive sensor, moisture measurement, relaxation oscillator, transformer ratio arm (TRA) bridge.

I. I NTRODUCTION APACITIVE sensors are used for measuring various nonelectrical quantities such as force, humidity, pressure, vibrations, displacement, level, and so on because of their small size, accuracy, low power consumption, and low temperature dependence. Detection electronic circuit is desired to be simple to convert the change in capacitance into voltage or frequency [1]–[3]. Continuous effort has been made to develop an efficient and simple technique to measure the liquid level using capacitive method [4] or detection of objects [5]. Some recent applications of interdigital electrode planar-type capacitive impedance sensor include monitoring of water pollutants [6], [7] and quality of dairy products [8]. Capacitive technique is used for nondestructive measurement of moisture content in grains, nuts, and so on [9]. Today, most of the humidity sensors employ capacitive technique utilizing

C

Manuscript received February 16, 2015; revised June 11, 2015; accepted June 13, 2015. Date of publication September 1, 2015; date of current version November 6, 2015. This work was supported by the Department of Science and Technology, New Delhi, India, under Grant SR/S2/CMP-011/2011. The Associate Editor coordinating the review process was Dr. Dario Petri. (Corresponding author: Tarikul Islam.) T. Islam and S. A. Khan are with the Department of Electrical Engineering, Faculty of Engineering and Technology, Jamia Millia Islamia University, New Delhi 110025, India (e-mail: [email protected]). M. F. A. Khan is with the Department of Electrical Engineering, Jamia Millia Islamia University, New Delhi 110025, India. S. C. Mukhopadhyay is with the School of Engineering and Advanced Technology, Massey University, Palmerston North 4410, New Zealand (e-mail: [email protected]). Color versions of one or more of the figures in this paper are available online at http://ieeexplore.ieee.org. Digital Object Identifier 10.1109/TIM.2015.2459473

different electrode structures with different types of porous and nonporous sensing materials [10]–[14]. The commercial hyperthin-film humidity sensors in ppm level employ capacitive method in different industrial applications [15]. Recently, several works have been reported to utilize the capacitive method for fabricating sol–gel thin-film γ -Al2 O3 -based moisture sensor in the trace level [16]. Many capacitive sensors have a very small change in capacitance with a relatively large initial offset capacitance [17]. Capacitive sensors that work on dielectric adsorption technique suffer from high zero offset capacitance and offset drift due to aging [11], [17], [18]. The accuracy of measurement is affected by various factors such as parasitic earth capacitance, ambient temperature, and op-amp offset voltage [1], [3], [8], [17]–[19]. Variation of ambient temperature affects the measurement accuracy for most of the capacitive humidity sensors [11], [20]. Smart sensors with on-board analog to digital converter and digital output in the form of frequency or duty cycle or pulsewidth-modulated wave will be increasingly useful in the near future. The digital form of signal provides high noise immunity, low output signal power, wide dynamic range, high accuracy of frequency standards, simplicity of communication, and ease of interfacing, integration, and coding [19]–[23]. Relaxation oscillator-based signal conditioning circuit having op-amp-based integrator and comparator is widely applied for interfacing resistive or capacitive sensors [19], [20], [24]–[26]. In such circuits, the capacitance changes or resistance changes are detected in digital form, which can be easily interfaced with microcontroller for further signal processing [9], [11], [15], [16]. In this paper, a simple signal conditioning circuit using transformer ratio arm (TRA) bridge and relaxation oscillator for capacitive humidity sensors has been presented. The frequency output of the oscillator is directly related to the change in capacitance of the sensors. It can be used to measure small capacitance change neglecting very high offset capacitance drift and parasitic earth capacitances [1]–[3], [17]. The circuit employs TRA bridge to convert the differential capacitance change into frequency. Three-winding TRA bridge is one of the oldest and accurate techniques to measure small capacitance with earth capacitance elimination [1], [2], [27]. It is still treated as the most accurate and stable method for capacitance measurement [1], [27]. The working principle of the interfacing circuit has been described. Experiments have been performed with discrete ceramic capacitance and capacitive moisture sensor to convert the incremental capacitance into change in frequency. The moisture sensor is based on sol–gel thin film of γ -Al2 O3 . The fabricated moisture

0018-9456 © 2015 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission. See http://www.ieee.org/publications_standards/publications/rights/index.html for more information.

ISLAM et al.: RELAXATION OSCILLATOR-BASED TRA BRIDGE CIRCUIT FOR CAPACITIVE HUMIDITY SENSOR

Fig. 2.

Fig. 1. (a) Schematic of the signal conditioning circuit. (b) Three-terminal representation of the capacitive sensor.

sensor is suitable for trace moisture measurement [16]. For this capacitive sensor, the initial offset capacitance at dry moisture is large and the capacitance change due to change in moisture in ppm is small [16], [28]. Simulation in MATLAB has been carried out to study the effects of parasitic capacitances and ambient temperature on the output frequency. The emphasis of the proposed circuit is to interface capacitive-type sensor widely employed in humidity detection [11]. The capacitance value of humidity sensor may vary from few picofarads to several thousands of picofarads. Measurement of small capacitance in picofarads is an issue because of error due to parasitic earth capacitance or lead capacitance. The initial offset capacitance and change in capacitance due to moisture depend on: 1) electrode structure and size; 2) thickness of the sensing film; 3) gap between the electrodes; and 4) pore morphologies of the nanostructure [13]. II. W ORKING OF THE S IGNAL C ONDITIONING C IRCUIT The schematic of the circuit is shown in Fig. 1(a). Fig. 2 shows the waveforms at different terminals of the circuit. Part A of Fig. 1(a) enclosed by the dotted block is the capacitive TRA bridge. The TRA can be formed using audio frequency center tap transformer. The primary of the transformer is excited by the square output of the comparator of the relaxation oscillator. At the secondary side of the transformer, two voltages Vx and −V x are almost equal in amplitude and exactly 180° out of phase. Transformer is normally used for isolation in high-frequency communication circuit for minimizing loading effect. A simple center tap transformer with transformer ratio (η < 1) can reduce the secondary voltage to a suitable level. The output of the TRA is connected to the capacitors Cs and C x , where C x is the sensing capacitance and Cs is the reference capacitance. The sensor is enclosed in a grounded conducting shield (metallic sensor chamber).

3415

Significant waveforms at different output terminals of the circuit.

The C x can be represented equivalently by a π-network given in Fig. 1(b). C x1 and C x2 are the two parasitic earth capacitances for C x . Similarly, Cs can also be represented by π-network with parasitic capacitances Cs1 and Cs2 , respectively. C x1 and Cs1 capacitances appear across the equal arms of the transformer, and (C e = C x2 + Cs2 ) is the parasitic capacitance at the detector terminal P. If the impedances of the transformer primary excitation V Rb and secondary voltage sources ±V x are low, then the effect of C x1 and Cs1 for bridge balance is negligible. This is because these two capacitances appear across the voltage sources that are having low impedances [1], [3], [27]. This condition can be easily achieved by selecting transformer with highpermeability core. The detector terminal P of the op-amp is at virtual ground, and if the common point of the three-winding transformer is at the ground potential as shown in Fig. 1(a), then the parasitic capacitance Ce can be neglected. The unbalance detector current I3 can be measured by a current meter or by a voltage meter after converting the detector current into voltage. However, in voltage mode of measurement, care should be taken to reduce the effect of the parasitic capacitance Ce as shown in Fig. 1 [24]. In the present circuit, detector current due to change in C x is converted into voltage signal Vcf using a capacitor C f at the feedback path of the op-amp A1. Circuit of the relaxation oscillator for capacitance to frequency conversion is shown in Part B of Fig. 1(a). The detector output Vcf of the capacitance bridge with a buffer and the comparator output VRb are applied as inputs to the inverting summing integrator of the oscillator. The output of the integrator is connected to the positive input terminal of the comparator. When the output of the bridge circuit is connected to the summing integrator, the square-wave current IRg flowing through Rg is added with another square-wave current I R0 flowing through R0 . Applying KCL at node P and converting unbalance current into voltage Vcf [3], [27] 1 Vcf = −(I1 + I2 ) . (1) j ωC f As I1 = j ωCs Vx and I2 = − j ωC x Vx Vcf = −( j ωCs Vx − j ωC x Vx )

1 C x −Cs or Vcf = Vx . j ωC f Cf (2)

3416


Replacing Vx in (2) by ηV Rb Vcf = δηV Rb .

(3)

δ = (C x − Cs /C f ), and η is the transformation ratio of the voltage from the primary to the secondary side. For C x = Cs , the bridge is balanced and Vcf = 0. A. When C x Is Replaced by Capacitive Humidity Sensor When the measured capacitance C x is replaced by the capacitive humidity sensor with C x = C x + C x , C x is the change in capacitance due to the change in humidity. Humidity sensor is based on the change in effective dielectric constant so that the capacitance of the sensor changes in proportion to the change in humidity [16], [28]. Equation (2) can be written as (C x + C x ) − Cs Vcf = Vx Cf or C x − Cs C x Vcf = Vx (4) Vx + Cf Cf

Fig. 3.

Equivalent circuit of the Part A of the proposed TRA Bridge.

During the unbalance TRA bridge condition, the integrator output can be written as Vc0 =

C x C x Vx = ηVRb . Cf Cf

T =2 or

(5)

Equation (5) shows that the detector output voltage is directly proportional to capacitance change C x . Therefore, the change in voltage is directly proportional to the change in capacitance of a sensor and, thus, humidity. III. C APACITANCE TO F REQUENCY C ONVERTER The circuit of the relaxation oscillator for capacitance to frequency conversion is shown in part B of Fig. 1(a). If the peak voltage level of comparator output |+V s | = |−V s | = Vs is symmetrical, the output voltage VCO ( p− p) of the integrator is a triangular wave with peak-to-peak amplitude 2(Ra /Rb )Vs set by the comparator hysteresis. The details for the expression VCO ( p − p) = 2(Ra /Rb )Vs can be found in [19] and [26]. The output switches in half of the time period T /2. The integrator equation T /2 T /2 1 1 Vcf dt − VRb dt. (6) Vco = − Rg C0 0 R0 C 0 0 At the bridge balance condition, Vcf = 0 and replacing VRb by −Vs shown in Fig. 2 Vco = −

Vs 1 T (−Vs ) = T. R0 C 0 2 2R 0 C0

(7)

Equating Vco ( p − p) = 2(Ra /Rb )Vs with (7), the time period T is given by Ra (8) T = 4R0 C0 Rb and the frequency of the triangular (or square) wave at the balance condition Rb f0 = . (9) 4R 0 C0 Ra

(10)

and the time period and the frequency of the output signal can be given by

where Vcf is the modified voltage output of the detector. Then, the change in voltage Vcf = Vcf − Vcf =

T T ηδVs + Vs 2Rg C0 2R0 C0

f =

Rb 2R a

Ra Rb

1 1 2R0 C 0

+

ηδ 2Rg C 0

(11)

1 R0 ηδ = f0 1 + + ηδ . 2R0 C0 2Rg C0 Rg

(12)

A. Frequency Relation for Humidity Sensor For C x = Cs , the bridge is in the balance condition at the central frequency f = f 0 , and when C x = C x + C x (humidity sensor) δ= and

C x Cf

R0 C x f = f0 1 + η . Rg C f

(13)

Thus, the change in frequency is directly proportional to the change in capacitance of the humidity sensor. IV. E RROR A NALYSIS A. Error in Output Frequency Due to Parasitic Earth Capacitances The equivalent circuit of the TRA bridge (Part A) is shown in Fig. 3 [1], [2], [31]. The expressions for reactances using the Laplace operator are X Cs =

1 , sC s

XCx =

1 , sC x

X Cs1 =

1 , sC s1

X C x1 =

1 sC x1

and X (AC f ) =

1 . s AC f

(14)

The voltage across the parasitic capacitance Cs1 is VC =

Vx sC1s1 RS +

1 sC s1

=

Vx 1 + s R S Cs1

(15)


and across C x1 is Vt =

−V x sC1x1 RS +

1 sC x1

=

−Vx . 1 + s R S C x1

(16)

The currents through the capacitances Cs and C x are I1 = (VC + V )sC s

(17)

I2 = (Vt + V )sC x .

(18)

Let C T = Cs2 + C x2 + AC f and the reactance X CT =

1 1 = s(C s2 + C x2 + AC f ) s(C e + AC f )

which is the same expression shown in (4) obtained for an ideal op-amp condition. Equations (7) and (24) can be combined as ⎤ ⎡ C Cs x T /2 AηV Rb ⎣ 1+s R S C x1 − 1+s R S Cs1 ⎦ −1 Vc0 = Rg C0 0 1 + s Z 0 C f Cs + C x + (Ce + AC f ) T /2 1 VRb dt. − R0 C 0 0 Integrating over the time period 0–T /2 and replacing VRb by −V s VCO =

where Ce = Cs2 + C x2 Ze = I3 =

1 Z in s(Ce +AC f) 1 Z in + s(Ce +AC f )

=

Z in 1 + s Z in (Ce + AC f )

−V (1 + s Z in (Ce + AC f )) −V = . Ze Z in

Applying KCL at node P Cs Cx − I1 + I2 = I3 sV x 1 + s R S Cs1 1 + s R S C x1 s Z in (Ce + AC f ) = −V s[Cs + C x ] − V Z in where |s Z in (Ce + AC f )| 1 Cx Cs Vx − 1 + s R S Cs1 1 + s R S C x1 = −V [Cs + C x + (Ce + AC f )].

(19) (20)

(21)

AV sC1 f Z0 +

1 sC f

=

AV . 1 + s Z 0C f

Putting V from (22) in (23), Vcf is ⎤ ⎡ C Cs x V − x 1+s R S C x1 1+s R S C s1 A ⎣ ⎦. Vcf = 1 + s Z 0C f Cs + C x + (Ce + AC f )

T AηVs × 2R g C0 1 + s Z 0C f ⎤ ⎡ Cx Cs − 1+s R S C x1 1+s R S C s1 T ⎦+ ×⎣ Vs . Cs + C x + (Ce + AC f ) 2R 0 C0

2

Ra T AηVs Vs = × Rb 2R g C0 1 + s Z 0C f ⎤ ⎡ C Cs x 1+s R S C x1 − 1+s R S C s1 ⎦ + T Vs . ×⎣ Cs + C x + (Ce + AC f ) 2R 0 C0

Thus, the time period of the output signal ⎡ 2Ra Rb

⎢ ⎢ ⎢ ⎢ ⎣

Aη 1 2Rg C 0 × 1+s Z 0 C f

⎤

1 Cx 1+s R S Cx1

s − 1+s C R C

S s1

C s +C x +(C e +AC f )

+ 2R10 C0

(24)

For ideal op-amp conditions s Z 0 C f 1 as Z 0 is very small, C f is in picofarads. If f 0 = 104 Hz, C f = 10−10 F, Z 0 = 100 , and A = open loop gain of the op-amp (ideally infinite), then s Z 0 C f = 104 × 102 ×10−10 = 10−4 1. Similarly, R S —the source resistance is negligible. For C x varying from 500 to 600 pF typically for trace moisture capacitive sensor, Cs + C x + (Ce + AC f ) is approximated as AC f . If the reference capacitance Cs = 500 pF at dry moisture condition and the parasitic capacitance Ce = C s2 + C x2 = 200 pF (at most), then (24) can be approximated as Vx [C x − Cs ] (C x − Cs ) Vs = Vcf = A AC f Cf

⎥ ⎥ ⎥. ⎥ ⎦

(26)

(22)

(23)

(25)

Equating VCO ( p − p) = 2(Ra /Rb )Vs with (25)

T =

The voltage Vcf at the output of op-amp A1 in Fig. 1(a) is Vcf =

3417

And the frequency of the output signal is ⎡ Aη Rb ⎣ 1 × f = 2Ra 2Rg C0 1 + s Z 0C f ⎤ ⎤ ⎡ Cx Cs − 1+s R S C x1 1+s R S C s1 1 ⎦+ ⎦. × ⎣ Cs + C x + (Ce + AC f ) 2R0 C0

(27)

B. Error in Output Frequency Due to Change in Ambient Temperature The temperature coefficient in capacitance (TCC) describes the maximum change in capacitance over a specified temperature range. The capacitance value is specified by a manufacturer normally at a reference temperature of 25 °C. TCC should always be taken into consideration for the application of capacitance below or above this reference temperature [1], [29]. Class 1 capacitors (temperaturecompensating-type monolithic ceramic capacitors) are highly stable with temperature and are referred as temperature compensating. The TCC specification for class 1 capacitors is of the order of ppm/°C. The capacitance change due to temperature variation is calculated by C(T ) =

C × TCC × T 1 000 000

(28)

3418


where C is the nominal value of the capacitance Cm − C25 × 106 [ppm/°C] C25 × (T − 25) where T is the maximum design ambient temperature at which the capacitor can be continuously used, C25 is the capacitance value at 25 °C (reference temperature), Cm is the capacitance value at temperature T °C, and T is the change in temperature from the reference value. The expression of capacitance C x and Cs with ambient temperature is [C x + (C x × TCC × T /1 000 000)] and [Cs + (Cs × TCC × T /1 000 000)], respectively. Hence, the expression of frequency f in terms of ambient temperature is ⎡ η Rb ⎣ 1 − f = 2Ra 2R0 C0 2Rg C0 ⎞⎤ ⎛ C s ×(TCC)×T + − C C x + C x ×(TCC)×T s 1 000 000 1 000 000 ⎠⎦. ×⎝ Cf TCC =

(29) C. Error in Output Frequency Due to Input Offset Voltage in Amplifiers A1 and A2 Considering the input offset voltages V01 and V02 of the amplifiers A1 and A2, respectively, the relative frequency error is given by [32]–[34] V02 V01 V02 1 − 2Rg C 0 Vs Vs + 2R 0 C 0 Vs δf = . (30) 1 f + ηδ 2R 0 C 0

2R g C 0

Considering V01 = V02 = 5 mV (LF-351), Vs = 4 V, and δ = 0.86, δ f / f yields to be approximately 0.438 × 10−2 , which can be considered as negligible. D. Limiting Error in Output Frequency Due to Components Used in the Circuit The relative limiting error in X due to the tolerance of the components can be determined by the relation δ X / X = ±(δx 1 /x 1 + δx 2 /x 2 ) and δ X/ X = ±((x 1 / X)δx 1 /x 1 + (x 2 / X)δx 2 /x 2 ) for the product and sum of two components, where the errors in values x 1 and x 2 are ±δx 1 and ±δx 2 , respectively [23], [29]. While calculating the limiting error in output frequency f , the standard values of resistances and capacitances were used as Rb = 10 k, Ra = 1 k, R0 = 240 k, Rg = 36 k, Cs = 486 pF, C x = 400 pF, C f = 100 pF, and C0 = 1 nF, considering ±0.5% tolerance for the resistances and capacitances used. The limiting error in output frequency is calculated as f = 13402.81 Hz ± 2%. V. E XPERIMENTAL R ESULTS A. Response Characteristics of the Capacitive Moisture Sensor in ppm Level The capacitive sensor that works on dielectric adsorption principle is briefly discussed. The sensor has been fabricated by depositing thin of nanoporous layer of γ -Al2 O3 of ∼5-μm thickness deposited in between two gold electrodes using the

Fig. 4.

Variation of capacitance of the sensor with moisture.

sol–gel dip-coating method. The first gold electrode has been deposited on the alumina substrate by screen printing method and sintered at 900 °C for 1 h. Metal–oxide sensing film is deposited on the electrode by dip coating of AlO(OH) sol, and the film was sintered at a temperature of around 400 °C for 1 h. Then, another gold electrode is deposited on top of the sensing film. The size of the electrode on the top of the sensing film is 14 mm × 16 mm. Electrodes and metal–oxide sensing film are finally sintered at 900 °C for 1 h. Complete sensor fabrication and its characterization were reported in [16]. To determine the capacitive response curve, the sensor is placed in a sample chamber of ∼50-cc volume with output terminals connected to the impedance analyzer (4294 A, Agilent, Vac = 500 mV, and f = 1 kHz). The moisture level in the range of 0–110 ppm has been created by precision volume flow controller. Moisture concentration has been varied by mixing moist air with dry air (N2 ). Initially, the sensor is refreshed by dry N2 gas to a minimum moisture level of 5.2 ppm. SHAW (UK) dew point meter has been used to monitor the moisture in the chamber. The capacitance value with change in moisture measured by the impedance analyzer is shown in Fig. 4. Response curve shows that the initial offset capacitance at dry humidity (5.2 ppm) is nearly 486 pF, and the maximum change in capacitance for 107-ppm moisture change is approximately 30 pF. The capacitance change above 20 ppm is almost linear, and over the full moisture range, the nonlinearity is ∼0.96% and the sensitivity is 0.3 pF/ppm. Transient response curve for response and recovery times and the reproducibility of the sensor have been determined using the detection electronic circuit. A comparative literature for capacitive humidity sensor in ppm level is given in Table I [12]–[14]. B. Hardware Realization of the Circuit and Determination of the Response Characteristics of the Sensor The circuit was implemented using op-amp with high slew rate and low response time. Fast active device has been selected to minimize the frequency response error due to time delays of the device. The op-amp used is LF-351. The power supplies are ±12 V. The central frequency f 0 is designed at 11.4 kHz for the moisture sensor using the component values: C0 = 0.954 nF, R0 = 0.232 M, Ra = 0.978 k, and


3419

TABLE I C OMPARATIVE L ITERATURE OF ppm-L EVEL H UMIDITY S ENSOR

Rb = 9.89 k. Other relevant parameters used are Rg = 35 k and C f = 100 pF. For capacitive TRA bridge, an audio frequency center tap transformer with transformation ratio η = 1/20 has been selected. The secondary output of the transformer (±Vx ) is 1.18 V. Both primary and secondary windings are placed on high-permeability core. The secondary voltage source has small internal impedances so that C x1 and Cs1 can be neglected. The effect of other parasitic capacitance Ce is canceled by connecting the terminal P to an inverting current to voltage converter circuit. Thus, the capacitive TRA bridge circuit can measure the capacitance change with high degree of accuracy and resolution. The initial experiments have been conducted with discrete ceramic disc capacitances and variable gang capacitors. Cs is used as reference capacitance, and C x is used as unknown variable capacitance arm. The voltage output of the op-amp A1, Vcf , has been observed with a change in discrete capacitance for both Cs > C x and Cs < C x . Fig. 5 shows the voltage output with the incremental change in capacitance. It shows that the output linearly changes with the change in discrete capacitance as predicted in (3). The minor asymmetry in the response behavior is observed, but it is small as the slope of the curve for lower values of C x ≤ 110 pF is −0.0089 V/pF and for higher values of C x > 110 pF is 0.0082 V/pF. A handheld LC R meter of Agilent (model no U1733C) was used to measure the discrete capacitances at 1-kHz signal frequency whose accuracy is ±(0.5% + 5) for the 0–200-pF range, where accuracy is given as ± (% of reading + counts of least significant digit). The accuracy specification of the meter for the 0–20-pF range is not mentioned. This asymmetry may be due to more measuring error of the discrete capacitance with value less than 50 pF. To verify the minimum capacitance that can be converted into frequency, the circuit has been tested with lower discrete capacitance from 8 to 280 pF as shown in Fig. 5. However, the circuit was tested with ppm-level humidity sensor. The op-amp A1 output Vcf has also been noted by replacing the C x by capacitive humidity sensor. The reference capacitance Cs = 486 pF is selected corresponding to capacitance

Fig. 5.

Variation of Vcf with discrete capacitance.

Fig. 6.

Variation of Vcf with moisture (ppm).

under dry condition. Variation of Vcf with change in humidity in ppm concentration is shown in Fig. 6. It increases with the increase in moisture starting from zero value. This response curve is almost similar to the capacitance response with moisture as shown in Fig. 4. A dummy reference humidity sensor can be used in places of Cs to compensate for the initial offset capacitance in dry humidity and to minimize the ambient temperature of the actual sensor C x . In the interface electronic circuit, a ceramic discrete capacitance of suitable value has been used for Cs . The circuit has been tested with metal–oxide aluminum thin humidity sensor that has been fabricated by sol–gel dip-coating method. The fabricated sensor is thermally stable as it has a very small change in capacitance with variation of ambient temperature. The result of temperature error has been shown in [16]. Therefore, for testing the present sensor, no reference humidity sensor has been used. But for other types of capacitive sensors that are having temperature error, reference sensor can be used to minimize the temperature error. To verify the frequency output of the oscillator, Vcf has been applied as one of the inputs of the summing integrator. The change in frequency of the square-wave output is measured using digital oscilloscope (Agilent Technologies DSO1002A). First, the frequency response of the interface electronics is determined with the variation of discrete capacitance for C x . The component values of the circuit are redesigned for a reference frequency of nearly 3.95 kHz. The response curve with discrete capacitance values is shown in Fig. 7. The frequency output increases with the increase in capacitance as predicted. Experiments have been performed with capacitive humidity sensor. Fig. 8 shows the change in frequency of the output signal with variation in moisture concentration

3420

Fig. 7.


Variation of frequency with discrete capacitance.

Fig. 8.

Variation of frequency with moisture (ppm).

Fig. 9.

Repeatability of the sensor with a moisture change of 94 ppm.

from 5.2 to 107 ppm. The frequency sensitivity of the circuit for the full-scale moisture measurement is 10.94 Hz/ppm ( f /ppm), and the nonlinearity is 1.2%, which is close to the nonlinearity value obtained from the capacitive response shown in Fig. 4 using the impedance analyzer. Frequency change is significant even though the capacitance change of the sensor for the lower humidity is small. Further experiments have been conducted using the proposed circuit to determine some of the important characteristics of the sensor such as response time, recovery time, and reproducibility of the sensor. The results are shown in Fig. 9. Fig. 9 shows the transient frequency response curve of the sensor for four identical step changes of moisture from 13 to 107 ppm to determine the response, recovery time, and reproducibility. These times indicate the nature of the time variation in input frequency f (VRb ) when the input moisture in ppm exhibits a step change. The sensor is refreshed to 13 ppm moisture, and it is suddenly exposed to a maximum moisture of 107 ppm. The frequency output rises from 11.46 to 12.36 kHz and reaches the saturation value. When the sensor is refreshed by dry N2 gas, moisture molecules desorb from the porous film and reaches an initial

Fig. 10.

Moisture measurement using the prototype.

moisture of 13 ppm. Response time is the time taken by the prototype to reach the output from 10% to 90% of its saturation value. Recovery time is the time taken to reach 10% of its saturation for the step change in moisture to a minimum value. In practical situations, such an instantaneous step change of moisture rarely occurs, and thus we have applied a worst situation in determining the time response. The response and recovery times are 45 and 120 s, respectively. The results obtained are comparable with the values reported in [16]. The output of the sensor is highly reproducible. Since the output of the sensor is in pulse wave form, it is applied to double NOT gates, and then directly interfaced with microcontroller. For detection of oscillation frequency and subsequent signal processing, an 8-bit PIC16F877A microcontroller was used. The square-wave signal with variable frequency after suitable scaling is applied as one of the inputs to an 8-bit counter TMR0. The 8-bit counter counts every rising edge of the square wave for exactly 1 s. The number of counter overflows multiplied by 256 plus the number of counts in TMR0 will give the total number of counting pulses and, hence, the frequency. The counting unit can measure a minimum frequency of 1 Hz. The frequency of relaxation oscillator has been calibrated in terms of ppm using lookup table. In the lookup table, the counter output N and the corresponding moisture values in ppm are stored in the memory of the microcontroller. Following the measurement, the input value N is looked in the table and the correct moisture value is displayed on the liquid crystal display (LCD). The counter outputs are stored in the ascending order in the L memory locations, and the corresponding moisture values are stored in the following L locations. Thus, if N is found at the J th memory location from the start of the table, the actual moisture value is found in the (L+ J ) location. The reading displayed on the LCD is compared with the commercial dew point meter (SHAW, UK). The meter is having the measurement range of 0–1000 ppm and an accuracy of ±1 ppm. The accuracy of the moisture measurement is found to be nearly ±1%. The resolution of the frequency measurement of the interface circuit is 1 Hz. The comparison between the humidity as measured by the prototype with the commercial dew point meter is shown in Fig. 10. The maximum error is 2 ppm, and the nonlinearity is 0.03%. It may be possible to cause small frequency error due to nonideal behaviors of the op-amps, particularly the switching delay of the comparator IC. Selecting high slew rate device, such an error can be minimized [20], [32]–[34].


3421

VI. C ONCLUSION

Fig. 11.

Variation of frequency with parasitic capacitances.

Fig. 12.

Variation of frequency with change in ambient temperature.

The main sources of errors of a relaxation oscillator-based resistance to frequency converter circuit have been studied, and the compensation scheme is suggested in [34]. C. Simulation Results of Parasitic Earth Capacitances and Ambient Temperature Simulations are carried out in MATLAB to study the effect of parasitic earth capacitances and ambient temperature variations on output frequency using (27) and (29), respectively. The simulation results of frequency variation with parasitic capacitances are shown in Fig. 11. It is observed that the output frequency ( f ) is affected by the variation of parasitic capacitances when the value of the parasitic earth capacitances is above 10−7 F. However, the parasitic capacitance value is of the order of picofarads and such a large value rarely occurs in practice. The variation in output frequency with the change of the capacitances (Cs and C x ) due to change in ambient temperature from −55 °C to 105 °C is studied. The simulation work is carried with Cs = C x = 486 pF. The simulation results of output frequency variation considering the effect of ambient temperature on Cs and C x are shown in Fig. 12. The response curve shown in dots indicates the variation of frequency due to the temperature effects of only C x , while the response in triangle indicates the temperature effect for Cs . The response curve in diamond symbol indicates the temperature effect of both C x and Cs . It is observed that due to differential property of the TRA bridge, the change in output frequency with change in ambient temperature is almost independent. Small difference can be attributed due to the difference in temperature coefficient of Cs and C x .

In this paper, our aim is to develop a linear, sensitive, and simple capacitive bridge circuit for converting incremental capacitance change into frequency with high precision and accuracy. The output signal in digital form can be easily acquired by a microcontroller. A simple audio frequency transformer is used for interfacing the capacitive sensor. The proposed circuit is simple to operate and requires only few components for its hardware implementation. It is capable of measuring incremental capacitance change of different types of capacitive sensors for different applications including tracelevel moisture measurement. The experimental results confirm the theoretical value. The effectiveness of the circuit has been tested for a γ -Al2 O3 -based thin-film moisture (ppm) sensor. The simulation results show that the output of the circuit is stable due to parasitic capacitance and ambient temperature. For calibration and display, the frequency output of the circuit, which is a function of change in humidity, is interfaced with microcontroller. The resultant overall nonlinearity is also compensated for using the lookup table which is stored in microcontroller memory. However, because of the transformer, the circuit is not CMOS compatible, and for lossy capacitance measurement, conductance component should be considered. ACKNOWLEDGMENT The authors would like to thank the reviewers and editors for their valuable suggestions for improving this manuscript. Dr. T. Islam would also like to thank A. Hossain, his father, for his constant support and motivation in research. R EFERENCES [1] B. Hague and T. R. Foord, Ed., Alternating Current Bridge Method, 6th ed. London, U.K.: Sir Isac Pitman and Sons Ltd, 1971. [2] W.-C. Heerens, “Application of capacitance techniques in sensor design,” J. Phys. E, Sci. Instrum., vol. 19, no. 11, pp. 897–906, 1986. [3] A. A. A. Aish and M. Rehman, “Development of a capacitive mass measuring system,” Sens. Actuators A, Phys., vol. 151, no. 2, pp. 113–117, 2009. [4] S. C. Bera, H. Mandal, S. Saha, and A. Dutta, “Study of a modified capacitance-type level transducer for any type of liquid,” IEEE Trans. Instrum. Meas., vol. 63, no. 3, pp. 641–649, Mar. 2014. [5] S. Avramov-Zamurovic and R. D. Lee, “A high-stability capacitance sensor system and its evaluation,” IEEE Trans. Instrum. Meas., vol. 58, no. 4, pp. 955–961, Apr. 2009. [6] X. Wang, Y. Wang, H. Leung, S. C. Mukhopadhyay, M. Tian, and J. Zhou, “Mechanism and experiment of planar electrode sensors in water pollutant measurement,” IEEE Trans. Instrum. Meas., vol. 64, no. 2, pp. 516–523, Feb. 2015. [7] S. C. Mukhopadhyay, C. P. Gooneratne, G. S. Gupta, and S. N. Demidenko, “A low-cost sensing system for quality monitoring of dairy products,” IEEE Trans. Instrum. Meas., vol. 55, no. 4, pp. 1331–1338, Aug. 2006. [8] A. I. Zia et al., “Technique for rapid detection of phthalates in water and beverages,” J. Food Eng., vol. 116, no. 2, pp. 515–523, 2013. [9] C. V. K. Kandala, C. L. Butts, and S. O. Nelson, “Capacitance sensor for nondestructive measurement of moisture content in nuts and grain,” IEEE Trans. Instrum. Meas., vol. 56, no. 5, pp. 1809–1813, Oct. 2007. [10] W.-H. Zhou, L.-C. Wang, and L.-B. Wang, “A method to measure the dynamic characteristics of microhumidity sensors,” IEEE Trans. Instrum. Meas., vol. 63, no. 12, pp. 2993–3001, Dec. 2014. [11] Z. M. Rittersma, “Recent achievements in miniaturised humidity sensors—A review of transduction techniques,” Sens. Actuators A, Phys., vol. 96, nos. 2–3, pp. 196–210, 2002.

3422


[12] T. Islam, S. S. Islam, and H. Saha, “Porous silicon based moisture detector in the ppmV range,” Sensor Lett., vol. 6, no. 5, pp. 746–751, Oct. 2008. [13] F. Hossein-Babaei and P. Shabani, “A gold/organic semiconductor diode for ppm-level humidity sensing,” Sens. Actuators B, Chem., vol. 205, pp. 143–150, Dec. 2014. [14] X. Shi, Q. Chen, J. Fang, K. Varahramyan, H.-F. Ji, “Al2 O3 coated microcantilevers for detection of moisture at ppm level,” Sens. Actuators B, Chem., vol. 129, no. 1, pp. 241–245, 2008. [15] G. K. McMillan and D. M. Considine, “Humidity and moisture system,” in Process/Industrial Instruments and Controls Handbook, 5th ed. New Delhi, India: McGraw-Hill, pp. 4.194–4.247. [16] T. Islam, L. Kumar, and S. A. Khan, “A novel sol–gel thin film porous alumina based capacitive sensor for measuring trace moisture in the range of 2.5–25 ppm,” Sens. Actuators B, Chem., vol. 173, pp. 377–384, Oct. 2012. [17] N. Karlsson, “A study of a high-resolution linear circuit for capacitive sensors,” IEEE Trans. Instrum. Meas., vol. 48, no. 6, pp. 1122–1124, Dec. 1999. [18] T. Islam and H. Saha, “Study of long-term drift of a porous silicon humidity sensor and its compensation using ANN technique,” Sens. Actuators A, Phys., vol. 133, no. 2, pp. 472–479, 2006. [19] N. Nizza, M. Dei, F. Butti, and P. Bruschi, “A low-power interface for capacitive sensors with PWM output and intrinsic low pass characteristic,” IEEE Trans. Circuits Syst. I, Reg. Papers, vol. 60, no. 6, pp. 1419–1431, Jun. 2013. [20] H. Shubata, M. Ito, M. Asakursa, and K. Watanabe, “A digital hygrometer using a polyimide film relative humidity sensor,” IEEE Trans. Instrum. Meas., vol. 45, no. 2, pp. 564–569, Apr. 1996. [21] S. Middelhoek, P. J. French, J. H. Huijsing, and W. J. Lian, “Sensors with digital or frequency output,” Sens. Actuators, vol. 15, no. 2, pp. 119–133, 1988. [22] K. Mochizuki, K. Watanabe, T. Masuda, and M. Katsura, “A relaxationoscillator-based interface for high-accuracy ratiometric signal processing of differential-capacitance transducers,” IEEE Trans. Instrum. Meas., vol. 47, no. 1, pp. 11–15, Feb. 1998. [23] E. O. Doebelin, Measurement Systems Application and Design, 5th ed. New York, NY, USA: McGraw-Hill, 2003. [24] C. M. G. Meijer, Ed., Smart Sensor Systems. New York, NY, USA: Wiley, 2008. [25] A. De Marcellis and G. Ferri, Analog Circuits and Systems for VoltageMode and Current-Mode Sensor Interfacing Applications. Berlin, Germany: Springer-Verlag, 2011. [26] T. Islam, L. Kumar, Z. Uddin, and A. Ganguly, “Relaxation oscillatorbased active bridge circuit for linearly converting resistance to frequency of resistive sensor,” IEEE Sensors J., vol. 13, no. 5, pp. 1507–1513, May 2013. [27] S. M. Huang, A. L. Stott, R. G. Green, and M. S. Beck, “Electronic transducers for industrial measurement of low value capacitances,” J. Phys. E, Sci. Instrum., vol. 21, no. 3, pp. 242–250, 1988. [28] E. Traversa, “Ceramic sensors for humidity detection: The state-ofthe-art and future developments,” Sens. Actuators B, Chem., vol. 23, nos. 2–3, pp. 135–156, 1995. [29] E. W. Golding and F. C. Widdis, “Capacitors, capacitance, and dielectric,” Electrical Measurements and Measuring Instruments, 5th ed. New Delhi, India: Reem Pub., 2009, pp. 135–179. [30] M. Gasulla, X. Li, and G. C. M. Meijer, “The noise performance of a high-speed capacitive-sensor interface based on a relaxation oscillator and a fast counter,” IEEE Trans. Instrum. Meas., vol. 54, no. 5, pp. 1934–1940, Oct. 2005. [31] A. S. Sedra and K. C. Smith, Microelectronics Circuits, 5th ed. London, U.K.: Oxford Univ. Press, 2004. [32] N. M. Mohan and V. J. Kumar, “Novel signal conditioning circuit for push-pull type capacitive transducers,” IEEE Trans. Instrum. Meas., vol. 56, no. 1, pp. 153–157, Feb. 2007. [33] S. N. Nihtianov, G. P. Shterev, B. Iliev, and G. C. M. Meijer, “An interface circuit for R-C impedance sensors with a relaxation oscillator,” IEEE Trans. Instrum. Meas., vol. 50, no. 6, pp. 1563–1567, Dec. 2001. [34] T. Islam, A. U. Khan, and J. Akhtar, “Accuracy analysis of oscillator-based active bridge circuit for linearly converting resistance to frequency,” in Proc. Int. Conf. Multimedia, Signal Process. Commun. Technol. (IMPACT), Aligarh, India, Oct. 2013, pp. 305–309.

Tarikul Islam received the M.Sc.Eng. degree in instrumentation and control systems from Aligarh Muslim University, Aligarh, India, in 1997, and the Ph.D. degree from the Department of Electronics and Telecommunication Engineering, Jadavpur University, Kolkata, India, in 2007. He is currently a Professor with the Department of Electrical Engineering, Jamia Millia Islamia University, New Delhi, India. He has authored or coauthored over 100 papers in peer-reviewed journals, conferences and book chapters. His current research interests include thin film sensor, electronic instrumentation, and soft computing techniques for signal conditioning.

Shakeb A. Khan received the M.Sc.Eng. degree in instrumentation and control from Aligarh Muslim University, Aligarh, India, in 1994, and the Ph.D. degree in instrumentation from IIT Delhi, New Delhi, India, in 2005. He is currently a Professor with the Department of Electrical Engineering, Jamia Millia Islamia University, New Delhi. His current research interests include electronics instrumentation, and soft computing techniques for signal conditioning techniques.

Md. Firoz A. Khan is currently a Ph.D. Research Scholar with the Department of Electrical Engineering, Jamia Millia Islamia University, New Delhi, India.

Subhas Chandra Mukhopadhyay (M’97– SM’02–F’11) received the bachelor’s (Hons.) degree in electrical engineering from the Department of Electrical Engineering, Jadavpur University, Kolkata, India, the master’s degree in electrical engineering from the Indian Institute of Science, Bangalore, India, the Ph.D. (Eng.) degree from Jadavpur University, and the D.Eng. degree from Kanazawa University, Kanazawa, Japan. He is currently a Professor of Sensing Technology with the School of Engineering and Advanced Technology, Massey University, Palmerston North, New Zealand. He has over 25 years of teaching and research experiences. He has authored or co-authored three books and over 300 papers in different international journals, conferences, and book chapters. He has edited 12 conference proceedings. He has also edited 11 special issues of international journals as a lead Guest Editor, and 19 books out of which 17 are with Springer-Verlag. His current research interests include sensors and sensing technology, instrumentation, wireless sensor networks, electromagnetics, control, electrical machines, and numerical field calculation. Dr. Mukhopadhyay is a fellow of the Institution of Engineering and Technology in U.K., and the Institution of Electronics and Telecommunication Engineers in India. He is a Topical Editor of the IEEE S ENSORS J OURNAL, an Associate Editor of the IEEE T RANSACTIONS ON I NSTRUMENTATION AND M EASUREMENTS , and a Technical Editor of the IEEE T RANSACTIONS ON M ECHATRONICS . He is the Co-Editor-in-Chief of the International Journal on Smart Sensing and Intelligent Systems.