A Simplified Control Algorithm for Shunt Active Power ... - IEEE Xplore

A Simplified Control Algorithm for Shunt Active Power Filter Without Load and Filter Current Measurement Engin Ozdemir

Mehmet Ucar

Metin Kesler

Murat Kale

Kocaeli University Technical Education Faculty Electrical Education Department Umuttepe Kocaeli TURKEY [email protected]

Kocaeli University Technical Education Faculty Electrical Education Department Umuttepe Kocaeli TURKEY [email protected]

Kocaeli University Technical Education Faculty Electrical Education Department Umuttepe Kocaeli TURKEY [email protected]

Kocaeli University Technical Education Faculty Electrical Education Department Umuttepe Kocaeli TURKEY [email protected]

Abstract – In this study, a new control algorithm for three-phase shunt Active Power Filter (APF) is proposed without load and filter current measurement. The proposed control method, based on Instantaneous Reactive Power (IRP) theory, requires only measuring the source currents in order to reduce the number of Current Sensors (CSs) required in the conventional control approach. The source currents are exactly in phase with the line voltages and approximately sinusoidal waveform after compensation. The proposed control technique has been simulated Matlab/Simulink and tested under thyristor bridge rectifier load condition, and validated with a 10kVA/380V experimental prototype based on Digital Signal Processor (DSP) TMS320F2812. Both simulation and experimental test results demonstrate viability of the proposed method, successful in meeting the IEEE 519-1992 recommended harmonic standard limits.

I. INTRODUCTION

In the conventional p-q theory based control approach for the shunt APF, the compensation current references are generated based on the measurement of load currents as shown in Fig. 3. However, the current feedback from the shunt APF output is also required and therefore, minimum four CSs are desired in a balanced system. In the proposed control method; the number of required CSs is minimized as shown in Fig. 3 and 4. In addition, the reference current calculation algorithm are simplified and easily implemented in the experimental prototype. In the proposed control algorithm, sensing only two-phase voltages, two source currents and a DC-link voltage is adequate to compute reference currents of the three phase shunt APF. In this way, the overall system design becomes easier to accomplish and the total implementation cost is reduced.

Harmonic current pollution of three-phase electrical power systems is becoming a serious problem due to the wide use of non-linear loads, such as diode or thyristor rectifiers and a vast variety of power electronics based appliances. Traditionally, passive LC filters have been used to eliminate the current harmonics and to improve the power factor. However, passive LC filters are bulky, load dependent and inflexible. They can also cause resonance problems to the system. In order to solve these problems, APFs have been reported [1-4] and considered as a possible solution for reducing current harmonics and improving the power factor. Fig. 1 shows the basic compensation principle of the three phase shunt APF. It is designed to be connected in parallel with the non-linear load to detect its harmonic and reactive current and to inject into the system a compensating current. Therefore, the current drawn from the power system at the coupling point of the shunt APF will result sinusoidal as shown in Fig. 2 and calculated by iS=iL+iC .

iL

iS

3∼

vS

ZLoad RS LS

AC Source

iC

RL

LL Non-linear Load

CDC

RC LC

VDC

Active Power Filter Fig. 1. The basic compensation principle of the shunt APF

iL (A)

iC (A)

(1)

In recent years, some methods based on the IRP theory [5-7], also known as p-q theory, have been used to obtain the current reference for the APFs. Although several improved algorithms are proposed [8-10], most control circuits are complicated and not easy to implement.

iS (A) t(s) Fig. 2. Non-linear load, compensation and source current waveforms

1-4244-0136-4/06/$20.00 '2006 IEEE

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vS

iSa

RS LS

iLa

CS

RL LL

p-q transform. vSa

Non-linear load

0

AC Source

vSb

iCa

iSa

PT

iSb

α-β Transf. (Eq. 6)

Ref. current calc.

vα vβ

iα α-β Transf. iβ (Eq. 7)

p

Instant. Power Calcul.

LPF

(Eq. 9)

0

p

q

QAH

α-β i∗Sα Current ∗ Referen. iSβ

α-β i ∗Sa Inverse ∗ Transf. i Sb

(Eq. 11)

(Eq. 12) -

VDC

CS

* VDC

RC

6

VDC

Fig. 3. Schematic diagram of the conventional shunt APF control method vS

RS LS

CS

iSa

iLa

RL LL

Non-linear load

0

AC Sources

PI

QBH QBL QCH QCL

iSc

DC voltage regulator

-

Σ

-

In this paper, p-q theory is employed as a suitable method to the analysis of non-linear three-phase load and for control of the shunt APF. The p-q theory [5] is based on the α-β-0 transformation, also known as the Clarke Transformation. It consists in a real matrix to transform measured three-phase source voltages (vSa, vSb, vSc) and source currents (iSa, iSb, iSc) into the α-β-0 stationary reference frame as shown in (2) and (3).

Active Power Filter

CDC

Σ

QAL

i ∗Sc

ploss

+

Σ

Fig. 5. Control block diagram of the proposed method

LC

DSP Control Unit

-

Hysteresis Band Current Controller

iCa

v0    vα  = v   β

1/ 2 1/ 2 1/ 2  vSa    2  - 1/2 - 1/2  vSb   1 3  0 3 /2 - 3 /2 vSc   

(2)

i0    iα  = i   β

1/ 2 1/ 2 1/ 2  iSa    2  - 1/2 - 1/2  iSb   1 3  0 3/2 - 3 /2 iSc   

(3)

PT

RC LC

DSP Control Unit

6

In this study, the shunt APF is connected to a three-phase three-wire system, which has sinusoidal and balanced source voltages and load condition. Thus, the sums of the instantaneous source voltages and source currents of three phases are zero. Measuring of two source voltages and currents are adequate for the reference current calculations. Therefore, the Clarke Transformation of the source voltages and currents are calculated as shown in (4) and (5) [11].

Active Power Filter

CDC

VDC

Fig. 4. Schematic diagram of the proposed shunt APF control method

The waveform of the source current based on the proposed control method can be approximated as a sinusoidal waveform after compensation. Simulations and experimental tests are performed in a laboratory prototype to confirm the validity of the proposed control algorithm. The proposed method is found quite satisfactory to compensate the reactive power and current harmonics of the nonlinear load. II. THE PROPOSED CONTROL METHOD The control block diagram of the proposed method for the three phase shunt APF is shown in Fig. 5. It consists of p-q transformation, a Low Pass Filter (LPF), one PI controller for dc-link voltage regulator, reference current calculation and hysteresis-band current controller. From Fig. 5, it can be realized that only source voltage and current signals are utilized in the control circuit of the proposed shunt APF to calculate the reference currents of the Voltage Source Inverter (VSI).

 vα   3/2  = v  β  1/ 2

0   2 

vSa    vSb 

(4)

iα   3/2  = i  β  1/ 2

0   2 

iSa    iSb 

(5)

In this paper, in order to perform easily implementation of the α-β transformations in the DSP software, (4) and (5)

2 / 3 and the simplified Clarke are multiplied with Transformations are obtained as in (6) and (7).

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 vα   1  = v  β  1/ 3

0   2/ 3 

vSa    vSb 

(6)

iα   1  = i  β  1/ 3

0  iSa    2/ 3  iSb 

The ‘c’ phase current reference is calculated by

(7)

i ∗Sc

In the p-q theory, instantaneous real power (p) and instantaneous imaginary power (q) are calculated as  p   vα   = − v  q   β

v β  iα    . v α  i β   

(8)

(9)

The instantaneous real power, including ac and dc values, can be expressed as p= p+ ~ p.

(10)

In the proposed control algorithm, the instantaneous imaginary power is set to zero (q=0) and a single dc component of the real power ( p ) is therefore selected as compensation power reference for harmonic and reactive power compensation. The compensation current references in α-β coordinates are calculated by (11). In order to produce dc component of the instantaneous real power ( p ), instantaneous active power (p) is filtered by using 1st order LPF with a cut-off frequency at 50 Hz. The average real power ( ploss ) is added to the dc component of the instantaneous real power ( p ) to cover the VSI losses of the shunt APF. i∗  1  Sα  = ∗  2 2 iSβ  vα + v β

 vα  v β

-v β   p + ploss    vα   0 

(11)

In this study, the compensation current references in the α-β coordinates are then transformed back into the a-b-c coordinates through the inverse simplified Clarke Transformation as given by i∗   1  Sa  =  i∗  - 1/2  Sb 

0   3 /2

i∗   Cα  . ∗  iCβ 

.

(13)

These reference currents should be supplied to the power system by switching of the IGBT. The method for generation of the switching pattern is achieved by the instantaneous current control of the shunt APF line currents. The actual source currents ( iSa , iSb ) are measured instantaneously and actual ‘c’ phase current ( iSc ) is obtained as equation (13) and

In the conventional p-q theory based control algorithm with load current measurement; the instantaneous imaginary power (q) is calculated and reverse of it is used in control algorithm for reactive power compensation. In the proposed method, calculation of the ‘q’ is not required with the proposed method, since it is not desired to draw the ‘q’ from the source. Only calculation of the instantaneous real power (p) is adequate as shown in (9). Therefore, the proposed control scheme has been simplified in the calculations of reference currents in DSP software. p = v α iα + v β i β

∗ ∗ = − (i Sa + i Sb )

(12)

∗ , ∗ , ∗ ) then compared to the reference currents ( iSa i Sb i Sc generated by the control algorithm in the hysteresis-band current controller. Three hysteresis-band current controllers generate the switching pattern of the VSI. The switching logic is formulated as follows: ∗ _ HB) higher switch is OFF and lower switch If iSa < ( i Sa is ON for leg “A” (QA=1). ∗ + HB) higher switch is ON and lower switch If iSa > ( i Sa is OFF for leg “A” (QA = 0).

The switching functions QB and QC for legs ‘B’ and ‘C’ are determined similarly, using corresponding reference and measured currents and hysteresis bandwidth (HB). The hysteresis-band current control is the fastest control method with minimum hardware and software but variable switching frequency is its main drawback [12]. The additional average real power ( ploss ), to cover the VSI losses, is obtained from dc-link voltage regulator. Dc-link voltage regulator is designed to give both good compensation and an excellent transient response. The actual dc-link capacitor voltage is compared by a reference value and the error is processed in a PI controller which is employed for the voltage control loop since it acts in order to zero the steady-state error of the dclink voltage. III. SIMULATION AND EXPERIMENTAL RESULTS A. Simulation Results

The performance of the proposed control algorithm was first tested by simulation of a typical three-phase system with non-linear load by Matlab/Simulink power system toolbox. In this study, the load considered is a three-phase halfcontrolled thyristor bridge rectifier feeding an R-L load. Fig. 6 and Fig. 7 show the simulation block diagram and simulation results of the proposed control method including source voltage and current respectively. It can be shown in Fig. 7 that source voltages and currents are balanced and sinusoidal after compensation. The results obtained have clearly shown that, even using only source current measurement, the shunt APF performance is quite similar to that of conventional solutions.

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N

A B C

A B C

3-phase source

A B C

A B C vabc iabc

v-i measure

A B C vabc iabc

v-i measure RC LC

valfa vabc vbeta

A B C

p

A B C

thyristor-bridge rectifier

Rdc

valfa vbeta icalfa*

icalfa* iabc*

p

icbeta*

icbeta*

abc-αβ trans. αβ cur. ref. calc. αβ-abc trans. p calc. dc voltage reg.

iabc* pulses iabc

hyst.-band cur. control

Fig. 5

A

+

B

+ v -

C

C pulses

Discrete system Ts=1e-006

vabc pulses alpha_deg

Ldc

Vdc

iabc vabc

A B C

A + B C pulses -

step pulse generator

A B C

iabc

v-i measure

A A B B C C RL LL

-

3-Leg VSI Block Fig. 6. The simulation block diagram

Moreover, as the proposed circuit requires only source current feedback signals where the load CSs can be saved, the circuit design can be thus simplified significantly. In order to get transient response of the proposed algorithm; firing angle of the three-phase thyristor bridge rectifier is increased to 45o from 15o. The source current waveforms reveal that the shunt APF is effective in compensating for load harmonic current and reducing the Total Harmonic Distortion (THD) of source current from 24.5% to about 3.4% and from 40.8% to about 4.7% after load has changed in 0.3 second as shown in Fig. 7. 400

vSa (V) 200 0

iSa (A) -200

-400 0.24 400 vSa (V) 200 0 iLa (A) -200 -400 0.24 40 20 iCa (A) 0 -20 -40 0.24 850

5x iSa

vSac

0.26

0.28

0.3

0.32

0.34

0.36

0.38

0.26

0.28

0.3

0.32

0.34

0.36

0.38

0.26

0.28

0.3

0.32

0.34

0.36

0.38

0.26

0.28

0.3

0.32

0.34

0.36

0.38

5x iLa

vSac

Vdc (V) 800 750 0.24

time (s)

Fig. 7. The simulation results of the proposed method including source voltage and current

B. Experimental Results

The feasibility of hardware implementation for the proposed control algorithm was evaluated by design and implementation of the-phase three-wire shunt APF. A threephase thyristor bridge rectifier with the R-L load is connected to ac mains to demonstrate the effectiveness of the shunt APF with the proposed method. Fig. 8 shows the configuration of the 10 kVA experimental prototype. The current control of the shunt APF is implemented by hysteresis-band current controller on the DSP. The proposed control technique is implemented on the TMS320F2812 DSP using C programming language. It was verified by implementing an algorithm programmed by C language using floating-point arithmetic (IQ math) in the DSP. TMS320F2812 eZdsp has an internal 36 kB RAM, 2 kB OTP ROM, 256 kB flash, 16 channels PWM, 12-bit 16 channels ADC and an external 128 kB SRAM, expansion interfaces and parallel port JTAG interface. It can perform parallel multiply and ALU on integer or floating point data (using IQmath floating point engine) in a 6.67 ns single cycle instruction time with a peak computation rate of 150 MIPS. Fig. 9 and Fig. 10 show source voltage and current waveforms before and after filtering respectively. It can be shown an improvement in the aspect of the waveform of source current when compared with the waveform without filtering. After compensation, source current becomes sinusoidal and in phase with the source voltage; hence, both harmonics and reactive power are compensated simultaneously. Before harmonic compensation the THD of the supply current was 27.5% and after the harmonic compensation, it was reduced to 4.75% which complies with the IEEE 519 harmonic standards. The design specifications and the main parameters of the prototype are indicated in Table I. The photograph of the shunt APF prototype is shown in Fig. 11.

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vS

RS LS

iSa

RL LL

iLa

RDC

CS 0

LDC PT

iCa

AC Source 380V, 50Hz

Thyristor bridge rectifier Non-linear load

RC

vSa vSb iSa iSb

LC

CS- Hall effect Current Sensor PT-Potential Transformer

Shunt Active Power Filter

Pre-Charging Resistors

A CDC

B C Semiconductor Protection Fuses

IGBT Drive Board (gate drive-isolation-short circuit and overcurrent protection)

Source Currents

(iSa, iSb)

Source Voltages (vSa, vSb)

Current-Voltage Measurement Interface Boards

Isolation Amplifier

VDC

DC Voltage Measurement Interface Board (overvoltage protection )

DSP Control Board TMS320F2812 eZdsp

Parallel Port Cable

PC TI C2000 Code Composer Studio IDE (compile, link and assemble)

Fig. 8. The shunt APF configuration of the experimental prototype

iSa

vSa

vSa

iSa

Fig. 10. Source voltage and current waveform after filtering (vSa=100V/div, iSa=20A/div)

Fig. 9. Source voltage and current waveform before filtering (vSa=100V/div, iSa=20A/div)

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TABLE 1 MAIN PARAMETERS OF THE PROTOTYPE

The results obtained have clearly shown that, even using only source current measurement, the APF performance is quite similar to that of standard solutions. Moreover, as the proposed circuit needs only supply current feedback signals where the load CSs can be saved, the circuit design can be thus simplified significantly.

System Parameters Source Voltage Source Frequency

VSabc fS

220Vrms/Ph.-neut. 50Hz

APF Parameters Dc Bus Voltage Dc Side Capacitance Ac Side Inductance Switching Frequency

VDC CDC LC fS

V. ACKNOWLEDGMENT 800V 1100µF 3.75mH 10kHz

This research was supported by TUBITAK Research Fund, (No: 105E182-HD-08) and Kocaeli University Research Fund, (No: 2001/13). VI. REFERENCES

Load Parameters Ac Side Inductance Dc Side Resistance Dc Side Inductance

LLA RLD LLD

1mH 20Ω, 145mH

[1]

[2] [3] [4] [5] [6] [7]

[8] Fig. 11. The photograph of the shunt APF prototype

[9]

IV. CONCLUSION This paper proposes a new p-q theory based control method for the shunt APF with only source current detection. With the proposed technique detection of neither load current nor APF output current is not necessary. Therefore, computations and circuit implementation of the control system become quite simple compared to the conventional load current detection algorithms. In order to confirm the effectiveness of the proposed control algorithm, the approach has been tested through the simulation and experimental validation. Experimental test results using a DSP TMS320F2812 are given to demonstrate the performance of the proposed method. Both simulations and experimental results confirm that the proposed control method presented in this paper has some advantages simple and easy to implement.

[10]

[11] [12]

[13]

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