Design and Layout Implementa nd Layout ...

Design and Layout Implementation of Microstrip Balanced Diode Filter Salman M. Khan, Jamshed Iqbal and Faisal Masood llah

Abstract—The paper describes the design and simulation of a balanced mixer using a 180º hybrid ring to combine Radio Frequency (RF) input and Local Oscillator (LO) and a pair of non-linear elements. The centre RF and Intermediate Frequency (IF) have been set at 10 GHz and 400 MHz respectively and there is no band inversion. The 180º hybrid is implemented as a 90º hybrid with an extra quarter wavelength transmission line. The Schottky diodes act as non-linear elements. The design has all elements needed to have the diodes correctly biased in DC. Computer Aided Design (CAD) based softwares have been used to design and simulate the mixer. Prior to design, the performance has been evaluated. The conversion loss and the optimum LO power to minimize the loss has been determined. The bandwidth of the mixer at -3dB has been computed and simulation results provide plots of conversion losses as a function of frequencies. Extension of this plot shows the image frequency rejection. Computation of LORF isolation as well as the LO leakage to the IF port have also been carried out. Other necessary performance parameters e.g. noise figure, noise contribution by each element and 3rd order intercept point etc have been determined. Following the performance evaluation of the proposed mixer, finally a Printed Circuit Board (PCB) layout has been designed. Thus the present work completely encompasses related issues of microstrip balanced diode filter design.

I. INTRODUCTION

Two bands are theoretically obtained: sum and difference. For down converters, the wanted signal is the difference. Intermediate frequency (IF) is the difference between signal frequency fRF and LO frequency fLO. Image frequency produces the same IF as the input signal. A double-balanced diode mixer normally make use of four diodes in a ring or star configuration with both the LO and RF being balanced. All ports of the mixer are inherently isolated from each other. The advantages of a double-balanced design over a single balanced design are increased linearity, improved suppression of spurious products (all even order products of the LO and/or the RF are suppressed) ant the inherent isolation between all ports. The disadvantage is that they require a higher level LO drive. A mixer uses the non-lineal characteristics of the diode to generate an output consisting on the sum and the rest between two frequencies (RF and OL). After the diodes, there is a low pass filter so as to discard RF + OL frequency. The general scheme of the design is shown in Figure 1.

Fig. 1. Block diagram of a low pass filter

Mixers are frequency translation devices. They allow the conversion of signals between a high frequency (the RF frequency) and a lower Intermediate Frequency (IF) or baseband. In communications systems the RF is the transmission frequency, which is converted to an IF to allow improved selectivity (filtering) and an easier implementation of low noise and high gain amplification. This paper details the design of mixer circuits, concentrating on low cost Printed Circuit Board (PCB) based designs using discrete Surface Mount Technology (SMT) components. Mixers are used to shift the frequency band of a given signal to a different center frequency. It is accomplished by multiplying the input signal with a tone.

The hybrid coupler is usually implement with a 180degree hybrid but in this case is used a 90-degree hybrid because in the layout the output ports are nearer than in the 180-degree one. The used diodes will be a pair that offers the ADS library. They are two identical diodes, which are polarized through a network of transmission lines. In this case, the mixer is analyzed with a harmonic balance simulation that will allow us calculate the inter-modulation products. This simulation needs the HB controller (Simulation-HB library). The basic needed values are the frequency of the inputs, the harmonics of each frequency and the maximum order of the inter-modulation products. Table I shows the mixer specifications. TABLE I MIXER SPECIFICATIONS Value Parameter 10 GHz Input Frequency 400 MHz Output Frequency Microstrip 90º hybrid plus Hybrid 90º TL Schottky Diodes Band inversion No

(1) Manuscript received August 29, 2010. S. M. Khan is with the School of Information Technology & Engineering (SITE), Ottawa Carleton Institute for Electrical and Computer [email protected]) Engineering, Canada (email: [email protected] J. Iqbal is with the ADVanced Robotics (ADVR) Department, Italian Institute of Technology (IIT) and University of Genova, Italy (phone: [email protected] 010-71781481; fax:+39-010-720321; e-mail: [email protected]). F. Masood is a Senior Design Engineer in Powersoft19 Inc. (e-mail: [email protected]).

275

With these specifications, the first step in the design phase was to select the fundamental components in a mixer. Block diagram of the mixer is shown in Figure 1. The approach for a diode passive mixer employs a 180º hybrid mixer to combine the LO with RF to feed the two signals using the non linear element which is a pair of Schottky diodes HSMS8202 suitable for frequency range of 10GHz. Using a 180º hybrid (based on 90º hybrid plus a 90º phase shift TL), we kept the LO with 180º phase shift in the two outputs of the hybrid and fed together into the diodes added in counter phase. This technique will automatically isolate the LO and the IF output. Furthermore, a coupled line RF input Band Pass Filter (BPF) for image rejection purposes and a step impedance Low Pass Filter (LPF) for the IF output has been incorporated into the mixer chain. This approach keeps microstrip Printed Circuit Board (PCB) substrate implementation simple. The transmission lines have been designed taking into consideration the implementations limitations in standard microstrip boards.

Fig. 3. I-V characteristics of Shottky diode

If the diode is excited with a sum of two signals, I = a1·(cos(w1·t ) + cos(w2 ·t )) + a2 ·(cos(w1·t ) + cos(w2·t ))2 + ...(3)

Expanding the formula and applying the trigonometric relation,

2·cos(w1t )·cos(w2t ) = cos((w1 − w2 )t ) + cos((w1 + w2 )t ).......(4) The interesting product of a mixer is the first order product; which gives addition and subtraction of inputs. The higher frequencies components at 2·RF and 2·LO have been obtained and needed to be filtered out. The mixer presented works with the subtraction of LO and RF. Because of the requirement of having no band inversion, the LO has to be at frequency of 9.6 GHz to give a 400 MHz output. IF = RF − L.O IF = 10Ghz − 9.6Ghz = 400 Mhz RF _ Im = RF − 2· IF = 10Ghz − 800 Mhz = 9.2Ghz.....(5)

Fig. 2. Block diagram of the mixer circuit

The rest of the paper is organized as follows: In Section II, we will introduce the concept from theoretical point of view. Section III contains the design details and implementation results. The performance evaluation and simulation results have been presented in Section IV. Section V presents the layout implementation of the proposed design. Finally, Section VI discusses the conclusion.

Where RF_im is the image frequency at the input that will be converted to the same IF frequency at the output. RF_Im must be filtered. The diode mixers have usual loses of 5 to 8 dB. The basic schematic of a single diode mixer is shown in Figure 4.

Fig. 4. Schematic diagram of a diode mixer

II.

THEORETICAL ANALYSIS

Mixers are frequency translation devices thereby permitting the conversion of signals between a high frequency (the RF frequency) and an intermediate frequency, usually lower than the RF. Most of the mixer circuits are based on the non linear response of a component, which can be a diode or a transistor. Diode is used as the non-linear element. Schottky diode offers better solution because of having a low forward voltage drop and a very fast switching action. The Schottky diode can be modeled using (1) while its IV characteristics have been shown in Figure 2. I = a 1 ·V + a 2 ·V

2

+ a 3 ·V

3

The LO in a diode mixer has to reach signals of about 8 dBm of power, so it is a good idea to cancel the output from the IF port. A technique to do this, used in proposed microstrip as well is to design a phase shift network (Fig. 5). The 180º hybrid ring makes a 180º phase shift only in the LO section, thus, when added together in the diode, the two waveforms will be cancelled with each other. An attenuation of 40 dB has been achieved in our simulation using this technique.

+ ...( 2 )

276

B. 180º Hybrid Design The The next next subsystem subsystem is is the the hybrid hybrid ring that permits feeding RF signal signal and LO signals to the diode the sum of filtered RF keeping the the isolation isolation and and port port match match at at correct correct pair thereby keeping levels. The ring has been implemented and simulated in a levels. The ring has been implemented and to 12 12 show show the corresponding separate circuit. Figures 9 to results. MTEE_AD S Tee1 Subst="MSub1" W 1=W _50 W 2=W _35 W 3=W _50

Fig. 5. Block diagram for Phase shift network

III. CIRCUIT DESIGN

The circuit implementation is done in the HP Advanced Design System Computer Aided Design (CAD) program, using the Microstrip Tl library and the component library for the diode model. Substrate parameters are based on a commercial microwave substrate, obtained directly from the datasheet and placed in the MSUB ADS section. Following the block diagram presented in Section I, the complete mixer implementation has been divided in several subsystems.

MLIN TL5 Subst="MSub1" W =W _50 L=L_50

MLIN TL8 Subst="MSub1" W =30.839488 m il L=864.889764 mil

Term Term 3 N um=3 Z =50 Ohm

MTEE_AD S Tee4 Subst="MSub1" W 1=W _35 W 2=W _50 W 3=W _50

MLIN TL6 Subst="MSub1" W =W _50 L=L_50

MLIN MLIN TL2 Subst="MSub1" Subst="MSub1" W =W _35 W =W _35 L=L_35 L=L_35


Term Term 4 N um=4 Z=50 Ohm


Va r Eqn

Goal OptimGoal1 Expr="dif _coupling" SimInstanceN ame="SP1" Min=-2.5 Max=-1 W eight= R angeVar[1]="f req" R angeMin[1]=9.95 Ghz R angeMax[1]=10.05 Ghz

Va r Eq n

S_Param SP1 Start=9 GH z Stop=11.0 GH z Step=10 Mhz

GOA L

GOAL

OPTIM Optim Optim1 OptimTy pe=R andom U seAllGoals=y es MaxIters=50 Sav eC urrentEF=no D esiredError=0.0 StatusLev el=4 FinalAnaly sis="N one" N ormalizeGoals=no SetBestValues=y es Seed= Sav eSolns=y es Sav eGoals=y es Sav eOptim Vars=no U pdateD ataset=y es Sav eN om inal=no Sav eAllIterations=no U seAllOptVars=y es

S-PA RA METERS

VAR VAR VAR 1 VAR 1 L_50=220.185 mil {o} L_50=220.185 mil {o} W _50=26.9633 m il {o} W _50=26.9633 m il {o} W _35=46.3406 m il {o} W _35=46.3406 m il {o} L_35=194.424 mil {o} L_35=194.424 mil {o}

Mea s Eq n

GOA L Goal OptimGoal3 Expr="isolation" Sim InstanceN am e="SP1" Min= Max=-30 W eight= R angeVar[1]="f req" R angeMin[1]=9.6 Ghz R angeMax[1]=10 Ghz

Goal Optim Goal2 Optim Goal2 Expr="port_match" Expr="port_match" Sim InstanceN ame="SP1" Sim InstanceN ame="SP1" Min= Max=-30 Max=-30 W eight= W eight= R angeVar[1]="f req" R angeVar[1]="f req" R angeMin[1]=9.6 Ghz R angeMin[1]=9.6 Ghz R angeMax[1]=10 Ghz R angeMax[1]=10 Ghz

VAR VAR 2 P_R F=-40 P_LO=-10 LOF req=9.6 Ghz R F Freq=10 Ghz

MeasEqn Means1 port_match=dB(S(1,1)) isolation=dB(S(1,3)) dif _coupling=dB(S(1,2))

Fig. 8. Schematic diagram of the hybrid ring design implemented in order to achieve port match at correct levels m5 freq= 10.00GHz dif_coupling=-2.835 optIter=49

MCFIL CLin2 Subst="MSub1" W=31.12 mil S=45.446 mil L=211.066 mil


Port P2 Num=2

-2.5

0 -10

dif_coupling


dB(S(1,4)) dB(S(1,3)) dB(S(1,2)) dB(S(1,1))

Port P1 Num=1

-20 -30 -40

-4.5 9.0

9.2

9.4

9.6

9.8

m6 freq= 9.600GHz isolation=-28.439 optIter=49

-20 -30

m7 freq= 10.00GHz isolation=-29.360 optIter=49

-40 -50

9.2

9.4

9.6

9.8

9.4

9.6

9.8

10.0 10.2 10.4 10.6 10.8 11.0

freq, GHz

-10 -20

m6

m7

-30 -40 -50 -60

10.0 10.2 10.4 10.6 10.8 11.0

9.0

freq, GHz

Fig. 7. Coupled lines input filter simulated response

9.2

Fig. 9. Hybrid ring simulated response to keep isolation and match the ports at correct levels

-10

9.0

9.0

10.8 11.0 10.0 10.2 10.4 10.6 10.8

freq, GHz

isolation

m3 freq= 10.12GHz dB(S(1,2))=-5.034

-3.5

-4.0

-60

m1 m2 freq= 10.00GHz freq=9.890GHz dB(S(1,2))=-4.942 dB(S(1,2))=-1.844 m1 m 2 m3 0

m5

-3.0

-50

Fig. 6. Schematic diagram of coupled lines input filter

dB(S(1,2))

Term Term 2 N um=2 Z=50 Ohm

M Su b MSU B MSub1 H =10.0 mil Er=2.17 Mur=1 C ond=5.8e7 H u=3.9e+034 mil T=0.15 m il TanD =0 R ough=0 mil

A. Coupled lines Input Filter Implementation of the input filter is based on a microstrip coupled lines sub circuit and theoretical calculations from the filter theory. The filter order is N=3 and the input and output impedances are 50 ohms. The schematics of coupled lines input filter is shown in Figure 6 while Figure 7 illustrates the corresponding simulated response.


MLIN TL1 Subst="MSub1" Subst="MSub1" W =W _35 W =W _35 L=L_35 L=L_35


Term Term 1 N um=1 Z =50 Ohm


9.2

9.4

9.6

9.8

10.0 10.2 10.4 10.6 10.8 11.0

freq, GHz

Fig. 10. Ports isolation simulated response: matching ports at accurate levels

The pass band attenuation is -1.844 dB and the bandwidth for 3 dB attenuation with reference to band pass attenuation is 230 MHz.

277

m8 freq= 10.00GHz port_match=-30.027 optIter=49

-10

port_match

-20

m9

m8

-30

m9 freq= 9.600GHz port_match=-29.209 optIter=49

-40 -50 -60 9.0

9.2

9.4

9.6

9.8

10.0 10.2 10.4 10.6 10.8 11.0

freq, GHz

Theoretically a LPF at 400 MHz plus half of the bandwidth will attenuate the undesired signals. However in practice, 400 MHz is a very low frequency compared with the 9.6 MHz frequency of LO. At 400 MHz a microstrip filter must have a big size transmission lines, because the wavelength is more less 70 cm. The solution is to design a LPF but with a comfortable cutoff frequency near 9.6 GHz that can be implemented with step impedances. The corresponding captured schematic is shown in Figure 14.

Fig. 11. Port matching simulated response: matching ports at accurate levels

S-PARAMETERS

MSub

m4 phase(S(3,4)) phase(S(3,2))

phase(S(1,4)) phase(S(1,2))

200

100

0

-100

100

0

S_Param SP1 Start=1.0 GHz Stop=20 GHz Step=10 MHz

MSUB MSub1 H=10.0 mil Er=2.17 Mur=1 Cond=5.8e7 Hu=3.9e+034 mil T=0.15 mil TanD=0 Rough=0 mil

200

m3

-100

m2 m1 -200

-200

9.0 9.0

9.2

9.4

9.6

9.2

9.4

9.8 10.0 10.2 10.4 10.6 10.8 11.0

9.6

9.8

10.0 10.2 10.4 10.6 10.8 11.0

freq, GHz

freq, GHz

m2 freq= 9.600GHz phase(S(1,4))=-155.907 optIter=49

m4 freq= 9.600GHz phase(S(3,2))=103.271 optIter=49



Fig. 12. Phase shift simulated response: matching ports at accurate levels

MLIN TL6 Subst="MSub1" W=228.181102 mil L=30.858386 mil

Term Term1 Num=1 Z=50 Ohm

MSTEP Step1 Subst="MSub1" W1=228.18 mil W2=2.8518 mil





The final implementation of hybrid ring is shown in Figure 13. Term Term2 Num=2 Z=50 Ohm

MLIN TL16 Subst="MSub1" W=30.838583 m il L=225.253150 m il MTEE_ADS Tee6 Subst="MSub1" W1=30.839 m il W2=30.839 m il W3=30.839 m il

MTEE_AD S Tee2 Subst="MSub1" W 1=W_35 W 2=30.839 m il W 3=W_50

MLIN TL1 Subst="MSub1" W=W_35 L=L_35

MLIN TL10 Subst="MSub1" W=30.839488 m il L=432.444882 m il

MLIN TL5 Subst="MSub1" W=W _50 L=L_50

MTEE_ADS Tee4 Subst="MSub1" W1=W _35 W2=30.839 m il W3=W _50

MLIN TL2 Subst="MSub1" W=W _35 L=L_35

MTEE_ADS Tee3 Subst="MSub1" W1=30.839 mil W2=W _35 W3=W _50


m2 freq=9.600GHz dB(S(1,2))=-41.902

-10 di_hp_HSMS8202_20000301 D1

MLIN TL22 Subst="MSub1" W =30.838583 m il L=112.626772 m il MLIN TL24 Subst="MSub1" W=30.838583 mil L=112.626772 m il

MLIN TL15 Subst="MSub1" W=30.838583 m il L=225.253150 m il


m1

0

MLIN TL6 Subst="MSub1" W=W_50 L=L_50 MTEE_ADS Tee5 Subst="MSub1" W1=30.839 mil W2=30.839 mil W3=30.839 mil

MLIN TL8 Subst="MSub1" W=30.838583 mil L=432.444882 m il



Fig. 14. Schematic diagram of step impedance Low pass filter

MLIN TL7 Subst="MSub1" W=30.839488 mil L=216.222047 m il

MSOBND_MDS Bend2 Subst="MSub1" W=30.839488 m il

dB(S(1,2))

MTEE_ADS Tee1 Subst="MSub1" W1=30.839 mil W2=W_35 W3=W_50


m2 -40

-60 0

MSOBND_MDS Bend1 Subst="MSub1" W=30.839488 m il

For the hybrid ring, the distance and impedances have been computed using ‘linecalc’ function. The final implementation has a number of bend lines to keep the layout easy to build. Another important point to mention in this schematic is the use of two λ/4 short circuit terminated stubs at the middle frequency between the LO and the RF. This is to provide a path to ground for the DC signal generated by the diodes in the mixing process.

m1 freq=3.800GHz dB(S(1,2))=-3.265

-30

-50

MLIN TL23 Subst="MSub1" W =30.838583 m il L=225.253150 m il

Fig. 13. Schematic diagram of hybrid ring circuit with filtered RF signals and LO signals as inputs to the diode pair

-20

2

4

6

8

10

12

14

16

18

20

freq, GHz

Fig. 15. Step impedance LPF simulated response: avoiding output of LO/RF frequencies and inter-modulation products from the IF band

D. λ/4 Stub Using an output filter will bounce back the LO and RF energy to the ring because at 9.6 GHz and 10 GHz the filter shows high impedance. The solution is to put a stub in the diode middle output with the behavior of a band stop filter, with enough bandwidth to cover the entire LO and RF frequencies. This element is the microstrip radial stub (Figure 16). Using the Tune tool in ADS, the center band reject frequency can be adjusted as depicted in Figure 17.

C. Step Impedance LPF The LPF in a mixer has to avoid the output of the LO and the RF frequencies and also other inter-modulation products from the IF band. For the mixer in our case, the IF was 400 MHz and the bandwidth was about 200 MHz.

278

MRSTUB Stub1 Subst="MSub1" Wi=30.83 mil L=146.22 mil Angle=70

Mixer Conversion Gain, Isolation, and Port Impedance versus LO Power Sweep

Edit these values Var Eqn

RF and IF spectra, conversion gain, isolation and all port impedances for a single-ended mixer.

MSub Term Term2 Num=2 Z=50 Ohm

Term Term1 Num=1 Z=50 Ohm

MSUB MSub1 H=10.0 mil Er=2.17 Mur=1 Cond=5.8e7 Hu=3.9e+034 mil T=0.15 mil TanD=0 Rough=0 mil

S-PARAMETERS S_Param SP1 Start=1.0 GHz Stop=15.0 GHz Step=100 Mhz

I_Probe I_load

I_Probe I_RFin VRFin

Vload

RF

mixer180_con_diodos_optim_final X1

I_1Tone SRC2 I= Freq=RFfreq+LOfreq I_USB=1 mA

I_1Tone SRC1 I= Freq=IFfreq I_USB=1 mA I_LSB=1 mA

LO

I_Probe I_LOin VLOin

P_nHarm PORT3 Num=3 Z=50 Ohm Freq =LOfreq P[1]=polar(dbmtow(P_LO),0) P[2]=polar(dbmtow(P_LO-100),0) P[3]=polar(dbmtow(P_LO-100),0)

0

dB(S(1,2))

H AR MONIC BALANC E

Term IFport Num=2 Z=Zload

IF

P_1Tone RFport Num=1 Z=Zsrc P=dbmtow(P_RF) Freq =RFfreq

Fig. 16. Schematic diagram of microstrip radial stub

-20

VAR VAR1 P_RF=-20 LOmin=-10 LOmax=15 LOstep=0.2 RFfreq =10 GHz LOfreq=9.6 GHz Bias=3.3 V Zsrc=50 Ohm Zload=50 Ohm

m2 freq=10.00GHz dB(S(1,2))=-32.261

m1 m2

-60

Var Eqn

VAR VAR2 IFfreq=mag (LOfreq -RFfreq ) P_LO=0

Dis p Temp

DisplayTemplate disptemp1 "GS_Mix_SE_CG_LOswp"

Fig. 18. Schematic for evaluation of conversion loss

m1 freq=9.600GHz dB(S(1,2))=-31.514

-40

HarmonicBalance HB2 MaxOrder=6 Freq[1]=LOfreq Freq[2]=RFfreq Order[1]=9 Order[2]=3 SS_MixerM ode=yes SS_Freq =1.0 kHz SweepVar="P_LO" Start=LOmin Stop=LOmax Step=LOstep

m2 indep(m2)=6.800 plot_vs(Mix2_Down_ConvGain, HB.P_LO)=-7.775 Conversion Gain (dB)

2

4

6

8

10

12

14

Mix2_Up_ConvGain Mix2_Down_ConvGain

0

16

freq, GHz 0

-5

dB(S(1,1))

m2

0

-80

-10

-10 -20 -30 -40 -50 -60 -70 -10

-5

0

-15

5

10

15

LO Power, dBm

Fig. 19. Conversion loss as a function of LO power -20 2

4

6

8

10

12

14

16

freq, GHz

Fig. 17. Center band reject frequency adjustment

IV. PERFORMANCE EVALUATION AND SIMULATION RESULTS The complete mixer circuit is evolved as a result of combining several subsystems discussed in Section III. The resultant overall circuit having three ports: RF input, LO input and IF output is shown in Figure 16 (at the end of paper). The next step is essentially to investigate the performance of the designed mixer. Various conversion losses have been analyzed and important parameters like noise figure and 3rd order interception point have been computed. A. Conversion Loss versus LO Power and Isolation In this simulation scenario carried out by the HP ADS 2008 assistant for mixers, the conversion loss versus LO power and isolation has been evaluated. The mixer sub circuit has to be placed in the test point, and the frequency parameters must be adjusted to the mixer specifications (Figure 18). The simulation results include plot of the conversion loss with LO power with the RF input kept at -40 dBm (Figure 19 to 20).

For an optimum point we have chosen LO Power of 6.8 dBm, the conversion loss was 7.77 dB for the whole mixer, including all the filters. The port to port isolation simulation results are shown in Figure 20. The isolation between the LO port and the IF port was -79 dB and between the LO to RF ports, it was -40.7 dB. Port-to-Port Isolation (dB) -20

m3

m4 indep(m4)= 6.800 plot_vs(Mix2_LO2IF, HB.P_LO)=-79.061

-40

M ix2 _ R F 2 IF M ix 2 _ L O 2 R F M ix2 _ L O 2 IF

0

-60

m4

-80

m3 indep(m3)= 6.800 plot_vs(Mix2_LO2RF, HB.P_LO)=-40.724

-100 -120 -10

-5

0

5

10

15

LO Power, dBm

Fig. 20. Port to port isolation simulated response

B. Conversion Loss versus Input Frequency Using the mixer simulation assistant, the mixer has been simulated with the same template. In this case, the parameter to sweep was input frequency. Figure 20 depicts values of parameters used in this simulation scenario.

279

D. Third Order Interception Point The third order interception point has been simulated with two tones and a power sweep to see the input power needed to obtain a third order harmonic power equal to the main conversion signal output. We have fed the mixer with two tones separated by a small frequency difference and have simulated the mixer for a sweep of input power (Figure23). The two input tones were separated by 125 MHz, centered on 10 GHz, so the outputs will be 10.0625 GHz – 9.6 GHz and 9.9475 GHz – 9.6 GHz. The third order intermodulation was placed at 375 MHz (n*125MHz where n=3). The results are presented in Table II. Fig. 21. Parameters value for evaluation of conversion loss as a function of input frequency

RF_INPUT P_nTone PORT1 Num=1 Z=50 Ohm Freq[1]=RFFreq + FSpacing/2 Freq[2]=RFFreq - FSpacing/2 P[1]=polar(dbmtow(P_RF),0) P[2]=polar(dbmtow(P_RF),0)

To get a frequency response including the image rejection response, the sweep will be from 9 GHz to 10.3 GHz with a step size of 50 MHz step size. The corresponding results are shown in Figure 21. The 3dB bandwidth was 200MHz centered at 10 GHz and the image rejection at 9.2GHz was 42.39dB. In Figure 21, the false gain peak at 9.6 GHz is because the LO output is in the IF. m4 RFfreq= 9.200E9 Mix3_Down_ConvGain=-42.390

Mix3_Down_ConvGain Mix3_Up_ConvGain

20

m2 RFfreq= 1.009E10 Mix3_Down_ConvGain=-10.271

m4

-40 -60

RF Power (dBm) -40 -10 0 10 20 30


-80 -100

10.40G

10.20G

10.00G

9.800G

9.600G

9.400G

9.200G

9.000G

RF (Input) Frequency

Fig. 22. Conversion loss as a function of input frequency simulated response

C. Noise Figure The Noise Figure (NF) and contribution of each element of the proposed mixer has been analyzed using the ADS 2008 has an assistant. The results have been shown in Figure 23.The simulation results show a NF of 7.281 dB with LO power of 6.8 dBm. Single-Ended Mixer Single Side Band Noise Figure and Conversion Gain Single Sideband Noise Figure, dB

Output Frequency 400.0 MHz

Source Impedance 50.0 / 0.0

Noise Contributions

7.281

Load Impedance

LO Power (dBm)

50.0 / 0.0

6.8

0 1 2 3 4 5 6 7

_total X1.D1 X1.D1.DIODE1.id X1.D1.DIODE1.Rs X1.D1.DIODE2.id X1.D1.DIODE2.Rs X1.DA_CLFilter1.CL... X1.DA_CLFilter1.CL...

382.5 344.0 222.6 121.7 201.9 115.0 91.47 77.16

LO Frequency

Input Frequency 10.00 G

9.600 G

A SSB noise figure is calculated at the specified RF input frequency. The noise contributions at the image frequency, LO harmonics, and output termination are also included in the total output noise.

mixer180_con_diodos_optim_final X1 LO_INPUT

P_1Tone PORT2 Num=2 Z=50 Ohm P=polar(dbmtow(P_LO),0) Freq=LOFreq

HARMONIC BALANCE HarmonicBalance HB1 MaxOrder=10 Freq[1]=LOFreq Freq[2]=RFFreq +FSpacing/2 Freq[3]=RFFreq - FSpacing/2 Order[1]=10 Order[2]=10 Order[3]=10


m3 m1m2

-20

VAR VAR2 P_RF=20 P_LO=6.8 {t} LOFreq=9.6Ghz RFFreq=10 Ghz FSpacing=125 Mhz

R R3 R=50 Ohm {-t}

Fig. 24. Schematic of analysis of 3rd order interception point

Conversion Gain (dB) 0

Var Eqn

IF_OUTPUT

freq

400.0 MHz

Fig. 23. Results from the Noise Figure analysis

ConvGain

-7.744

pV pV pV pV pV pV pV pV

TABLE II SIMULATION RESULTS 3rd Order O/p Power IF O/p Power (dBm) (dBm) -225 -48 -91 -18 -69 -10 -35 -8 -30 -13 -19 -22

The simulation results are in coincidence with the theoretical concept that explains that 3rd order interception point is about 10 dB or 15 dB above the 1dB compression point. V. LAYOUT IMPLEMENTATION After design and evaluation of the mixer, Agilent ADS has been used to implement the microstrip layout of the schematic. Steps for implementing the proposed layout have been mentioned below. The final layout of the Printed Circuit Board (PCB) has been shown in Figure 26. 1. Include extra T.L to be able to solder the connectors (SMA in-circuit female type). 2. Import the components and Microstrip lines to layout. 3. Move them to the right positions to keep the board without extra tracks. 4. Identify the needs of transmission line bends to fit the point. 5. Place the bends in the schematic. 6. Start again the process from point 1 until no more bends required.

280

REFERENCES

VI. CONCLUSION [1] [2]

The simulated performance of the optimized mixer has been tabulated in Table III.

[3]

TABLE III MIXER PARAMETERS Parameter Optimum LO power Conversion loss Bandwidth 3 dB Image rejection Isolation LO vs. IF Isolation LO vs. RF Noise Figure 3rd Order interception point

[4]

Value 6.8 dBm 7.7 dBm 200 MHz 42.39 dB -79 dB -40.7 dB NF = 7.2dB. + 30 dBm (RF I/p power)

[5]

[6] [7]

[8]

From these parameters, it is clear that the proposed work presents a realistic diode mixer. Further optimization can be done in the hybrid ring, biasing to process to obtain more isolation between the LO and IF or RF. A conversion loss of 7.7 dB is acceptable taking into account that the mixer incorporates the input and the output filters. We have designed the components to be implemented using only microstrip techniques, except the transistor and a low pass coil to eliminate the DC level at the output, because this coil has to be able to present a high impedance at 400 MHz which is too low frequency for a microstrip transmission line. A prototype of PCB has also been designed and has been sent for fabrication. MLIN TL16 S ubst="MS ub1" W=30.838583 mil L=225.253150 mil

180 R ing at 10 Ghz plus diodes

MTE E _A DS Tee1 S ubst="MS ub1" W 1=30.839 mil W 2=W _35 W 3=W _50

RF_INP UT P ort P1 Num=1

rf_inpu_filtr DA _CLFilter1_mixer180_con_diodos_optim DA _CLFilter1 S ubst="MS ub1" Fs1=9.2 GH z Fp1=9.9 GH z Fp2=10.1 GHz Fs2=10.8 GHz A p=3 dB A s=30 dB N=0 Zo=50 Ohm CouplingType=Coupled Line Transformer Input Delta=0 mil

Var Eqn

MLIN TL1 S ubst="MS ub1" W =W_35 L=L_35

MLIN TL10 S ubst="MS ub1" W=30.839488 mil L=432.444882 mil


MRS TUB S tub1 S ubst="MS ub1" W i=30.83 mil {t} L=146.22 mil {t} A ngle=70 {t}

MLIN TL7 S ubst="MS ub1" W =30.839488 mil L=216.222047 mil


MLIN TL6 S ubst="MS ub1" W =W _50 L=L_50

MTE E _A DS Tee4 S ubst="MS ub1" W1=W_35 W2=30.839 mil W3=W_50

MS ub

MS UB MS ub1 H=10.0 mil E r=2.17 Mur=1 Cond=5.8e7 Hu=3.9e+034 mil T=0.15 mil TanD=0 Rough=0 mil


MTE E _A DS Tee3 S ubst="MS ub1" W 1=30.839 mil W 2=W _35 W 3=W _50


di_hp_H S MS 8202_20000301 D1

MTE E _A DS Tee5 S ubst="MS ub1" W1=30.839 mil W2=30.839 mil W3=30.839 mil


VAR V A R3 L_50=220 mil W_50=26.97 mil W_35=46.34 mil L_35=196 mil

MTE E _A DS Tee6 S ubst="MS ub1" W 1=30.839 mil W 2=30.839 mil W 3=30.839 mil

MTE E _A DS Tee2 S ubst="MS ub1" W 1=W _35 W 2=30.839 mil W 3=W _50

LO_INP UT P ort P3 Num=3

David M. Pozar, “Microwave Engineering”, 2nd ed. Sung Tae Choi, Design of Microstrip Balanced Diode Mixer for 38 GHz Band L. Pradell, Design and Analysis of RF and MW Systems Course Slides, UPC, Barcelona, Spain 2008 Max W. Medley, “RF and MW Circuits: Analysis, Synthesis and Design”, Artech House Inc, December 1992 Hunter, M.T.J. “The Basics of System Design”, Proceedings of the IEE Tutorial Colloquium on “How to Design RF Circuits”, Wednesday 5th April 1999, Savoy Place, London Maas, S.A. “Microwave Mixers”, Artech House, ISBN 0-89006-605-1 Walker, J.L.B. “Filters”, Proceedings of the IEE Tutorial Colloquium on “How to Design RF Circuits”, Wednesday 5th April 1999, Savoy Place, London Walker, J.L.B. ”Improvements to the Design of the 180° Rat Race Coupler and its Application to the design of Balanced Mixers with High LO to RF Isolation”, IEEE MMT-S Digest, 1997, Vol. II, pp 747-750

MLIN TL22 S ubst="MS ub1" W=30.838583 mil L=112.626772 mil MLIN TL24 S ubst="MS ub1" W=30.838583 mil L=112.626772 mil


MLIN TL23 S ubst="MS ub1" W=30.838583 mil L=225.253150 mil MS OB ND_MDS B end1 S ubst="MS ub1" W=30.839488 mil

Fig. 25. Schematic of overall mixer circuit

Fig. 26. Schematic Diagram of Mixer layout PCB

281

MS OB ND _MDS B end2 S ubst="MS ub1" W =30.839488 mil

P ort P2 N um=2

MTE E _A DS Tee7 S ubst="MS ub1" W 1=30.839 mil W 2=30.839 mil W 3=30.839 mil

DC _Feed DC _Feed1



MS TE P S tep1 S ubst="MS ub1" W 1=228.18 mil W 2=2.8518 mil



MS TE P S tep5 S ubst="MS ub1" W1=228.18 mil W2=2.8518 mil




MS TE P S tep6 S ubst="MS ub1" W1=228.18 mil W2=2.8518 mil
