Design and Validation of a Reconfigurable Single ... - IEEE Xplore

IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 61, NO. 2, FEBRUARY 2013

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Design and Validation of a Reconfigurable Single Varactor-Tuned Reflectarray Francesca Venneri, Member, IEEE, Sandra Costanzo, Senior Member, IEEE, and Giuseppe Di Massa, Senior Member, IEEE

Abstract—An aperture-coupled reflectarray element giving a full phase tuning range with a single varactor diode is proposed in this paper for pattern reconfigurability applications. The full phase agility is achieved by a proper optimization of the phase tuning line, thus providing an alternate inductive/capacitive effect able to avoid the use of two varactor diodes, usually adopted in similar existing configurations. The proposed active element structure is adopted to design a demonstrative reflectarray prototype of 3 15 radiators. Furthermore, an own synthesis procedure is applied to obtain the proper biasing voltages giving the prescribed H-plane field. Test examples of beam-scanning, multibeam, and shaped-beam patterns are discussed to demonstrate the effectiveness of the approach. Index Terms—Beam-scanning antennas, microstrip arrays, reconfigurable antennas, reflectarray antennas, reflector antennas, varactors.

I. INTRODUCTION

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ICROSTRIP reflectarrays have received great interest since their first use as microwave or millimeter-wave antennas [1], [2]. They consist of a printed array illuminated by a low directivity antenna. Each radiating element is properly designed to give a phase response able to create a total reradiated field coherent along a desired direction or, more in general, characterized by a custom-shaped pattern. Many different reflectarray configurations have been proposed in literature, each one based on the idea of varying one or more geometrical/electrical parameters of the single radiator, in order to produce a proper change in the reflection phase [3]. The reflectarray configuration offers significant advantages over traditional reflector antennas. As a matter of fact, the combined use of the spatial feeding approach with the microstrip technology leads to high-efficiency antennas with low profiles and reduced mass. The only real limitation of reflectarray antennas is related to the narrowband behavior, which is caused by two principal factors: the intrinsic reduced bandwidth of the single microstrip radiators and the frequency dependence of the differential spatial delay among the different feed-elements paths. However, the great efforts devoted to overcome this last problem have led to very appealing reflectarray configurations, which offer quite a large operational frequency band [4]–[7]. Manuscript received June 21, 2012; revised August 21, 2012; accepted October 09, 2012. Date of publication October 24, 2012; date of current version January 30, 2013. The authors are with Dipartimento di Elettronica, Informatica e Sistemistica, Università della Calabria, 87036 Rende, Italy (e-mail: [email protected]). Color versions of one or more of the figures in this paper are available online at http://ieeexplore.ieee.org. Digital Object Identifier 10.1109/TAP.2012.2226229

The overall benefits resulting from the use of microstrip reflectarrays make this type of antennas very attractive also for applications requiring beam-scanning capabilities or pattern reconfigurability. As a matter of fact, tunable reflectarrays may offer many advantages over traditional phased-arrays, due to the absence of complicated beam-forming networks. Several reconfigurable reflectarray elements have been investigated in literature, even if only some of them have been validated through the realization and the measurement of a reflectarray prototype. A mechanical approach has been adopted in [8] and [9], where miniature motors have been employed to reconfigure, respectively, the elements rotation and the displacement of a dielectric rod under each radiator. More recently, many other reflectarray configurations have been proposed, which are based on the use of electronically tunable microstrip elements. MEMs-based structures have been presented in [10]–[15], a tunable liquid-crystal reflectarray has been designed in [16], while in [17] the performances of a reconfigurable element based on BST thick-film ceramic have been discussed. Different solutions based on the integration of a microstrip radiating element with one or more semiconductor devices, such as PIN or varactor diodes, have been proposed in [18], [19], and [20]–[23], respectively. The present work focuses on this last approach and is based on the use of varactor-tuned reflectarray elements. This configuration offers a continuous phase tuning over a range of about 360 [20]–[23], thus providing a higher degree of reconfigurability. In [20], a rectangular patch is loaded with a shunt connected varactor diode, whose position is properly chosen in order to maximize the phase tuning range. In [21], a microstrip dipole is centrally loaded with a varactor diode, while in [22] the two halves of a rectangular patch are connected through two surfacemounted varactors. All these solutions offer a good phase agility essentially due to the variation in the element resonant frequency caused by the tuning of the varactors capacitance. In [22], the effectiveness of the proposed structure is also demonstrated in the implementation of a 5.8-GHz beam-scanning reflectarray prototype with 10 7 elements. However, despite their good phase agility, the above configurations are characterized by a reflecting surface with several discontinuities due to the presence of varactor diodes, biasing circuitries and solders, which in turns may create undesired scattered field components. A different concept, overcoming the above problem, is adopted in [23], where the use of an aperture-coupled configuration, first proposed in [24], is adopted to obtain an active structure by the insertion of two varactor diodes on the tuning line. In this case, a uniform reflecting surface is obtained by hosting the active network on a different

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layer from the radiating one. The functionality of this last configuration is demonstrated on a 5 6 elements reflectarray prototype with beam-scanning capabilities at the operating frequency of 5.4 GHz [23]. However, the above tunable configuration requires the use of two diodes on each reflectarray element in order to achieve a full phase tuning range. In the present work, a configuration based on the use of an aperture-coupled patch electronically driven by a single varactor diode is considered. A microstrip line printed on a different substrate is designed to realize the phase tuning of the field reflected by the patch. One line side terminates on a variable capacitance diode, while the other one is open ended. The radiating patch receives a linearly polarized wave, which couples to the varactor loaded line through the aperture in the ground plane. By changing the bias voltages across the diode, a variable phase shift is added to the coupled wave, that is reflected back to the patch and then reradiated in the same polarization direction. By properly sizing the line length, with particular attention to the extent of the open ended section, the phase agility of the antenna is maximized. In fact, as already demonstrated by the authors in [25], the above reflectarray element may offer a full phase tuning range of about 320 . Although a similar configuration has been proposed in [26], a limited 120 -phase range has been demonstrated. The aim of this work is to prove the effectiveness of the proposed tunable element in the implementation of a reconfigurable reflectarray antenna. At this purpose, the design of a reflectarray prototype with beam-scanning capabilities is fully discussed to validate the approach. As a further proof of the good degree of reconfigurability, the realized reflectarray antenna is properly controlled in order to obtain some examples of multibeam and cosecant shaped patterns. The prototype is composed by 3 15 elements, each one loaded by a varactor diode. An electronic DAC-board is integrated to the antenna in order to supply the desired bias voltages to the diodes. The small size of the antenna is expressly fixed in order to reduce the costs and the complexity of the structure. However, one cut of the array is large enough to realize the reconfiguration of the radiation pattern in the corresponding plane. The paper is organized as follows. Section II describes the design details of the single reflectarray element. First, a parametric analysis of the reflection phase behavior against the length of the varactor loaded line is discussed. Then, simulation and experimental results relative to the designed element are presented. Section III describes the realization of the reflectarray prototype, while Section IV illustrates the results obtained from the synthesis and the measurements of the reflectarray antenna for some prescribed radiation pattern behaviors. Finally, conclusions are outlined in Section V. II. RECONFIGURABLE REFLECTARRAY UNIT CELL The reconfigurable reflectarray unit cell consists of a rectangular patch printed on a grounded dielectric substrate and slot coupled to a varactor loaded microstrip line (Fig. 1). The line is composed by two sections, respectively given by a segment terminating on the varactor diode and an open ended stub . A dc-bias network, printed on the same layer hosting the line, imposes a tunable reverse voltage across the diode.

Fig. 1. Reflectarray element: (a) top view and (b) side view.

By changing the amplitude of the dc signal, the different capacitance associated to the varactor creates a phase variation in the field reflected by the element. The tunable phase shift introduced by a single diode usually varies within a limited 180 range, as a result of the capacitive nature of the varactor reactance [26]. However, the introduction of a proper inductive load in the varactor-based tuning circuit increases the range of allowable phase variation. Starting from this concept, the antenna phase agility is maximized by properly sizing the length of both line sections and . In particular, by adjusting the length of the stub , usually adopted in the aperture coupled antenna to cancel the reactance associated to the slot [27], a suitable inductive effect is added to the varactor loaded line . In this manner, it is possible to obtain a tunable coupled line offering a capacitive reactance for large values of the biasing voltage and a pure inductive impedance for small values of . In such cases, the coupled-line behaves like a tunable LC circuit, thus acting as a 360 phase shifter. A. Characterization of the Varactor-Based Tunable Line The varactor loaded line, representing the tunable element of the proposed reflectarray configuration, is schematized under Fig. 2(a). The input impedance at section AA (slot side) is the series combination of the transmission line segments and . By properly sizing the two different sections, an alternate capacitive-inductive behavior is obtained in the phase of the reflected signal towards the coupled patch.

VENNERI et al.: DESIGN AND VALIDATION OF A RECONFIGURABLE SINGLE VARACTOR-TUNED REFLECTARRAY

Fig. 2. Details of the tunable line at section AA (slot side): (a) equivalent circuit model of the coupled-line; (b) ideal impedance of the varactor diode; and (c) nonideal impedance of the varactor diode.

Fig. 3. Phase of coefficient versus diode capacitance computed for difand [ideal case of Fig. 2(b)]. ferent lengths of the line sections

This fact can be clearly observed in Fig. 3, which illustrates the phase of the coefficient , evaluated at the section AA of a 50 -line with a variable capacitive load [Fig. 2(b)]. The reflection phase is computed by varying the capacitance from 0 to 4 pF, for different lengths of the two line sections and . In the same figure, it can be observed that a proper choice of the lengths and results in a wider phase tuning range. Furthermore, it can be stated that, while the section introduces the desired inductive effect, the length is tuned in order to match the maximum phase variation with the available varactor capacitance range. For example, assuming to employ a varactor model with a capacitance ranging from 0.2 to 2 pF, the best behavior in terms of phase agility is provided by the curve computed in correspondence of the lengths 0.46 and 0.4 (Fig. 3). In this case, the tunable line alternately behaves as follows: an open circuit for low values of the varactor capacitance, a short circuit for approaching to values giving the resonance of the line, and finally a decreasing inductive load that for higher values of may tend back to an open circuit. In this way, a full phase tuning range approaching the 360 is achieved.

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Fig. 4. Phase of coefficient versus diode capacitance computed for difand [nonideal case—Fig. 2(c)]. ferent lengths of the line sections

Fig. 5. Amplitude of coefficient different lengths of the line sections

versus diode capacitance computed for and [nonideal case of Fig. 2(b)].

In practical cases, the impedance associated to a varactor diode is given by the equivalent circuit model illustrated in Fig. 2(c), which takes into account for the package parasitic effects and the diode losses . These parameters affect the resonance condition of the tuning circuit, so that the line length must be modified in order to recover the desired performances within the assumed capacitance range. Figs. 4 and 5 show the coefficient (phase and amplitude) computed when the diode series resistance is fixed at a value of 0.7 and the device is characterized by values of parasitic inductance and capacitance respectively equal to 0.2 nH and 0.15 pF. By comparing the phase curves in Fig. 4 with those depicted in Fig. 3, it can be observed that the new load configuration imposes a dominant inductive behavior for smaller values of the diode capacitance, due to the presence of a series package inductance. As a consequence of this, the maximum phase tuning range is now achieved when the length of the sections and are fixed to 0.46 and 0.35 , respectively. Furthermore, Fig. 5 shows a reduction in the amplitude of the reflection coefficient due to the series resistance of the diode. These losses have a maximum at the line resonance condition and they will affect the

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Fig. 6. Amplitude of coefficient versus diode capacitance computed for 0.35 0.46 . different values of the diode series resistance R

total reflection loss of the aperture-coupled reflectarray element, as demonstrated in the next paragraph. However, this negative effect can be reduced by choosing a diode with a lower series resistance, as illustrated in Fig. 6. B. Element Design and Simulation The proposed radiating structure is adopted for the design of a reconfigurable reflectarray element operating at a frequency 1 1.5 GHz. The synthesized radiator consists of a rectangular patch, having dimensions 9.3 mm and 8.2 mm (Fig. 7), printed on a substrate composed by a 0.762-mm-thick layer of Diclad870 with a relative dielectric constant 2.33, and a layer of air with a thickness of 0.762 mm. The patch is coupled to a phase tuning line through a rectangular aperture sliced in the ground plane, which gives the best coupling when its sides are fixed, respectively, to 0.6 mm and 5.8 mm. The varactor loaded line is printed on a dielectric substrate with permittivity 2.33 and thickness 0.762 mm. The line width is fixed to the value 3.07 mm, which corresponds to an intrinsic impedance equal to 40 . An optional metallic plate is placed at a distance 6.5 mm from the line layer, in order to reduce the back-radiation. As discussed above, the two line sections and must be determined in order to maximize the phase agility of the element. At this purpose, the results obtained in the case of the isolated tunable line, illustrated in Fig. 2(a), are properly corrected in order to include the effects of the slot reactance. In other words, the length of the stub , that is usually adopted to compensate for the excess reactance of the slot coupled antenna, is tuned to give the appropriate inductive load that, placed in series to the impedance of the varactor tuned line , maximizes the element phase tuning range. Parametric full wave simulations of the reflectarray element are performed by adopting the infinite array approach and assuming a normally incident plane wave. The cell size is fixed to at the operating frequency . The varactor load is modeled with the equivalent circuit illustrated in Fig. 2(c). A Microsemi MV31011-89 hyperabrupt varactor diode is considered [28]. The parasitic effects due the diode package are modeled by a capacitance 0.15 pF and

Fig. 7. Synthesized aperture coupled patch: (a) top view and (b) input impedance.

an inductance 0.2 nH. The varactor series resistance is assumed equal to 1.36 , as derived from the manufacturer datasheet [28]. The parametric analysis of the radiating structure is carried out by varying the diode capacitance from 0.2 pF to 2 pF for different values of the stub length, starting from 2 mm, which corresponds to the matched case, up to 7.8 mm, that is equivalent to an inductive load of 2.12 nH at the operating frequency [see Fig. 7(b)]. The length of the line is fixed to the value of 6.5 mm, which places the maximum variation of the computed reflection phase curves within the considered varactor capacitance range (Fig. 8). In the same figure, it can be observed that a wider phase tuning range is obtained by increasing the inductance associated to the stub . As illustrated in Fig. 8, the reflection losses increase when the phase agility of the element is improved. These losses are essentially due to the resonance of the tuning line. This effect can be reduced by adopting a diode with a higher and a lower series resistance. Typically, the high varactor models which provide a capacitance variation range suitable for the proposed reflectarray configuration, are characterized by a minimum value of equal to 0.7 [28]. C. Element Prototype and Measurement A small array of 5 5 elements is considered as validation test for the designed reflectarray unit cell. A photograph of the

VENNERI et al.: DESIGN AND VALIDATION OF A RECONFIGURABLE SINGLE VARACTOR-TUNED REFLECTARRAY

Fig. 8. Simulated reflection response versus diode capacitance for different values of stub length .

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Fig. 10. Biasing line design: (a) layout of the printed biasing line connected to a 40 transmission line and (b) simulated scattering parameters.

Fig. 9. Reflectarray prototype under test for the single element validation: (a) patch layer and (b) varactor loaded lines with a particular of the biasing line.

fabricated antenna is reported in Fig. 9. Each element is loaded by a Microsemi MV31011-89 varactor, the same adopted in Section II.B, characterized by the parameters 0.15 pF, 0.2 nH, 1.36 , and providing a capacitance which varies from 0.2 to 2 pF in the presence of a bias voltage from 0 V (giving 2 pF) to 20 V (giving 0.2 pF). A dc biasing network (dc-line) is properly connected to the tuning lines, thus obtaining a parallel distribution of the applied bias signal. Each branch of the dc-line consists of a high impedance line with a couple of radial stub [Fig. 10(a)], so resulting as a RF-chock in the operational frequency band of the antenna, as illustrated in Fig. 10(b). The reflectarray prototype is tested with a measurement system composed by two transmitting and receiving horn antennas, both connected to a vectorial network analyzer [25]. The instrumentation detects the amplitude and phase of the far-field component reflected by the antenna along the broadside direction. The adoption of a small array instead of a single

Fig. 11. Measured reflection phase versus frequency and varactor bias voltage.

reflectarray element improves the directivity of the test, so making the measurements less vulnerable to the external noise. Furthermore, the adopted measurement setup takes into account the mutual coupling effects due to the surrounding elements. The phase curves of the antenna are measured at different frequencies within the operating band, by varying the applied bias voltages from 0 to 20 V. As remarked at the beginning of the section, this corresponds to a variation of the diode capacitance from 0.2 to 2 pF, thus providing results comparable to those obtained in the parametric analysis of Section II-B. The measured curves reported in Fig. 11 demonstrate a phase tuning range of about 320 , within the frequency range from 11 GHz up to 11.6 GHz. Furthermore, a good agreement between simulations and measurements can be observed in Fig. 12, illustrating the reflection phase vs varactor bias voltage for different frequencies.

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Fig. 14. Photograph of the reflectarray protoype.

Fig. 12. Comparisons between simulated and measured reflection phase versus bias voltages for different frequencies.

Fig. 13. Scheme of the designed reflectarray antenna (H-plane view).

III. DESIGN OF RECONFIGURABLE REFLECTARRAY ANTENNA A reflectarray prototype of 3 15 radiators is fabricated in order to prove the effectiveness of the proposed reconfigurable element. An interelement spacing of is fixed at the operating frequency of 11.5 GHz. A scheme of the entire radiating structure is reported in Fig. 13. The array is illuminated by an -band horn, which is placed in the E-plane at a distance of 34 cm from the reflecting surface, with an offset angle of 15 . The horn is characterized by an aperture size equal to 4.8 mm 5 mm, giving a pattern with 10-dB beamwidths equal to 127 and 102 , respectively, in the two principal Hand E-planes. A circuit board composed of 12 digital-to-analog converters (DACs) driven by a microcontroller is placed in the back side of the antenna [29]. The output lines of the DAC board is properly connected to each reflectarray tunable element, so delivering the desired bias voltages to all varactor diodes. A copper plate of dimension equal to 6.5 cm 28.4 cm is placed between the antenna and the DAC board (Fig. 13), thus reducing the back-radiation caused by the slots, while eliminating at the same time any unwanted coupling between the electronic control board and the radiating structure. By varying the bias voltages distribution on the varactor loaded elements, the radiation pattern behavior of the antenna can be properly modified. However, the small number of elements composing the illustrative prototype imposes some restrictions on the size of feasible solutions admitted by the antenna pattern synthesis problem. In particular, the adopted

array lattice limits the ability to reconfigure the radiation pattern shape only in the H-plane, as it contains the maximum number of elements. The reconfigurability of the fabricated antenna is first proven by demonstrating the beam-scanning capabilities in the H-plane, then the radiation pattern is reconfigured in order to obtain a multibeam and a cosecant beam shape. The reflectarray prototype is designed with the only purpose of verifying the reconfigurability of the proposed tunable configuration, therefore the other antenna performances are not optimized. As a matter of fact, the small size of the reflecting surface, in conjunction with the large feed beamwidth, causes very high spillover losses; thus, the aperture efficiency of the antenna is very low. Furthermore, the feed position is not optimized for the minimization of the blockage effects in order to preserve the integrity of the antenna pattern behavior, but it is given by the right compromise between the need of structure compactness and the fulfillment of the assumption that the phase curve computed and/or measured along the broadside direction can be effectively used as the design curve for each element position. In other words, this assumption neglects the effect of the impinging wave incidence angle on the phase of the field reflected by the elements. As it is demonstrated in [30], this condition is valid when the incidence angle on every array element is less than 40 . The designed structure guarantees a maximum incidence angle of about 28 , so satisfying the above requirement. A photograph of the complete reflectarray prototype is illustrated in Fig. 14. The single layers composing the antenna are reported under Fig. 15, following the stratification order given by the adopted configuration (Fig. 1), i.e., the layer containing the 45 radiating patches [Fig. 15(a)], the ground plane with the coupling slots [Fig. 15(b)], and the varactor loaded lines connected to the biasing network [Fig. 15(c)]. The last two layers are printed, respectively, onto the top and the bottom of Diclad 870 copper-clad laminate. Finally, the DAC board driving the antenna elements is illustrated in Fig. 15(d). A more detailed description of the electronic control board can be found in [29] and [31]. IV. SYNTHESIS OF RECONFIGURABLE REFLECTARRAY: NUMERICAL AND EXPERIMENTAL RESULTS The reconfigurability features of the reflectarray prototype are verified by properly changing the biasing voltages across the diodes in order to obtain various shaped radiation patterns. The varactor voltages distributions are computed by adopting a synthesis procedure that receives as input the desired radiation

VENNERI et al.: DESIGN AND VALIDATION OF A RECONFIGURABLE SINGLE VARACTOR-TUNED REFLECTARRAY

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Fig. 16. Radiation pattern for a uniform varactors bias voltages distribution.

When applying the synthesis algorithm to the proposed varactor loaded reflectarray, the set is associated to the reflection coefficient of the unit cell obtained by varying the varactor biasing voltages, while the set is defined by the array patterns lying between a lower bound and an upper bound masks, namely and , with being the free space propagation constant, and the angular direction. The solution of the synthesis problem is computed by the following iterative equation [34]: (1) Fig. 15. Reflectarray prototype: (a) patch layer; (b) ground plane with slots; (c) varactor loaded lines and biasing lines; and (d) DAC board.

features, in terms of pattern behavior, and automatically returns the required excitation phase on each reflectarray element. The effectiveness of the proposed tunable reflectarray structure is then experimentally verified by measuring the radiation pattern for the various computed configurations of varactors bias voltages. Measurements are performed into the anechoic chamber of the University of Calabria, equipped with both farfield and near-field facilities [32], [33]. A double ridged broadband horn, working in the range from 1 up to 18 GHz, is adopted as measurement probe. The reflectarray antenna is placed in its far-field region at a distance of about 5.6 m from the aperture of the probe. The integrated DACs board is directly controlled by a PC, thus imposing the desired bias voltages to each diode. A. Synthesis Algorithm Specification A synthesis algorithm based on the iterative projection method [34], [35] is applied to compute the proper phase distributions on the reflectarray elements so as to compensate for the phase delay in each feed-element path, while obtaining, at the same time, the prescribed field pattern. As described in [34], the synthesis problem is formulated as the finding of an array factor satisfying the prescribed field requirements. It finds the optimum solution within the intersection , where is the set of array factors that fulfill the pattern requirements, while the set specifies the allowable excitation coefficients.

and define the projection operators on the sets where and , respectively. A rigorous and detailed description of (1) can be found in [34]. The measured phase curves (Fig. 11), relating the element reflection phase to the supplied bias voltage, are considered as input design data for the implemented algorithm. The set imposing the radiation pattern behavior, is properly specified for the various cases of single-beam, multibeam, and shaped-beam patterns, as described in the next paragraphs. As outlined above, the realized prototype, having a reduced number of elements in the E-plane, is considered only for illustrative purpose, so the pattern synthesis is uniquely applied in the H-plane. As a starting point, the reflectarray behavior is considered in the absence of a proper phase distribution compensating for the phase delay imposed by the feed. The corresponding theoretical and experimental H-plane patterns, obtained for a uniform voltage supply of 10 V on each varactor diode, are illustrated in Fig. 16. As discussed in the following paragraphs, a proper synthesis of the varactor bias voltages leads to reconfigure the radiation pattern of Fig. 16 into a desired pattern shape. B. Beam-Scanning Operation Mode To verify the beam-scanning operation mode, the synthesis algorithm is applied for obtaining the correct phase distributions giving a main beam pointed to some specific directions in the H-plane. The array grid spacing, in conjunction with the array size, allows us to actively move the radiated mainlobe along all directions ranging from 25 to 25 , without occurring in

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Fig. 17. Synthesized radiation patterns for (H-plane).

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0

10

Fig. 19. Measured co-polar and cross-polar patterns for

0 .

Fig. 20. Measured co-polar and cross-polar patterns for

10 .

Fig. 21. Measured co-polar and cross-polar patterns for

20 .

20

Fig. 18. Measured radiation patterns for different varactors bias voltage distributions (H-plane).

grating lobes appearance. As the H-plane first null beamwidth is equal to 11 , a minimum step of 10 is assumed when driving the varactors voltages, uniquely to avoid overlapping between adjacent beams. An operating frequency equal to 11.25 GHz is considered. Fig. 17 shows the synthesized patterns obtained when is fixed to 0 , 10 , and 20 , respectively. In the same figure are also reported the lower and upper bound masks adopted for the case of a pattern with 0 . The same couple of masks, opportunely shifted of 10 and 20 , is adopted for the synthesis of the other two considered cases. The radiation patterns, measured at the frequency 11.25 GHz (Fig. 18), show a good agreement with the constraints imposed to the main beam positions during the synthesis step. They prove a very good beam-scanning capability for the proposed single-varactor loaded reflectarray configuration. Furthermore, the measured co-polar and cross-polar patterns are compared in each considered case (Figs. 19–21). It can be observed that all cross-polar components are characterized by a lower intensity level (about 23 dB) with respect to the relative co-polar patterns. In the case of a bias voltages distribution giving the broadside main beam (Fig. 19), a frequency gain characterization is also performed, with a measured value of 9.3 dBi obtained at the

Fig. 22. Measured gain versus frequency in the broadside direction for 0 .

design frequency of 11.25 GHz. The complete gain curve versus frequency is reported in Fig. 22. For the beam directions 10 , 20 , relative to the case illustrated in Figs. 20 and 21, a gain value of 9.22 and 6.76 dBi, respectively, is experimentally estimated at the central design frequency of 11.25 GHz.

VENNERI et al.: DESIGN AND VALIDATION OF A RECONFIGURABLE SINGLE VARACTOR-TUNED REFLECTARRAY

Fig. 23. Synthesized multibeam radiation pattern (H-plane).

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Fig. 25. Comparison between measured and synthesized multibeam radiation patterns.

Fig. 24. Synthesized E-field contour plot (normalized values). Fig. 26. Measured multibeam radiation patterns.

C. Multibeam Patterns The adopted synthesis algorithm is modified in order to design a multibeam reflectarray. In particular, the set is properly defined as the union of distinct masks to impose a radiation pattern guaranteeing the simultaneous presence of main lobes. The example reported in this subsection consists of a pattern having two main beams directed along 10 and 25 , respectively. The operating frequency is again fixed to 11.25 GHz. The synthesized pattern and the bound masks adopted in this case are reported in Fig. 23. A contour plot of the synthesized pattern is also illustrated in Fig. 24. Both figures show the presence of two main lobes, which are characterized by a similar intensity level. The varactor bias voltages returned by the synthesis algorithm are adopted to supply the reflectarray elements, thus obtaining the measured radiation pattern illustrated in Fig. 25. A good agreement can be observed between measured and synthesized data, thus confirming the validity of the proposed tunable reflectarray configuration. As a further proof of the ability in pattern reconfigurability, the measurement results obtained for other two synthesized multibeam solutions are reported in Fig. 26. In these cases, the varactor biasing voltages are computed in order to obtain a first

solution characterized by three main beam directed along the directions 25 0 25 , and a second radiation pattern with two main lobes pointing at 10 and 10 , respectively. For the multibeam configuration with two main beams directed along 10 and 25 , a gain estimation is experimentally performed at the central design frequency of 11.25 GHz, obtaining a value of 9.1 and 5 dBi, respectively along 10 and 25 . D. Cosecant Shaped Patterns The fabricated reflectarray antenna is finally adopted for the synthesis of a cosecant shaped pattern at the frequency 11.25 GHz. The lower and upper bound masks prescribed in the synthesis stage are shown in Fig. 27, where the radiation pattern given by the synthesis algorithm is also illustrated. A quite good agreement between the masks and the pattern can be observed. The contour plot of the synthesized pattern is depicted in Fig. 28, while the measured H-plane of Fig. 29 shows a quite good agreement with the theoretical one. A mean gain of 7 dBi in the shaped zone is estimated from measurements at the design frequency of 11.25 GHz.

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on the use of a patch aperture coupled to a varactor loaded microstrip line, has been optimized for a full phase tuning range. Simulated and measured data have been reported to demonstrate the phase tuning ability of the proposed configuration, which has been successful applied to the realization of a 3 15 elements reflectarray antenna operating in the -band. The varactor biasing voltages have been properly synthesized in order to reconfigure the H-plane pattern for a variety of test cases, namely: a main beam scanning within an angular range of about 40 , a set of multibeam radiation patterns and a cosecant beam shape. The experimental validations performed for each test case have demonstrated the effectiveness of the proposed tunable reflectarray configuration. REFERENCES Fig. 27. Synthesized cosecant shaped radiation pattern (H-plane).

Fig. 28. Synthesized E-field contour plot (normalized values).

Fig. 29. Comparison between measured and synthesized cosecant shaped patterns.

V. CONCLUSION A reconfigurable reflectarray element, tuned by a single varactor diode, has been considered for the realization of a reconfigurable reflectarray prototype. The proposed element, based

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VENNERI et al.: DESIGN AND VALIDATION OF A RECONFIGURABLE SINGLE VARACTOR-TUNED REFLECTARRAY

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Francesca Venneri (M’04) received the degree in information technology engineering from the University of Calabria, Italy, in 1998 and the Ph.D. degree in electronic engineering from the University “Mediterranea” of Reggio Calabria in 2002. Currently, she is an Assistant Professor at the University of Calabria. Her research interests focus on microstrip reflectarrays and antenna analysis and synthesis.

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Sandra Costanzo (M’00–SM’11) received the Laurea degree (summa cum laude) in computer engineering from the University of Calabria, Italy, in 1996 and the Ph.D. degree in electronic engineering from the University of Reggio Calabria, Italy, in 2000. Currently, she is an Assistant Professor at the University of Calabria, where she teaches courses on electromagnetic waves propagation, remote sensing, and signal and waves in telecommunications. In the framework of the European School of Antennas, she was a teacher in the ACE (Antenna Centre of Excellence) course Phased Arrays and Reflectarray”, held at the TNO (Netherlands Organisation for Applied Scientific Research), Nederlands, in 2005. Since 1996, she has been involved in many research projects funded by international and national companies. She has authored or coauthored more than 100 contributions in international journals, books, and conferences. Her research interests focus on near-field/far-field techniques, antenna measurement techniques, antenna analysis and synthesis, numerical methods in electromagnetics, millimeter-wave antennas, reflectarrays, synthesis methods for microwave structures, electromagnetic characterization of materials, innovative antennas, and technologies for radar applications. Dr. Costanzo received the Telecom Prize for the Best Laurea Thesis in 1996. She is member of the IEEE South Italy Geoscience and Remote Sensing Chapter, the CNIT (Consorzio Consorzio Nazionale Interuniversitario per le Telecomunicazioni), and the SIEm (Società Italiana di Elettromagnetismo). In 2001, she was the Conference Chair of the Antenna Measurement Techniques Session at the IEEE AP-S International Symposium. In 2008, she was member of Technical Program Committee and Chairm of the session EuMC 34 Application Specific Antenna Designs 2 at the Thirty-Ninth European Microwave Conference. She serves as reviewer for various international journals and is member of the Editorial Board for the Electrical and Electronic Engineering journal, Scientific and Academic Publishing. She is the Editor of the book Microwave Materials Characterization (INTECH, 2012), and Lead Editor of the 2012 International Journal of Antennas and Propagation Special Issue on Reflectarray Antennas: Analysis and Synthesis Techniques.

Giuseppe Di Massa (SM’93) was born in Barano d’Ischia, Naples, Italy, in 1948. He received the Laurea degree in electronic engineering from the University of Naples, Federico II, Naples, Italy, in 1973. From 1978 to 1979, he was Professor of antennas at the University of Naples, Italy. In 1980, he joined the University of Calabria, Italy, as Professor of electromagnetic waves. Since 1985, he served as an Associate Professor, and since 1994 as a Full Professor, at the same university, where he teaches antennas and electromagnetic fields. From 1985 to 1986 he was a Scientific Associate at CERN, Geneva. In 1988, he was Visiting Professor at Brookhaven National Laboratory, Long Island, NY. From 1997 to 2002, he was the Dean of the Department of Elettronica, Informatica and Sistemistica and the President of Programming Committee at University of Calabria. Currently, he is the Chairman of the Telecommunication Engineering Program at the University of Calabria. He has authored/coauthored more than 300 scientific papers, mainly on international scientific journals or proceedings of international conferences. He is the Principal Investigator or Coordinator of many research programs, granted by national and international research organizations, as well as by leading national companies. His main research interests are focused on applied computational electromagnetics, microstrip antennas, microwave integrated circuits, reflectarrays, Gaussian beam solutions, millimeter wave antennas, near-field measurements, electromagnetic characterization of materials, innovative radar antennas and technologies. Dr. Di Massa was the Italian Delegate in the European COST 284 “Innovative Antennas for Emerging Terrestrial and Space-Based Applications from 2002 to 2006 and, from 2007 to 2011, in the COST Action IC0603: Antenna Systems & Sensors for Information Society Technologies (ASSIST). Currently, he is the Italian delegate in the Management Committee of the COST Action IC1102, Versatile Integrated and Signal-aware Technologies for Antennas (VISTA). From 2003 to 2007, he participated in the Network of Excellence Antenna Centre of Excellence (ACE), where he was part of the Governing Board and leader of Work Group WP 1.2-1: Antenna Measurement Services.