Generation, Modulation,and Detection of Signals in ...

Thesis for the degree of doctor of philosophy

Generation, Modulation, and Detection

of Signals in Microwave Photonic Systems

Andreas Wiberg

Photonics Laboratory

Department of Microtechnology and Nanoscience

CHALMERS UNIVERSITY OF TECHNOLOGY

G¨oteborg, Sweden 2008

Generation, Modulation, and Detection of Signals in Microwave Photonic Systems Andreas Wiberg G¨ oteborg, March 2008

c Andreas Wiberg, 2008 � ISBN 978-91-7385-083-4 Doktorsavhandlingar vid Chalmers Tekniska H¨ ogskola Ny serie Nr 2764 ISSN 0346-718X Chalmers University of Technolgy Department of Microtechnology and Nanoscience (MC2) Photonics Laboratory SE-412 96 Gteborg, Sweden Phone: +46 (0) 31 772 1000 ISSN 1652-0769 Technical Report MC2-112

Front cover illustration: A measured electrical spectrum of an optically generated 40 GHz carrier (top). An eye-diagram of a received 2.5 Gb/s data signal (right). A constellation diagram of a demodulated 2.5 Gsymbols/s 16-QAM near-baseband synchronous subcarrier signal (left). An optical spectrum showing the output from a carrier suppressed modulated Mach-Zehnder modulator (bottom).

Printed by Bibliotekets reproservice, Chalmers, G¨ oteborg, February 2008

Generation, Modulation, and Detection

of Signals in Microwave Photonic Systems

Andreas Wiberg

Photonics Laboratory, Department of Microelectronics and Nanoscience

Chalmers University of Technology, Sweden

Abstract

This thesis deals with the use of photonic technology in microwave and millimeterwave applications. The two major parts of this work have been techniques for transmission and signal generation. The transmission of analog microwave sig­ nals over optical fiber is known as radio-over-fiber and utilizes the advantages of the optical fiber in terms of loss, size, weight, cost and immunity to electromag­ netic interference. In this thesis, several techniques are presented for generation and modulation of dispersion tolerant millimeter-wave signals, in order to avoid power fading induced by chromatic dispersion. We have demonstrated systems op­ erating with 40 GHz millimeter-wave signals transmitted over optical fiber up to 44 km with 2.5 Gb/s data, including a short range wireless transmission. Further­ more, multiplexed modulation and simultaneous transmission over optical fiber of microwave and millimeter-wave signals are presented, and all-optical demultiplex­ ing using fiber Bragg gratings are successfully demonstrated. Subcarrier modulation can be used in high bit rate optical communication sys­ tems in order to send multilevel data in a simple manner, considering the optical link as a ”black box” with electrical input and output. A new concept for generat­ ing modulated subcarrier signals using binary digital electronics with up to 16-PSK modulation at 2.5 Gsymbols/s is presented. The performance of our in-house fabricated unitraveling-carrier photodiodes (UTC-PD) and a commercial PIN-PD is compared in the context of an analog link requiring high carrier-to-noise ratio and low distortion. We have found that the benefits of using a UTC-PD is mainly its superior spurious-free dynamic range. The generation of millimeter-wave signals is important for applications, where high frequency local oscillators are used, e.g. in antenna arrays. Using nonlin­ ear characteristics or phenomena of photonic technology, millimeter-waves can be generated which have frequencies several times higher than the original electrical signal. Different techniques for harmonic signal generation are presented including harmonic frequency generation using an optical phase modulator, an optoelectronic oscillator, optical four-wave mixing or chirped pulse mixing. Keywords: Microwave photonics, millimeter-wave photonics, mm-wave generation, photonic frequency multiplication, millimeter-wave communication system, radio over fiber, subcarrier modulation, uni-travelling-carrier photodiode, spurious free dynamic range third-order intercept point, PSK, QAM.

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List of papers This thesis is based on the following appended papers: [A] P.O. Hedekvist, B-E. Olsson, and A. Wiberg, “Microwave Harmonic Frequency Generation Utilizing the Properties of an Optical Phase Modulator,” J. Lightwave Technol., Vol. 22, No. 3, pp. 882–886, March, 2004. [B] A. Wiberg and P.O. Hedekvist, “Photonic Microwave Generator Utilizing Nar­ rowband Brillouin Amplification and a Fiber-Based Oscillator,” Conference on Mi­ crowave and Terahertz Photonics, Proceedings of SPIE, Vol. 5466, pp 148–156, 29­ 30 April, 2004, Strasbourg, France. [C] A. Wiberg, P. Per´ez-Mill´ an, M. V. Andr´es, and P.O. Hedekvist “MicrowavePhotonic Frequency Multiplication Utilizing Optical Four-Wave Mixing and Fiber Bragg Gratings,” J. Lightwave Technol., Vol. 24, No. 1, pp. 329–334, January, 2006. [D] J. Stigwall and A. Wiberg “Tunable Terahertz Signal Generation by Chirped Pulse Photomixing,” IEEE Photon. Technol. Lett., Vol. 19 No. 12, pp. 931–933, 2007. [E] A. Wiberg, P. Per´ez-Mill´ an, M. V. Andr´es, P. A. Andrekson and P.O. Hedekvist, “Fiber Optic 40 GHz mm-wave Link with 2.5 Gb/s Data Transmission,” IEEE Pho­ ton. Technol. Lett., Vol. 17, No. 9, pp. 1938–1940, 2005 [F] P. Per´ez-Mill´ an, A. Wiberg, M. V. Andr´es, and P.O. Hedekvist “Optical Demulti­ plexing of Millimeter-Wave Subcarriers for Wireless Channel Distribution Employing Dual Wavelength FBGs,” Optics Communications, Vol. 275, No. 2, pp. 335–343, 2007. [G] A. Wiberg, B-E Olsson, P.O. Hedekvist and P.A. Andrekson “DispersionTolerant Millimeter-Wave Photonic Link Using Polarization-Dependent Modulation,” J. Lightwave Technol., Vol. 25, No. 10, pp. 2984–2991, October, 2007. [H] A. Wiberg, B-E Olsson, and P.A. Andrekson “Near-Baseband Synchronous Subcarrier Modulation,” submitted to IEEE Photon. Technol. Lett. [I] A. Wiberg, J. Vukusic, H. Sunnerud, and P.A. Andrekson “Linearity and Noise Comparison of Uni-Traveling-Carrier- and PIN-Photodiodes,” submitted to IEEE Photon. Technol. Lett.

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Publications by the author related to this thesis: [J] P.O. Hedekvist, B-E. Olsson, and A. Wiberg, “Microwave Harmonic Frequency Generation Utilizing the Properties of an Optical Phase Modulator,” International Topical Meeting on Microwave Photonics, MWP’03, pp. 193-196, 10­ 12 September, 2003, Budapest, Hungary. [K] P. Per´ez-Mill´ an, A. Wiberg, M. V. Andr´es, and P.O. Hedekvist “38 GHz Microwave Photonic Generation utilizing Four-Wave and Fiber Bragg Gratings,” International Topical Meeting on Microwave Photonics, MWP’04, paper TD-4, pp. 213–216, 4­ 6 October, 2004, Ogunquit, Maine, USA. [L] A. Wiberg, B-E Olsson, P.O. Hedekvist and P.A. Andrekson “Selective Modulation of Optical Carriers Using Polarization Dependent Scheme for Dispersion-Tolerant Millimeter-Wave Photonic Links,” Proceedings 32nd European Conference on Optical Communication (ECOC), Vol. 3, paper We1.6.2, pp. 55–56, 24-28 September, 2006, Cannes, France [M] A. Wiberg, P. Per´ez-Mill´ an, M. V. Andr´es, P.O. Hedekvist and P.A. Andrekson “Evaluation of All-Optical Demultiplexing in Millimeter-Wave Subcarrier-System for Wireless Communication,” International Topical Meeting on Microwave Photonics, MWP’06, paper T2.2, 3-6 October, 2006, Grenoble, France [N] J. Vucusic, H Sunnerud, A. Wiberg, M. Sadeghi, P.A. Andrekson and J. Stake “Fab­ rication and Characterization of InGaAlAs/InP based Uni-Traveling-Carrier Photo­ diodes,” The Joint 31st International Conference on Infrared and Millimeter Waves and 14th International Conference on Terahertz Electronics, pp. 138, 18-22 Septem­ ber, 2006, Shanghai, China [O] J. Stigwall and A. Wiberg “Photonic GHz to THz Tunable Signal Generation by Cirped-Pulse Mixing,” 2006 European ISIS Workshop, Emerging Optical Broadband Technologies, May 29–June 1, 2006, Boppard am Rhein, Germany. [P] P. Per´ez-Mill´ an, A. Wiberg, P.O. Hedekvist, J.L. Cruz and M. V. Andr´es “Dis­ persion Induced Effects of High-Order Optical Sidebands in the Performance of Millimeter-wave Fiber-Optic Links,” Microwave and optical technology letters, Vol. 48, No. 7, pp. 1436–1441, 2006. [Q] A. Wiberg “Millimeter-Wave Photonics: Signal Sources and Transmission Links,” Licentiate Thesis, Chalmers University of Technology, No. MC2-42, 2005.

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Acknowledgement I would like to thank my colleagues in the Fiber optics group, especially Peter Andrekson and Magnus Karlsson. I would also like to thank Per Olof Hedekvist for giving me the opportunity to work in the microwave photonics field. Also thanks to the rest of the group, Henrik Sunnerud, Mathias Westlund and espe­ cially Mats Sk¨old, with their helping hands and good discussions. I would like to thank Bengt-Erik Olsson for always being enthusiastic and open-minded for new ideas and discussions. I would also like to thank Sheila Galt and Arne Alping for introducing me to this area through my master thesis. I would also like to thank the rest of the staff at the Photonics laboratory for creating an inspiring and friendly atmosphere to work in. Special thanks to Pere Per´ez-Mill´ an at the University of Valencia for the good collaboration and his enthusiasm during his two visits at the Photonics laboratory. Finally, I would like to thank my wife Sara, my son William, my brother Gustav, my father Kjell and the rest of my family for all their love and support. This work has been financially supported by the Swedish research council (VR) and the Swedish Foundation for Strategic Research (SSF). It has also been part of the European Networks of Excellence on Broadband Fiber Radio Techniques and its Integration Technologies (NEFERTITI) and on Infrastructures for Broad­ band Access in Wireless/Photonics and Integration Strengths in Europe (ISIS). Key equipment used in the experiments in this thesis was funded by the Knut and Alice Wallenberg Foundation. Travel grants for visiting conferences have been received from Kungliga och Hvitfeldtska stiftelsen and Stiftelsen Wilhelm och Mar­ tina Lundgrens Vetenskapsfond. Andreas Wiberg G¨ oteborg March 2008

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Abbreviations used in the text

AC AFPM ASE AWG BER BERT BRGC BS CATV CNR CS CW DAC DC DCF DFB DGD DWDM EAM EAT EDFA EOM FBG FWM GPS GSM HD2 HD3 HFC HNLF IEEE IF IMD3 IP3 LASER

Alternating current Asymetric Fabry-Perot modulator Amplified spontaneous emission Arrayed waveguide grating Bit-error rate Bit-error test set binary reflected Gray code Base station Common antenna television Carrier-to-noise ratio Central station Continuous wave Digital-to-analog converter Direct current Dispersion compensating fiber Distributed feedback Differential group delay Dense wavelength division multiplexing Electro-absorbtion modulator Electro-absorbtion transceiver Erbium-doped fiber amplifier External optical modulator Fiber Bragg grating Four-wave mixing Global positioning system Global system for mobile communications Second order harmonic distortion Third order harmonic distortion Hybrid fiber/coax Highly nonlinear fiber Institute of Electrical and Electronics Engineers, Inc. Intermediate frequency Third order intermodulation distortion Third order intercept point Light amplification by stimulated emission of radiation vii

LO MMF mmWP MU MWP MZI MZM OEO OFLL OIP3 OIPLL OOK OPLL PAM PBS PCF PD PDA-PD PM PMD PRBS PSK QAM RADAR RF RIN RoF SBS SCM SFDR SMF SNR SOA SPM SSB TW UMTS UTC-PD VCSEL WDM WG Wi-Fi WLAN XPM

Local oscillator Multi mode fiber Millimeter-wave photonics Mobile unit Microwave photonics Mach-Zehnder interferometer Mach-Zehnder modulator Optoelectronic oscillator Optical frequency locked loop Third order output intercept point Optical injection phase-locked loop On-Off keying Optical phase-locked loop Puls amplitude modualtion Polarization beam splitter Photonic crystal fiber Photodiode Partially depleted-absorber photodiode Phase Modulator Polarization-mode dispersion Pseudo random bit sequence Phase shift keying Quadrature amplitude modulation Radio detection and ranging Radio frequency Relative intensity noise Radio over fiber Stimulated Brillouin scattering Subcarrier multiplexing Spurious-free dynamic range Standard single-mode fiber Signal-to-noise ratio Semiconductor optical amplifier Self-phase modulation Single sideband Travelling wave Universal mobile telecommunications system uni-traveling-carrier photo diode Vertical cavity surface emitting laser Wavelength-division multiplexing Waveguide Wireless fidelity Wireless local area network Cross-phase modulation

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Table of contents Abstract

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List of papers

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Acknowledgement

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Abbreviations used in the text

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1 Introduction 1.1 Microwaves Photonics . . . . . . . . . . . . . . . . . 1.2 Microwave Photonic Applications . . . . . . . . . . . 1.2.1 Radio over Fiber . . . . . . . . . . . . . . . . 1.2.2 Microwave Photonic Communication Systems 1.2.3 Signal Generation and Analog Applications . 1.3 Motivation and Outline . . . . . . . . . . . . . . . .

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2 Photonic Technology and Components 2.1 Optical Sources . . . . . . . . . . . . . . . . . . . . . 2.2 Optical Modulators . . . . . . . . . . . . . . . . . . . 2.2.1 The Optical Phase Modulator . . . . . . . . . 2.2.2 The Mach-Zehnder Modulator . . . . . . . . 2.3 Optical Fiber Propagation . . . . . . . . . . . . . . . 2.3.1 Chromatic Dispersion . . . . . . . . . . . . . 2.3.2 Polarization Mode Dispersion . . . . . . . . . 2.3.3 Nonlinear Phenomena in Optical Fiber . . . . 2.4 Photo Detection . . . . . . . . . . . . . . . . . . . . 2.4.1 Photo Detectors . . . . . . . . . . . . . . . . 2.4.2 Properties of Photomixing [Paper G] . . . . . 2.5 Performance and Analysis . . . . . . . . . . . . . . . 2.5.1 Link Gain . . . . . . . . . . . . . . . . . . . . 2.5.2 Noise Sources . . . . . . . . . . . . . . . . . . 2.5.3 Distortion and Spurious-Free Dynamic Range 2.6 All-Optical Microwave Filtering . . . . . . . . . . . . 2.6.1 Fiber Bragg Gratings [Papers C, E & F] . . .

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2.6.2 Selective Brillouin Amplification [Paper B]

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3 Photonic microwave and millimeter-wave generation 31

3.1 Signal Generation with Dual Sidebands . . . . . . . . . . . . . . . . 31

3.2 Single Sideband Modulation . . . . . . . . . . . . . . . . . . . . . . . 32

3.3 Harmonic Frequency Generation . . . . . . . . . . . . . . . . . . . . 32

3.3.1 Mach-Zehnder Modulator Based Harmonic Generation [Papers C,

E, F & G] . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 33

3.3.2 Harmonic Frequency Generation with a Phase Modulator

[Papers A & B] . . . . . . . . . . . . . . . . . . . . . . . . . . 35

3.3.3 Harmonic Generation using Nonlinear Phenomena in Optical

Fibers [Paper C] . . . . . . . . . . . . . . . . . . . . . . . . . 37

3.3.4 Signal Generation with Chirped Pulse Photomixing [Paper D] 38

3.3.5 Mode-Locked Lasers and Frequency Comb Generation . . . . 39

3.4 Heterodyne Generation . . . . . . . . . . . . . . . . . . . . . . . . . 40

3.4.1 Dual Lasers Techniques . . . . . . . . . . . . . . . . . . . . . 40

3.4.2 Dual Wavelength Lasers . . . . . . . . . . . . . . . . . . . . . 41

3.5 Opto-electronic Oscillator . . . . . . . . . . . . . . . . . . . . . . . . 41

3.5.1 Opto-Electronic Oscillator with Harmonic Output [Paper B] 43

4 Microwave and millimeter-wave communication systems 45

4.1 Modulation Formats . . . . . . . . . . . . . . . . . . . . . . . . . . . 45

4.1.1 Performance Measures . . . . . . . . . . . . . . . . . . . . . . 47

4.2 Subcarrier Transmission . . . . . . . . . . . . . . . . . . . . . . . . . 48

4.2.1 Near-Baseband Synchronous Subcarrier Modulation [Paper H] 50

4.2.2 Binary Digital Electronic Near-Baseband Synchronous Sub-

carrier Signal Generation [Paper H] . . . . . . . . . . . . . . 51

4.3 Millimeter-Wave Photonic Links . . . . . . . . . . . . . . . . . . . . 54

4.3.1 Remote Local Oscillator Delivery to the Base Station . . . . 54

4.3.2 Directly Modulated Millimeter-Wave Links . . . . . . . . . . 55

4.3.3 Dispersion Tolerant Millimeter-Wave Photonic Links [Papers E,

F & G] . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 56

4.3.4 Full Duplex Millimeter-Wave Systems . . . . . . . . . . . . . 58

4.3.5 Multiplexed Millimeter-Wave Signal Delivery [Paper F] . . . 60

5 Conclusion and outlook

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6 Summary of papers

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Appendices Appendix A Appendix B Appendix C Appendix D

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References

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Papers A– I

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Chapter 1

Introduction n our daily life we are constantly in contact with, and influenced by, science

and technology offsprings, which have their origin in the former part of the 20 Icentury. Today its seems natural to receive radio and TV broadcasts, and to th

have direct contact with each other, whether it’s next door or at the other side of the earth, with wired and mobile phone, or through the Internet or with e-mail. However, one cannot forget that it is scientific work and technology advancement, that are behind this revolutionary change of our lives. This has taken us from the telephone wires in the beginning of the last century, through the wireless evolution of radio and satellite communication, to the world wide networks of fiber optic cables that connects the continents today. The area of microwave technology took a huge step forward in the middle of the 20th century, as a consequence of the World War II. Military systems were developed for wireless communication, and with that improved antennas. New technology for surveillance was also developed, better known as RAdio Detection And Ranging (RADAR). A basic part of microwave technology at that time was the electron tubes, which were space and power consuming. These were gradually replaced as a result of the development of semiconductor devices, such as the transistor, which were smaller, more reliable and power efficient. Technologies that originate from military applications have later on evolved into civilian applications which are part of everyday life, for example the mobile telephone, the microwave oven, GPS or the Internet. A new era in communication begun with the invention of the LASER in 1960 [1], which could be used as a coherent light source. A great effort was then made to find suitable transmission media for the light. In 1966 the optical fiber was devel­ oped, which could guide the light in a manner similar to electrons in copper wires. The first optical fibers had a huge loss of about 1000 dB/km, but a large step in the development of glass fibers was made in 1970 when the losses were decreased dramatically to 7 dB/km [2]. Around the same time, GaAs semiconductor lasers, continuously operating at room temperature, were demonstrated [3]. This simulta­ neous availability of a compact coherent light sources and low-loss optical fibers led to a fast development of fiber-optic communication systems. With the development of the fiber amplifier during the late 1980’s [4], the capacity could be increased and

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cost reduced further in the fiber optic networks by using wavelength division mul­ tiplexing (WDM). Since the 1970’s the progress has been remarkable, as evidenced by the many orders of magnitude increase in transmission capacity. Today’s state of the art research systems have demonstrated capacities of over 1 Tbit/s for a single optical channel [5,6], and over 20 Tbit/s for wavelength division multiplexed channels [7, 8].

1.1

Microwaves Photonics

The area of microwave photonics (MWP) has evolved in parallel to the develop­ ment of optical communication, as optical components have become available at microwave frequencies and beyond. We refer to [9–11] for the definition of MWP as the study of photonic devices operating at microwave frequencies and their applica­ tions in microwave and optical systems. The interdisciplinary field of MWP covers research within a frequency span from MHz to THz. The flexibility of microwave technology in combination with the low loss, large bandwidth and the insensitivity to electromagnetic disturbances of photonics, create a solid foundation for a vast number of MWP applications. These applications span from common antenna TV (CATV), to antenna remoting of communication systems and radars and to signal generation and signal processing. To clarify the nomenclature that is used in this thesis, microwave photonics (MWP) is used for the research field and, the term millimeter-wave photonics (mmWP) refers to operation at frequencies higher than 30 GHz. RF-photonics is regarded as the technology for transmitting radio signals over optical fiber and the TeraHertz-range refers to frequencies between 0.1 and 3 THz.

RF-input

Electrical transmission medium

RF-output

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RF-input

Electrical to optical converter

Optical to electrical ccnverter

RF-output

Optical transmission medium (b)

Figure 1.1: Comparison of RF transmission in electrical and microwave photonic link.

The microwave photonic link is one of the basic functionalities of MWP. Fig. 1.1 illustrates a conventional microwave link and microwaves sent over an equivalent

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photonic microwave link. A single or multiple analog microwave carriers could be transported over an optical fiber, in contrast to optical digital communication where baseband data is transmitted. However, the use of optical transmission re­ quires the signal to be converted from the electrical to the optical domain at the transmitter, and the optical domain to the electrical at the receiver side. In the fiber, the microwave signals are transported as intensity modulation on top of the optical carrier, as illustrated in Fig. 1.1. The objective of a microwave photonic link is to transport the electrical signal without distorting it, and thus to be trans­ parent to various signal formats. Depending on the application, the radio signal can be analog or digital data, or a pure carrier in form of a distributed local oscil­ lator (LO). However, apart from data transmission in microwave links, additional functions can be performed in the optical domain. For example, frequency up- and down-conversion (mixing) and filtering can be performed optically, to simplify the electrical design, which will be discussed further in the following chapters. The fundamental advantages of a MWP link compared to an electrical mi­ crowave link, where the signal is transported in a coaxial cable or a waveguide, are low loss and modulated frequency independent attenuation. Furthermore, it is in­ sensitive to electromagnetic interference, has low weight, a small size and low cost. If a comparison of weight and attenuation is made of a MWP link and a coaxial cable, it is clearly in favor for the MWP link, since the characteristics of the optical fiber is 1.7 kg/km and 0.5 dB/km, compared to 576 kg/km and 360 dB/km for the coaxial cable (at 2 GHz) [12].

1.2

Microwave Photonic Applications

One of the most important applications of MWP are communication, including radio-over-fiber systems, wireless bidirectional communication and broadcasting. Other important applications are high performance microwave and mm-wave signal generation and distribution of such signals, e.g. in RADAR systems or in antenna arrays. Depending on the application different requirements will be dominant and the feasibility of photonic solutions are determined by different aspects.

1.2.1

Radio over Fiber

The concept of modulating radio frequency (RF) signals on to an optical carrier for distribution over a fiber network is known as radio-over-fiber (RoF)1 . The typical RoF system connects a central station (CS) to a base station (BS), via an optical fiber, as illustrated in Fig. 1.2. Alternatively, an intermediate frequency (IF) signal or a digitized analog signal can be transmitted, depending on the application. The latter one is equal to conventional digital optical communication. The difference between RF and IF signals is only the carrier frequency of the wireless transmis­ 1 In

literature it is also called ”radio on the fiber”, ”radio on fiber”, ”hybrid fiber radio”, ”hybrid fiber/coax”, ”hybrid-fiber-radio” and ”fiber radio access”

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sion, where the IF signal has a much lower frequency than the RF and must be upconverted at the BS.

O/E

Central Station

Base Station

Baseband

IF

Opt.

Opt.

Baseband Modulation

Intermediate Frequency (IF) Modulation

RF

Opt.

RF Modulation

Figure 1.2: Illustration of three ways for signal transport over fiber, Baseband over fiber, IF over fiber and RF over fiber.

An early application of RoF was in CATV networks, where subcarrier multi­ plexed (SCM) TV-signals were sent over optical fiber. This was known as hybrid fiber/coax (HFC), since it has come to replace electrical coaxial cables [13]. Fur­ thermore, digitally modulated subcarriers could also be used in broadband data transmission. Subcarrier modulation is attractive compared to direct optical modu­ lation, since the phase of the modulated subcarrier is preserved after photo detec­ tion. The phase of an optical signal is lost at detection, because the generated photo current is proportional to the intensity of the light and not the field. This is further discussed in chapters 2 and 4. In mm-wave system designs, it could be beneficial to move functionality from the electrical domain to the optical. Some features of optical components can be used, such as the nonlinear properties of the optical modulators, the square-law photo detection in photodiodes or optical filtering. However, the requirement of the design od the optical transmission link is changed since chromatic dispersion affects the carrier more as frequency increases. Therefore, the choice of technical solution is dependent on the carrier frequency and the transmission distance. This will be addressed later in chapters 2 and 3.

1.2.2

Microwave Photonic Communication Systems

The use of broadband wireless access is expected to grow rapidly over the next decades. Today the radio spectrum is divided into narrow bands and those bands are strictly allocated for various applications, leaving only limited bandwidth for communication. As the number of mobile phones and WLAN equipped devices increases the lower part of the radio spectrum is becoming overcrowded. To meet

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the demand for both high bit rates and wireless connectivity, higher frequency bands, particulary in the mm-wave region, have been considered. Higher carrier frequencies and higher bit rates increase the requirements of the hardware as well as the overall system design. One of the advantages of using RoF systems is that they can give a flexible approach for multiple antennas connecting by using a centralized architecture. By allowing centralized control in the central station (CS), simplified base stations (BS) could be distributed closer to the users, see Fig. 1.3. By having many BSs covering large geographical areas divided in to cells, the centralized control could provide dynamic channel allocation in order to follow fluctuations in traffic load. One single CS can be connected to a core network by fiber or wireless, and distribute the information to the BSs. The system complexity may thus be moved to the CS, enabling the BSs to be very simple. The latter can hence be low-functionality, lowcost, small size, lightweight and reliable, which also lowers the maintenance and installation cost. A key benefit is the future-proof capability of such system if it is properly designed, since it could have multiple serviced and standards working in parallel, such as mobile telephony (GSM, UMTS) and WLAN, and future services and applications.

Mobile Unit

CS

Central Station

Base Station

Fiber Optic Transmission

Central Station

Figure 1.3: Outdoor cellular coverage and in-door coverage with distributed antenna systems.

The design of RoF systems for wireless communication will differ depending on which carrier frequency that is used, due to effects of fiber transmission and required bandwidth of components. The chromatic dispersion in optical fibers is not a significant limitation at frequencies under 10 GHz over modest reaches. Cellular and WLAN signals ( 430 mA, with very high saturation current [84] and RF power out of over 24 dBm at 2 GHz has been reported [83].

2.4.2

Properties of Photomixing [Paper G]

The detection process in the photodetector gives an output signal containing a DC component proportional to the average optical power. The squaring process of the optical field will also give mixing products between optical field components within the bandwidth of the photodetector. For example, two CW-laser sources with frequency separation f , will, when detected, create an amplitude modulated electrical carrier equal to the frequency separation of the lasers, as well as a DC component, which is illustrated in Fig. 2.9. This is known as photo mixing or heterodyne detection. A fundamental property is that the detection process is 100% polarization dependent, i.e. the polarization states must be parallel for efficient mixing. The consequence if the polarization states are not aligned is a decreased power in the AC-components. However, the power in the DC-components will remain constant. The polarization dependance can be used for intensity modulation of the heterodyne created microwave signal, which was used in Paper G.

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f

λ1

0

λ2

f

Figure 2.9: Illustration of photodetection of two optical carriers, i.e heterodyne detection.

2.5

Performance and Analysis

A complete photonic RF-link involves the conversion from the electrical domain into the optical domain, the transportation over optical fiber, and then conversion back to electrical again. At low microwave frequencies the electrical to optical (E/O) conversion can be performed by directly modulating a laser diode (LD), as illustrated in Fig. 2.10(a), but for mm-wave frequencies the performance of an external optical modulator, e.g. an electro-absorbtion modulator (EAM) or a MachZehnder modulator (MZM), is needed as illustrated Fig. 2.10(b). A photo-detector is used as the optical to electrical (O/E) converter or receiver. To avoid undesired power losses at the carrier frequency, the detector is required to have at least that frequency bandwidth. (a) Sd(f),Rd

Sl(f),Rl RF in

PD

LD

Pin

RF out Pout

Losses, a

(b) Plaser Laser RF in

Sd(f),Rd

Sm(f,Plaser),Rm

PD

EOM Losses, a

RF out Pout

Pin

Figure 2.10: Illustration of photonic RF-wave link with its parts. The electrical to optical con­ version is performed in (a) by a directly modulated laser diode (LD), and in (b) by an external optical modulator.

There are several measures of the performance of a photonic RF link. Since it is similar to a microwave link, it is evaluated in the same manner. A photonic RF-link could thereby be treated as a ”black box” with RF in and RF out. The

24

linearity is especially important in analog applications in order to keep the signal as unaffected as possible. Additional noise, such as relative intensity noise from the laser, can degrade the noise figure. Furthermore, nonlinearities of the modulator can reduce the spurious free dynamic range (see Sec. 2.5.3). Therefore important performance measures are: the RF gain and frequency response, the noise figure (NF) and the spurious free dynamic range (SFDR).

2.5.1

Link Gain

The linear RF gain of a link, g(f ), is defined as the ratio between the RF power, pout , at frequency f delivered to a matched load at the photo detector output and the RF power at the input, pin , transmitted from the RF source. An illustration of such a link is shown in Fig. 2.10. The RF gain is frequency dependent and generally the frequency response has the characteristic of a low pass filter, i.e. the gain decreases with higher frequency. For a link with a directly modulated LD the RF gain is expressed as [85] g=

Rd pout = sl (f )2 α2 sd (f )2 , pin Rl

(2.16)

and with an external optical modulator g=

pout Rd = sm (f, Plaser )2 α2 sd (f )2 , pin Rm

(2.17)

where Rd , Rl , and Rm are the detector load resistance, the effective laser resistance, and the modulator resistance, respectively. The equations show that the link gain depends quadratically on the slope efficiencies of the laser, sl (defined in Eq. 2.1), and the slope efficiency of the modulator, sm (defined in Eq. 2.5), the detector slope efficiency, sd (equal to the responsivity defined in Eq. 2.14, and the optical losses, α. The latter include coupling losses and fiber transmission loss. The slope efficiency of the laser depends on characteristics of the laser and the modulator slope efficiency depends on which modulator is used, but also on the laser output power, Plaser . High link gain can therefore be achieved with high laser power and with a high responsivity in the detector. To increase link gain, both the directly modulated and the external modulated link configuration can include optical amplifiers to amplify the optical signal. However, at very high optical power the photo-diode will be saturated, and undesired nonlinear phenomena may occur, as discussed in Sec.2.3.3.

2.5.2

Noise Sources

In an unamplified optical link there are three fundamental noise mechanisms: ther­ mal noise, shot noise and relative intensity noise. Noise will give rise to a random fluctuation of the photo current. The different noise contributors are independent random processes which can be approximated with Gaussian statistics [48]. It is

25

therefore convenient to use the variance of noise induced current fluctuations, since the sum of the different noise sources then simply is achieved by summarizing of the independent sources. Thermal noise3 originates from the random motions of the carriers moving in a conductor, which give rise to a fluctuation of the current. The thermal noise variance can be expressed as σT2 =

4kb T Δf RL

(2.18)

where kb is the Boltzmann constant, T is the absolute temperature and Δf the bandwidth. Shot noise is a quantum noise, which is caused by the quantum nature of light and the unpredictability of electron-hole pair generation in the photo detector. The shot noise variance can be expressed as the average photocurrent and the dark current σs2 = 2q(ID + Id )Δf

(2.19)

where ID is the average photocurrent, Id is the dark current, q the electron charge and Δf the bandwidth. Noise is present in the laser source in form of random fluctuations of the out­ put intensity. The mechanisms behind this fluctuation are spontaneous emission and random electron-hole pair recombination. The intensity fluctuation is often expressed in relative spectral density terms as relative intensity noise (RIN) [13], expressed in dB/Hz. The RIN varies with the output power of the laser and with frequency. However, it can be assumed to be constant over a small bandwidth. The RIN variance is written as 2 2 = RIN ID Δf, σRIN

(2.20)

note that thermal noise is not dependent on the photo current, while shot noise and RIN are. The carrier-to-noise ratio (CNR) is defined as the average signal power divided by the noise power CNR =

(mId )2 (mId )2 = 2 , 2 2 σT + σs2 + σRIN σtot

(2.21)

where m is the modulation depth defined as (Id,max − Id,min )/(Id,max + Id,min ). It is apparent that the dominating noise sources will change with photo current. Optical amplifiers, which are used to amplify the optical signal, are also con­ tributing with extra noise. The most common optical amplifier around 1550 nm is the Erbium doped fiber amplifier (EDFA). The output of the EDFA will contain the amplified signal as well as amplified spontaneous emission (ASE) generated in 3 Thermal

noise is also called Johnson noise or Nyquist noise

26

the amplifier. Upon detection, noise beating products will be created by beating between the signal and the ASE noise and noise beating with itself, which are called signal-spontaneous (sig-sp) beat noise and spontaneous-spontaneous (sp-sp) beat noise [48], and can be expressed as 2 σsig−sp 2 σsp−sp

= =

2�G2 PS qηFn Δf 2

2(ηqGFn ) Δνopt Δf

(2.22) (2.23)

where PS is the average optical signal power, G and Fn is the optical amplifier gain and noise figure, η detector quantum efficiency, Δνopt is the optical noise bandwidth, � is the responsivity. Since the signal is amplified, Eq. 2.19 and Eq. 2.20 can be modified by replacing ID with G�PS . By having a narrow optical filter, i.e. small Δνopt , the spontaneous-spontaneous beat noise can be substantially reduced.

2.5.3

Distortion and Spurious-Free Dynamic Range

The nonlinearity-induced signal distortion of a link is a limiting factor in some ana­ log applications, for example multi carrier transmission. Since the transfer function of a device or a link usually is not completely linear, harmonics of the fundamental component and intermodulation products between fundamental frequency compo­ nents will arise with increasing input power. The power of the harmonics and intermodulation products will grow faster with increased input power than the fundamental frequency, and when the power of one of these products increases over the noise floor, it will be the dominating source of distortion. A two tone measurement is the most common measure of the linear performance of a component or a link. The measurement is performed by sending two frequencies (f1 and f2 ) through the system with increasing power and study output frequency components. The most important nonlinear components are the second (HD2) and third (HD3) harmonic distortion and the third order intermodulation distortion (IMD3), illustrated in Fig. 2.11.(a) and (b). The harmonic distortions appear on integer multiples of the fundamental frequency, where HD2 and HD3 are the first. The IMD3 components appear at 2f1 − f2 and 2f2 − f1 , and are the cross-products of two fundamental frequencies, f1 and f2 . For a weakly modulated system, one can assume that the fundamental frequency components increases linearly with modulation power and that the IMD3 components increases cubically with input power [38]. From a measurement of the fundamental and the IMD3 frequency components, two lines can be extrapolated in a log-log plot and the crossing of the lines is called third order intercept point (IP3), marked in Fig. 2.11.(c). The intercept point is usually much higher than the practicable operation limit of the system, but gives a measure of the linearity of system. High values of IP3 indicate that the nonlinear components are suppressed, i.e. the system is very linear. An important measure is the spurious free dynamic range (SFDR), which is defined as the ratio between the power of the fundamental components and the

27

distortion products at the input power, where the distortion reaches the noise floor. The definition for the SFDR when the IMD3 is the dominating distortion, is illustrated in Fig. 2.11(c). The IMD3 is important since this distortion appears in-band and not at a harmonic frequency of the carrier. The unit for the third order intermodulation distortion SFDR is dB·Hz2/3 and could be calculated using [38] SFDR =

2 (OIP3 − Noise(1 Hz)) , 3

(2.24)

where OIP3 and Noise are in dBm and dBm/Hz, respectively. Fundamental HD2 HD3 f1

2f1

3f1

(a) Fundamental IMD3

IMD3

2f1-f2 f1

f2

Output power (log. scale)

IP3

Fundamental

SFDR

IMD3 Noise floor

Input power (log. scale)

2f2-f1

(b)

(c)

Figure 2.11: Illustration of (a) the second (HD2) and third (HD3) harmonic distortion and (b) the the third order intermodulation distortion (IMD3). (c) The output power of fundamental frequency,the IMD3 component and the noise floor as function of input power.

2.6

All-Optical Microwave Filtering

Microwave filtering is possible to achieve by tailoring the properties of the optical link in order to manipulate the optical frequency components of the microwave signal. Therefore, it is important to know the output optical field components. The transmission in the fiber itself will act as a filter since the chromatic dispersion (see sec2.3.1) will change the relative phase of the transmitted components. By carefully tailoring the length of the fiber a specific phase shift of the components is achieved and upon photo detection [86]. The periodic transfer characteristics with respect to frequency of a Mach-Zehnder interferometer (MZI) have been proposed for filtering. A more advanced filter design is created if several MZIs are cascaded and having carefully set time delay between them [87].

28

Other methods of filtering in the optical domain is to use fiber Bragg gratings or to use the narrowness of the Brillouin amplification to amplify selected spectral components.

2.6.1

Fiber Bragg Gratings [Papers C, E & F]

Bragg gratings are widely used in optical systems. A Bragg grating is a periodic perturbation of the medium which is guiding the light, e.g. an optical waveguide or an optical fiber. The perturbation is usually a periodic variation of the refractive index of the medium [88]. Depending on how this periodic variation is designed, custom-made Bragg gratings can be fabricated.

0 Transmitted power (dB)

Transmitted power (dB)

0

-5

-10 (a) -15 -60

-40

-20

0

20

40

60

-10

-20

-60

Relative frequency (GHz)

(b)

-30 -40

-20

0

20

40

60

Relative frequency (GHz)

Figure 2.12: Fiber Bragg gratings used in Paper C. (a) A dual overwritten FBG, and (b) a notch FBG filter.

Fiber based Bragg gratings are attractive since they could be used for a variety of applications such as filtering, with e.g. a uniform grating, and chromatic disper­ sion compensation, with a chirped grating. Their main advantages are their low loss, simple splicing to standard fibers, polarization insensitivity, low temperature coefficient and simple packaging. The gratings are fabricated by exposing special photosensitive optical fiber, e.g. a fiber doped with Germanium, to ultraviolet light. The exposure of ultraviolet light induces small permanent refractive index changes in the fiber which become the grating. Fiber Bragg gratings (FBGs) have been used in several papers in this thesis. they were used as filters in Papers C, E and F, both as notch filters and as dual pass-band filter. Examples of transfer characteristics of these filters used are shown in Fig. 2.12.

2.6.2

Selective Brillouin Amplification [Paper B]

As mentioned section 2.3.3, stimulated Brillouin scattering can be used for very narrow band amplification and has been shown for selective carrier amplification [59,60]. This can be used as an inverse filter by using very selective amplification to enhance specific spectral components, e.g. of one sideband [89]. The amplification

29

is done by sending one or more pump wavelengths propagating in the opposite direction to the signal signal in the fiber. At the output an optical circulator is used in order to direct the outgoing amplified signal from the incoming pump light with as low loss as possible. Two pump lasers that filter out two separate frequency components for the harmonic signal generation was used in Paper B, see Fig. 2.13 and by Schneider et.al. in [90]. A comparison of selective Brillouin amplification and EDFA amplified mm-wave signal generation has shown similar performance with a major difference being that the EDFA amplifies all frequency components, contrary to the selective amplification of the Brillouin amplifier [91]. 12

10

Lower sideband

Optical carrier

Upper sideband Optical frequency

νs

(b)

BS

Pump 1

BS

νB,2

νp,2

(c)

PC DFB 1

3dB

Optical Input

PC DFB 2 10.5 km DSF

2 -30 -20 -10 0 10 20 30 Frequency relative f0 (GHz) (d) 14

EDFA Optical Output

Normalized power

νp,1

4 0

Pump 2 Optical frequency

νB,1

6

12 10 8 6 4 2

Normalized power

(a)

8

Normalized power

14

12

Normalized power

14

10 8 6 4 2 0 -30 -20 -10 0 10 20 30 Frequency relative f0 (GHz) (e) 14 12 10 8 6 4 2

0

0

-30 -20 -10 0 10 20 30 Frequency relative f0 (GHz) (f)

-30 -20 -10 0 10 20 30 Frequency relative f0 (GHz) (g)

Figure 2.13: Schematic spectra of the signal (a) and the selective Brillouin amplification pumps(b). (c) Experimental setup dual selective Brillouin amplification. (d) Spectrum of unamplified signal. (e)-(g) Spectra of amplification of different spectral components of the signal.

30

Chapter 3

Photonic microwave and millimeter-wave generation

T

his chapter presents different techniques for micro- and mm-wave generation, which is important in analog application such as LO generation and refer­ ence signal distribution. Photonic mm-wave technology enables new possibilities, for instance harmonic generation and oscillators that use optical fiber as a delay line. Depending on the application, carrier frequency and transmission distance, the advantages and disadvantages of different techniques have to be taken under consideration, which is the focus of this chapter.

3.1

Signal Generation with Dual Sidebands

Modulating a microwave or a mm-wave frequency on an optical carrier with a di­ rectly modulated laser, or with an external optical modulator (EOM), is attractive since it is very simple and straight-forward. The frequency source is an electri­ cal signal generator and the modulated frequency output will be the same as the frequency source. The modulation will appear as new frequency components in upper and lower sidebands of the optical carrier, see Fig. 3.1. Directly modulated laser diodes have a drawback in the bandwidth limitation which is generally below mm-wave frequencies. However, lasers with direct modulation bandwidths up to 40 GHz have been reported [92]. An advantage of using external modulators is the higher bandwidth available, using both Mach-Zehnder modulators (MZM) [93] and electro-absorption modulators (EAM) [94, 95] are useful. EAMs are promising for very high frequency mm-wave generation (> 100 GHz), since estimated band­ widths of 95 GHz has been reported [96], whilst showing potential for even larger bandwidths. An obvious drawback of modulation generating dual sidebands is apparent if the signal is propagated in a dispersive optical fiber, which was discussed in section 2.3.1. The influence of chromatic dispersion can be counteracted by dispersion compensating fiber (DCF), which can be added to the link to cancel the dispersion corresponding for a specific distance. Another approach to cancel the dispersion is

31

RF out PD

EOM

Laser

f

f

El.

LO Opt.

Figure 3.1: Illustration of dual sideband modulation and transportation through optical fiber.

to use a chirped fiber Bragg grating together with an optical circulator [97], which also cancels the dispersion for a specific distance.

3.2

Single Sideband Modulation

The signal power fading due to chromatic dispersion can be overcome using the single sideband (SSB) technique. SSB modulation can be achieved with a dual-drive MZM [98], by driving both arms with the modulation signal. One of the arms must be fed with a π/2 phase-shift and the resulting output is SSB modulated, as shown in Fig. 3.2. This technique requires a bandwidth of the modulator in the order of the mm-wave and has low modulation efficiency at mm-wave frequencies. This results in weakly modulated signal onto the optical carrier, which leads to a relatively large carrier-to-sideband ratio and low optical modulation depth. This can be overcome by attenuating the optical carrier, and using an optical amplifier [99]. The optimum is achieved when carrier and sideband are equal, as shown in Appendix C. SSB modulation can also be generated from a dual sideband signal, where one of the sidebands is removed by an optical notch filter. This filter could for example be a fiber Bragg grating [100].

3.3

Harmonic Frequency Generation

Most photonic signal generation techniques use an electrical oscillator as the fun­ damental frequency source. In applications requiring high frequencies, up in the mm-wave and THz-range, harmonic frequency generation can be used to gener­ ate the required frequency. This frequency could be several times higher than the fundamental frequency. Photonic technology can be applied to generate such high frequencies, based on an electrical source which is multiplied which results in harmonic frequency output. There are many techniques that could be used for this, and some of them will be discussed in the following sections. Furthermore, harmonic frequency generation can also be used to get a frequency output which is higher than the bandwidth of the device used, e.g. the modulator. This could decrease the bandwidth requirement

32

PC

MZM

Laser Bias

π/2

/

Vπ 2 ff

f

Opt.

Figure 3.2: Illustration of single sideband modualtion (SSB) modulation using a dual drive MZM setup.

of components in the system, which usually is synonymous with high cost.

3.3.1 Mach-Zehnder Modulator Based Harmonic Generation [Papers C, E, F & G] The well known non-linear periodic transfer characteristics of the MZM (see sec­ tion 2.2.2), could be used for harmonic frequency generation. The preferred output is a single harmonic frequency, why the bias point and modulation amplitude must be carefully set. By studying Eq. 2.7, we note that odd and even order harmonics can be canceled out by appropriate bias, and also that the amplitude of different harmonics are controlled be the amplitude of the modulation signal. Frequency doubling using the MZM is a well known technique and presented in e.g. [46, 101]. If the MZM is biased for minimum transmission (ε = ±1, 3, 5 . . . in Eq. 2.3), and driven with a sine-wave with frequency f /2, the output will have the double frequency, i.e. f , as shown in Fig. 3.3(a). The spectral output from the modulator is understood by Eq. 2.7. From the bias setting, the component at ν0 , the optical carrier, will be suppressed as well as all even components. Two strong components will then appear at ν0 + f and ν0 − f , which amplitudes are proportional to J1 (α π2 ). However, higher order odd components will appear if the modulation amplitude is sufficiently high. In our case, the modulation amplitude is chosen so that the first odd components achieve maximum amplitude for which the amplitude of the second odd components remain negligible. From a transmission perspective this carrier suppressed modulation is very suit­ able, since the phase shift induced by chromatic dispersion between the main com­ ponents will not give power fading of the detected signal, but only a constant phase shift of the detected microwave signal (see Appendix A). From the transmission argument as well as frequency doubling, this MZM carrier suppression technique was chosen and successfully used in Papers C, E, F and G.

33

1

Optical field

Optical intensity

Optical Intensuty, Optical field

0.8

Output intensity

0.6 0.4 0.2 0 −3π/2 −π

−π/2

0

π/2

π

3π/2

2π

−0.2 −0.4 −0.6

Bias point Output field

−0.8

Modulation voltage

−1

(a)

Output intensity 1

Optical intensity

Optical field

Optical Intensuty, Optical field

0.8 0.6 0.4 0.2 0 −3π/2 −π

−π/2

0

π/2

π

3π/2

2π

−0.2 −0.4

Bias point

−0.6 −0.8

Modulation voltage

−1 Output field

(b)

Figure 3.3: Illustration of harmonic generation using the transfer characteristic of the MZM. (a) Second harmonic generation using carrier suppressed biasing. (b) Fourth harmonic generation.

In practice it is difficult to achieve perfect carrier suppression with MZM, be­ cause it is cumbersome to perfectly adjust the polarization of light in to the modu­ lator. If the MZM contains a polarizer, this might be slightly misaligned which also results in unmodulated light passing though the MZM. To improve the carrier sup­ pression two different techniques have been used in this thesis. In Paper E a FBG was used as a band-stop filter to improve the carrier suppression after the MZM.

34

RBW: 60 pm

Optical Power (dBm)

1544

(a)

RBW: 10 pm

40 GHz

(c)

0

1544.5 Wavelength (nm)

0 RBW: 60 pm -10 -20 -30 -40 -50 -60 1544 1544.5 Wavelength (nm)

1545

(b)

Optical power (dBm)

Optical Power (dBm)

10 0 -10 -20 -30 -40 -50 -60

−10 >40 dB −20

−30

−40

1545

−50 1550.2

1550.4

1550.6

1550.8

1551.0

1551.2

Wavelength (nm)

Figure 3.4: Spectra of improved carrier suppression using fiber Bragg grating and external polar­ izer. (a) Spectrum before and (b) spectrum after fiber Bragg grating. (c) Spectrum after MZM followed by an external polarizer.

The spectra without and with the FBG is shown in Fig. 3.4.(a) and (b). Using the FBG also enhanced the stability since the setup became less sensitive to bias drift of the MZM. Another technique was used in Paper G where a polarizer was put subsequent to the MZM. The polarizer removed all light which was misaligned in the MZM, leaving only modulated light to pass through. The result of this is shown in Fig. 3.4.(c), where a carrier suppression of over 40 dB was achieved. Furthermore, quadruple multiplication of the fundamental frequency using the MZM is also a possibility. If the MZM is driven with frequency f /4 and biased for maximum transmission, odd frequency harmonics will be suppressed, and with a carefully adjusted amplitude, two strong frequency components separated by f will be created [102], schematically as shown in Fig. 3.3(b).

3.3.2 Harmonic Frequency Generation with a Phase Modu­ lator [Papers A & B] Harmonic frequency generation can also be done for systems having transfer func­ tions similar to the MZM. In paper A three techniques for harmonic generation were presented, utilizing the polarization dependent properties of an optical phase mod­ ulator (see section 2.2.1). The presented techniques were used for generation of the fourth harmonic in three setups with a polarizer, a fiber loop-mirror configuration, and a Faraday-mirror and a polarization beam splitter, respectively, as polarization to amplitude converters. The best and most stable results were achieved with the setup including a Faraday-mirror and a polarization beam splitter and the setup

35

frf

(a) PC PBS

PC

td

Faraday mirror

0

(c) -10

Laser Pin

Phase modulator

Power (dBm)

Pout

(b) Amplitude (A.U.)

Analyzers

-20 -30 -40 -50 -60 -70

Time (10 ps/div)

0

10

20 30 Frequency (GHz)

40

50

Figure 3.5: (a) Setup of quadruple frequency generation using a optical phase modulator, a Faraday-mirror and a polarization beam splitter. (b) Oscilloscope trace and (c) electrical spectrum of the generated 40 GHz mm-wave.

is shown in Fig. 3.5.(a). By analyzing the setup the following expression for the optical power output could be derived

Pin Pout (t) = 2

�� � � π(A cos(2πfRF t) − ηA cos(2πfrf (t − td ))) 1 + cos , Vπ

(3.1)

where A and frf is the amplitude and frequency of the modulation, respectively, td is the delay between modulator and Faraday mirror, η is the modulation strength in the backward propagation of the modulator. However, the time delay between faraday mirror and modulator must be optimized to the frequency, for in-phase modulation, such as fRF td = n, where n in an integer. The harmonic generation output is shown in Fig. 3.5.(a) and (b), as an oscilloscope trace and frequency spectrum. The created 40 GHz carrier was mainly determined by the original 10-GHz signal generator, and the suppression of undesired lower order harmonic frequencies exceeds 30 dB. This technique has been adopted by others for the same purpose of fourthharmonic signal generation [103]. Furthermore, this technique was use as base for an optoelectronic oscillator in Paper B. An optical phase modulator could also be used in cascade with an MZM, in order to enhance the frequency multiplication. Running the MZM in carrier suppressed mode, and then adding the modulation of the phase modulator with the proper phase and amplitude, a sixth-harmonic is generated [104]. However, a drawback with this technique is that the amplitude of the sixth harmonic is in the same order of the other harmonics, and no single frequency is generated.

36

3.3.3 Harmonic Generation using Nonlinear Phenomena in Optical Fibers [Paper C]

LASER

Wavelength

Wavelength

Four Wave Mixing

Photo Detector

Mach-Zehnder Modulator

Electrical Input

-20

High-Power EDFA

Highly NonLinear Fiber

Fiber Bragg Grating Filtering

Electrical output

Electrical Power (dBm)

Optical Power

Wavelength Carrier Suppressed

Optical Power

Optical Power

The nonlinear fiber phenomena FWM is possible to use for harmonic generation, since the FWM process creates new frequency components. In the literature FWM is proposed for frequency comb generation where the desired frequency components are filtered out and heterodyne mixed on a photodiode. Signals with high spec­ tral purity, two and three times the original frequencies, are created by FWM in dispersion shifted fiber (DSF) [105]. In Paper C photonic frequency multiplying with FWM was refined and ex­ tended, and used for sixfold multiplication of the fundamental frequency. A MZM biased for carrier suppression created the two pump wavelengths, which gives a factor to in frequency multiplication as discussed in previous sections. The pumps were amplified with high power EDFA and the FWM was achieved in a subsequent highly nonlinear fiber (HNLF), generating new frequency components separated by six times the fundamental frequency. A schematic illustration is shown in Fig. 3.6 and the optical spectra after MZM and the HNLF fiber. Furthermore, it is seen that residual power in the undesired frequency components after the HNLF is re­ moved by two FBGs, which transmission characteristics are shown in Fig. 2.12. After the filtering only two strong frequency components are left, a seen the right spectrum. Having only two strong optical components makes this perfect for dis­ tribution in optical fiber since it is not affected by chromatic dispersion. In this paper it was shown for 40 GHz generation, mainly limited by the available compo­ nents, could easily be scaled to higher frequencies. By changing the fundamental frequency the created frequency can be much higher, e.g. starting with 40 GHz will give 240 GHz output. This technique is mainly limited by the bandwidth of the photodetector and the MZM, but UTC-PD has been presented having bandwidth exceeding 300 GHz [34]. The idea of using FWM to create a harmonic frequency RBW: 1 MHz

40 GHz

-30 -40 -50 -60 -70 0

10

20 30 40 Frequency (GHz)

50

fOUT = 6.fIN = 40 GHz

fIN = 6.67 GHz

Figure 3.6: Schematic illustration of optical six time multiplication utilizing FWM.

output has recently been adopted by other research groups. They have instead of

37

HNLF used FWM in semiconductor optical amplifier (SOA), based on two optical phase locked lasers [106] or two cascaded MZM [107].

3.3.4 Signal Generation with Chirped Pulse Photomixing [Paper D] Widely tunable signal generation can be accomplished by photo mixing of chirped optical pulse. The principle of operation of chirped pulse photomixing is that a chirped pulse is mixed with its time-delayed replica to create a widely tunable pulsed signal at frequencies in the GHz to THz range. The generated frequency arises from the difference frequency between a chirped pulse and its Δτ -time-shifted replica, as shown in Fig. 3.7.(a). The concept of mixing of chirped pulses from (a) 1

∆t

SHG intensity [au]

ν

(c)

T

ν

∆ν

0

0.8 0.6 0.4

f

2

4 6 Time [ps]

80

PC

PBS Photo Detector

fs-laser or FBG

Oscilloscope

Variable

(b) Autocorrelator

Voltage [mV]

DGD PC

10

(d)

t t =0 Dispersion

8

60 40 20 0 −1

−0.5

0 Time [ns]

0.5

1

Figure 3.7: (a) Time-frequency diagram illustrating the principle of chirped pulse photomixing. (b) Schematic overview of experimental setup. (c) Autocorrelator trace of generated 725 GHz signal. (d) Oscilloscope trace showing a generated 50 GHz burst.

Ti:sapphire lasers has been used for generation of THz radiation [108]. The chirp was then obtained by using a free-space grating pair, and the time-shift was in­ troduced by a Michelson interferometer. In Paper D a fundamentally mode-locked fiber-laser with a center wavelength of 1.55 µm, and instead of free space diffraction gratings, a chirped fiber Bragg grating (FBG) or chromatic dispersion in multiple kilometers of single-mode fiber was used. An overview of the experimental setup is illustrated in Fig. 3.7.(b). The pulse train from a passively mode-locked femtosec­ ond fiber laser was dispersed by either a spool of single-mode fiber or a chirped FBG. The time-shift was then created by launching equal amounts of light into the two axes of an adjustable differential group delay (DGD) element followed by a polarization controller and a polarizing beam splitter (PBS) to polarize the light at 45◦ with respect to the optical axes of the DGD element. To the outputs of the

38

PBS, an optical spectrum analyzer, a sampling oscilloscope, and/or an autocorre­ lator were connected. The result of the experiment showed excellent modulation depth and good phase stability, and that the frequency of the signal can easily be tuned by changing the time-delay between the chirped pulse and its replica or by changing the amount of dispersion. Generation of 725 GHz and 50 GHz is shown Fig. 3.7.(c) and (d). However, higher order dispersion results in that the signal acquires a frequency chirp. Analytical expressions linking the dispersion and the dispersion slope to the central difference frequency and the chirp in the difference frequency as presented in Paper D.

3.3.5

Mode-Locked Lasers and Frequency Comb Generation

Microwave and mm-waves can also be generated by mode-locked lasers or frequency comb generators. Mode locking at the desired frequency, will give a pulse train with repetition rate of the mode locking frequency, which at photo detection will give a strong signal at the mm-wave frequency. With a comb generator a broad spectrum of discrete frequency components are generated. Two of these components are filtered out and with photo mixing a mm-wave will be created equal to the frequency separation of the comb components. Pulse generation using actively mode-locked laser diodes have been proposed for usage in creating mm-waves. The mode-locking can be done in several ways. Short pulse operation can be made by changing the refractive index of the material within the laser cavity. With this method mode-locked laser has been presented based on an optically pumped Neodymium-doped LiNbO3 cavity, where locking up to 20 GHz with 96% modulation depth and high output power was achieved [109]. An alternative method of mode-locking of an optically pumped diode laser is by modulating the pump laser with a subharmonic frequency. Such a device has been demonstrated for 60 GHz mm-wave generation [110]. By monolithically integrating an electroabsorption modulator within the laser cavity mode selection can be done, and a pulsed output be achieved at 50 GHz with 25 GHz modulation [111]. As mentioned, from a frequency comb, discrete frequencies can be filtered out and used for mm-wave generation by photo mixing in the photo detector. A fre­ quency comb with 25-GHz spacing between the components has been proposed as light source in a multi-wavelength RoF application [112]. Here, a mode-locked laser diode was used as frequency comb generator stabilized with a 12.5 GHz RF oscil­ lator. For very high frequency generation, over 100 GHz, solutions which involve optical comb generators have been suggested. The optical comb can be used e.g. for injection locking of two lasers for frequency stabilization [113]. Here frequencies up to 110 GHz are reported, mainly limited by the bandwidth of the photodetector. Furthermore, an optical comb generator has also been proposed in a system with optical phase-locked loop [114], where a tunable laser was tracking a master laser with the feedback system and mm-wave generation over 150 GHz was achieved. However, a drawback with these two methods is the temperature sensitivity, which limits the locking range.

39

3.4

Heterodyne Generation

The heterodyne generation techniques utilize the fact that photocurrent is propor­ tional to the optical power and thereby proportional to the square of the optical field to create micro- and mm-waves (see section. 2.4). If two incident light waves on a photo detector with common polarization, separated by the desired mm-wave frequency, the resulting photo current will then have a modulation with a frequency identical to separation of the optical waves. This technique has advantage com­ pared to generation with modulators, since it inherently has full modulation depth, if the light sources have identical amplitude (see Appendix C), creates no harmon­ ics and is dispersion tolerant. The disadvantage with this technique is the relative stability between the lasers used to generate frequency. Using two free-running lasers for heterodyne signal generation will result in a phase unstable mm-wave signal. However, to overcome this problem different solutions have been suggested including dual frequency lasers and frequency locking of lasers.

3.4.1

Dual Lasers Techniques

Using two lasers for heterodyning is a straightforward technique. These lasers should have a wavelength difference equal to the desired mm-wave, since the photo mixing in photo diode will create an mm-wave. However, in order to achieve a stable mm-wave output, the lasers have to be stabilized relative each other. Free-running lasers have a linewidth broadening and wavelength drift due to temperature or environmental perturbations. This result in substantial phase noise of the generated signal and also frequency drift which results in a wide linewidth and also a frequency instability of the output mm-wave. Consequently, the wavelength difference must be kept constant and therefore some kind of feedback system must be used to lock or couple the lasers to each other. To accomplish this, the light emission of a laser can be coupled and locked to a master laser, by modulating this at a frequency f . The output is injected into the slave laser to seed it to lase at one of the harmonics of f from the master laser. The slave laser is then locked to the master laser and a good heterodyne signal has been achieved at 60 GHz [115]. The injection locking technique could also be used to lock more than one laser in order to have more channels, which has been demonstrated for mm-wave signals in the 60 GHz band [116]. Two examples of systems that use negative electrical feedback to control the output mm-wave quality are the optical frequency locked loop (OFLL) [117] and the optical phase locked loop (OPLL) [117, 118]. To generate mm-waves with high spectral purity, very narrow linewidth lasers are required in the OFLL compared to the OPLL, which requires lasers with high frequency modulation index and a fast feedback loop. The optical injection phase-locked loop (OIPLL) is a refinement of the previous described techniques. This method uses standard linewidth lasers with the long term stability offered by the OPLL, without the requirement for very short loop delay time [119]. In addition to a phase-lock loop configuration, an injection locking

40

path is added to lock the slave laser to a master laser modulation sideband. The benefit of this technique is that the bandwidth of the feedback loop is not a limiting factor and tunability from 26-40 GHz has been demonstrated. However a drawback is the complexity of the feedback system that must be used.

3.4.2

Dual Wavelength Lasers

A dual wavelength source for mm-wave generation with photo mixing is possible with a single device, e.g. a dual wavelength laser. Such laser could for example be distrusted feedback (DFB) laser diodes, with specially designed cavities. Presented is a dual-mode DFB laser which emit two optical wavelengths with a separation of 60 GHz [120]. Furthermore, with a two section gain coupled DFB-laser, a tunable dual wavelength source is achieved [121], ranging from 20 to 64 GHz. Two mono­ lithic integrated DFB lasers can generate mm-wave frequencies above 45 GHz with heterodyne detection [122]. Here optical injection locking was used for frequency stabilization. Moreover, dual wavelength sources from fiber based devices are also reported. The fiber based DFB laser was running in dual mode and the DFB section was made of fiber Bragg gratings printed in Erbium-Ytterbium co-doped fiber [123]. Linewidths less than 1 kHz are demonstrated for microwave frequencies between 1 and 3 GHz, where the tunability was done with temperature. Furthermore, a singlelongitudinal mode fiber laser using fiber Bragg gratings has also been presented [124]. A tunable fiber ring laser running at two frequencies with dual polarization for microwave generation has been demonstrated [125], with tunability from below 100 kHz to 14 GHz. Recently, a dual wavelength Brillouin fiber ring lasers, which utilizes the narrow bandwidth of Brillouin amplification to choose lasing mode and to achieve narrow mm-wave linewidth which is in the order of 10 Hz [126].

3.5

Opto-electronic Oscillator

All previously discussed signal generation techniques are based on an electrically synthesized signal. However, using both optical and electrical technology, a self oscillating cavity can be created, which have both electrical and optical output. These kinds of oscillators are referred to as opto-electronic oscillators (OEO) and belong to a new class of oscillators, which uses both electrical and optical technology [27]. An OEO utilizes the transmission characteristics of an optical modulator and a fiber optic delay to convert light energy into stable and spectrally pure microwave reference signals. The basic principle of an OEO is shown in Fig. 3.8. In this case, light from a pump laser is introduced into a MZM and then transmitted through a long fiber and detected with a photodetector. The output of the photodetector is amplified by an electrical amplifier and the fundamental frequency filtered out before it is connected to the electrical port of the modulator. This configuration will start to oscillate from noise if the gain of the loop is higher than the losses. The oscillating frequency is determined by the fiber delay length, the bias of the

41

modulator and the characteristics of the bandpass filter. The round trip time of the loop determines a set of possible oscillating modes, which is similar to a Fabry-Perot resonator. The mode frequencies can be expressed as 2π fk τ = 2kπ

k = 0, 1, 2 . . . ,

(3.2)

where k is the mode number, τ is the round trip time and fk is the corresponding frequency of mode k. An oscillating mode is selected by the bandpass filter, which has to be very narrow, especially if the delay line is long. Since an OEO is sensitive to temperature changes the fiber length will change slightly with temperature and thereby change τ . The round trip time determines the oscillating frequency which then will be changed. This can be compensated for by controlling the environment and/or with feedback systems with fiber stretchers to control the fiber length. Another configuration to select the oscillating mode is to use several loops [127]. With this configuration, each loop will have its own set of modes and there will be oscillation only at the frequencies where these overlap. The OEO can also be coupled or integrated with a mode-locked laser, in order to simultaneously create a low phase noise microwave signal and low-jitter optical pulses [128]. Recent results show that microwave generation with phase noise less than -150 dBc/Hz in the 10 to 100 kHz range and simultaneous generation of optical pulses with 2 fs jitter, is possible [129]. The main benefit of an OEO is that it can have extremely low phase noise, thanks to the long optical delay line [27]. In some applications where the phase noise is very important, the use of such device can be very beneficial. However, most proposed OEO design include a long fiber based delay lane. This is temperature sensitive which can cause frequency drift of the OEO. Therefore, it is needed to keep

Opto-electronic oscillator MZM

Coupler

Laser Bias Electical Output

Coupler

Optical Output

Optical Fiber

Filter Electical Optical

PD

Figure 3.8: Schematic illustration of an opto-electronic oscillator based on a MZM.

42

the fiber in a temperature controlled environment or use fiber-stretchers combined with a feedback system to compensate for loop lengths changes. Alternatively could more temperature change insensitive fibers be used, such as photonic crystal fibers (PCF), but a drawback these fiber are that they have large insertion loss due to mode mismatch with SMF [130]. Presently products are commercially available based on this technology [131] with phase noise below -142 dBc/Hz at 10 kHz offset from carrier.

3.5.1 Opto-Electronic Oscillator with Harmonic Output [Paper B] In Paper B was an OEO presented based on the concept of a phase modulator, a Faraday mirror and a polarization beam splitter in combination for signal gener­ ation, which was introduced in Paper A. The OEO was of cavity configuration, shown in Fig. 3.9, with cavity 1 and 2 in combination which determinate the oscil­ lation frequencies. The OEO is oscillating with 10 GHz as fundamental frequency selected with a narrowband electrical filter. An oscilloscope trace and a phase noise measurement of the fundamental frequency is presented in Fig. 3.9.(a) and (b). As can be seen in the phase noise measurement, the phase noise close to the carrier is quite high, which is due to no special care was taken to ensure long term stability, since the main objective of this OEO was to produce a comb of harmonics. Harmonics of the fundamental oscillating frequency were produced at the opti­ cal output, created by the high electrical amplification and the nonlinear transfer function of the setup. The desired harmonics were filtered out by amplifying them with optical narrow band Brillouin amplification, discussed in 2.6.2. The setup used and the results of the Brillouin amplification is seen in Fig. 2.13.(e)-(g). Frequencies up to 60 GHz where measured with a fundamental OEO frequency of 10 GHz. An oscilloscope trace of a generated 40 GHz signal shown in Fig. 3.9.(c). Furthermore, this technique could be used for generation of higher frequencies beyond 60 GHz.

43

PC PBS

PC

Phase modulator

Faraday mirror

(a)

Laser Cavity 1 Delay Cavity 2

OBPF 3 dB optical output

20 dB

Analyzers

PD

Phase Noise (dBc/Hz)

20ps/div

0

(b)

-40 -80 -120 -160

102

103

104

105

106

107

Offset Frequency (Hz)

Harmonics

(c)

.

.

Brillouin amplification.

PD Optical Output

20ps/div

Figure 3.9: Illustration of the Opto-Electronic oscillator with Brillouin amplified harmonic output. (a) Oscilloscope trace of the 10 GHz fundamental frequency. (b) Phase noise measurement of the 10 GHz signal. (c) Oscilloscope trace of a 40 GHz generated with selective Brillouin amplification.

44

Chapter 4

Microwave and millimeter-wave communication systems

C

ommunication is an important application of microwave photonics. In this chapter different aspects of communication and different methods for generat­ ing, modulating and propagating micro- and mm-wave signals are discussed.

4.1

Modulation Formats

To use the available bandwidth as much as possible, multilevel modulation formats can be utilized to increase the spectral efficiency [132]. In the binary case each bit is sent separately, i.e. the number of bits per symbol, b = 1, and the bandwidth Δf used is approximately equal to the data rate, R (bits/s). In multilevel modulation, two or more bits (b) are forming a symbol and the number of possible symbols is given by M = 2b , which corresponds to 2b possible b-bit sequences. In practice every symbol is represented by e.g. a specific amplitude and/or a phase-state, which is unique for every symbol. Thus, for M symbols an equal amount of different states are needed. This is called M-ary modulation. The symbol rate K is equal to B/b via Δf ∝ K =

B B = b log2 (M )

(4.1)

Consequently, if b is increased the used bandwidth is decreased and with the bit rate kept constant, the spectral efficiency is increased. However, noise will always put a limit on the maximum possible error-free transmission in a communication system. The maximum error-free transmission rate C (bits/s), for an ideal system with additative white Gaussian noise, was formulated by Shannon in 1948 and can be written as [133] � � S C = Δf log2 1 + N 45

(4.2)

incoherent OOK 01 BPSK/2-PAM 0

0

1

QPSK 00

11

1 M=2

M=2

10

M=4

16-QAM 16-PSK

8-PSK 011

001

010

110

000 100

111 101

M=16

M=8

M=16

Figure 4.1: Illustration of different multilevel modulation formats.

where S/N is the signal to noise ratio within the system bandwidth Δf . This equation sets the absolute limit for the condition for error free transmission. Thus, the information rate R must be less than C for theoretically having an error-free transmission. If R is larger than C, reliable transmission is not possible. The conclusion that can be drawn is that increased spectral efficiency requires higher signal-to-noise ratio. The simplest digital modulation format is binary intensity modulation, where only the power is modulated in on-off mode, so called on-off keying (OOK). This has been widely used in optical communication systems since it is straight forward to implement and no concern have to be taken about the phase of the carrier, which in this case is optical, since it is lost in the square-law detection. The detection is hence incoherent and is represented in Fig. 4.1 in the upper left corner. In electrical communication the signals are regarded as baseband or bandpass transmission, which in the latter case is encoded on to a carrier. In baseband transmission the modulation is encoded in the amplitude, in two or many levels. On the other hand, in bandpass transmission the information is encoded in the carrier’s amplitude, phase or frequency, or combinations of them. This could also be applied in optical communication since the information is en­ coded on the optical carrier. However, in order to retrieve the phase information, coherent or interferometric detection is required. The demand for higher bit-rates, together with the limited available frequency bandwidth for wireless transmission, have increased the interest of implementing multilevel modulation schemes to enhance the spectral efficiency. Widely used mul­

46

tilevel modulation formats are phase shift keying (PSK) and quadrature amplitude modulation (QAM), where the information is carried by both amplitude and phase in combination. Amplitude modulation such as pulse amplitude modulation (PAM) is also used but requires higher SNR for the same number of amplitude levels. The mentioned modulation formats are illustrated in Fig. 4.1 in the complex signal space, where the x-axis and y-axis corresponds to real (in-phase, I), and imagi­ nary (quadrature-phase, Q) axis, respectively. This representation of modulation formats in complex space is often recognized as constellation diagrams. Every dot represents a symbol, which corresponds to a bit combination. In order to minimize the number of bit errors in case a symbol is misinterpreted upon reception, Graycode are used to map bits to symbols in such way, that only one bit is changed for every adjacent symbol. This leads to, that for every symbol error there will only be one bit error. An example of this mapping is shown in Fig. 4.1 for 8-PSK. 1 Gb/s

231-1 10-5

44 km

10-6

2.5 Gb/s

BER

10-7 10-8 10-9

44 km

10-10

1.0 Gbps, 0 km 1.0 Gbps, 22 km 1.0 Gbps, 44 km 2.5 Gbps, 0 km 2.5 Gbps, 22 km 2.5 Gbps, 44 km

10-11 10-12 10-13 -14

-13

-12

-11 -10 -9 -8 Received optical power (dBm)

-7

-6

Figure 4.2: Example of BER measurement of signal sent with a 40 GHz carrier over optical fiber and wireless transmission at 1 and 2.5 Gb/s which was presented Paper E.

4.1.1

Performance Measures

The ultimate performance measure of any digital transmission system is the bit­ error-rate (BER). This can be measured on a real system by sending a known bit-sequence through the system and receiving the sequence on the output. The received bit-sequence is compared with the sent one and errors are counted. Systems which perform such measurements are generally referred to as a bit-error test sets (BERT). A BER-curve is commonly measured by changing e.g. the input power to the receiver, in order to demonstrate the sensitivity of the system. The commonly used sensitivity measure for optical communication is the optical power received in

47

order to have a BER of 10−9 , in other words one error in one billion bits sent [48]. An example of a BER measurement from Paper E is shown in Fig. 4.2. The conclusion from Eq. 4.2 is that the SNR and error rate are related. By assuming that the noise has a certain distribution, usually Gaussian, theoretical formulas can be derived which describes the relation between SNR and the proba­ bility of a symbol error to occur [132]. The formulas for calculating the probability of a symbol error for M-PSK, M-PAM, M-QAM, and, coherent and incoherent OOK, are listed in Appendix D. The probability of a bit error for some modu­ lation formats is presented in Fig. 4.3(a) as a function of the SNR per bit. To compare the different modulation formats, the SNR per bit which is needed to achieve a bit error rate of 10−6 is plotted in Fig. 4.3(b) as a function of the number of bits per symbol or relative bit-rate bandwidth. For M = 4, the performance of PSK and QAM will be similar, but for M > 4 higher SNR is needed for M-PSK than M-QAM in order to have the same performance. This is due to the fact, that the symbols are more closely spaced for M-PSK in the complex signal space and thereby the distance between the symbols is shorter, as seen in Fig. 4.1. However, in some cases it is an advantage to use M-PSK modulation since the signal has constant amplitude, for instance when the link has high nonlinearities. 1 Incoherent OOK

10

16-PAM

-4

10

-6

10

-8

24

64-QAM/ 16-PAM

BPSK/2-PAM/ QPSK/4-QAM 16-QAM/ 4-PAM

16-PSK 8-PSK Coherent OOK

10 -10 0

SNR per bit@(BER=10-6) (dB)

Probability of a bit error

10

-2

Relative bit-rate bandwidth 1/2 1/3 1/4 1/5

1/6

16-PAM

22 20 8-PAM

18 16 14

Incoherent OOK Coherent OOK 4-PAM 8-PSK

16-PSK

64-QAM

16-QAM

12 10

QPSK/4-QAM BPSK/2-PAM

8 5

10 15 20 SNR per bit (dB)

25

30

1

2

3 4 bits per symbol

5

6

Figure 4.3: (a) The probability of a symbol error as a function of the SNR per bit for some multilevel modulation formats. (b) A comparison of the SNR per bit needed to achieve a bit error rate of 10−6 for different modulation formats as a function of bits per symbol or relative bit-rate bandwidth.

4.2

Subcarrier Transmission

It is often convenient to consider the optical link as a ”black box” with an elec­ trical input and output, and use microwave subcarriers to carry the information through the link, as seen in Fig. 4.4. However, this requires that the optical link

48

is linear, so that the signals, which are sent through the system, are undistorted. Multiple data channels can be sent in parallel by using different subcarrier fre­ quencies. Subcarrier multiplexed (SCM) systems are for instance used in CATV distribution, as mentioned in chapter 1. Optical links could be implemented in such systems to distribute these subcarriers since the low loss in the optical fiber extend the transmission distance substantially compared with electrical cable dis­ tribution [13, 134]. The CATV SCM signal can consist of hundreds of low-bit rate channels. As mentioned, this places high requirement of the linearity of the link since any nonlinearity will cause interaction between the subcarriers, which will cause crosstalk and degrade the performance. SCM is also proposed for use in high bit rate optical communication [134] as an alternative to time division multiplexing, in order to increase data throughput [135]. In optical communication, multilevel modulation has obtained considerable atten­ tion in order to further increase the data rate and improve spectral efficiency [136]. However, with optical multilevel modulation of phase and amplitude, the complex­ ity of the transmitters and the receivers is increased. The increased complexity is due to the fact that the optical phase is lost upon photo detection and only the intensity envelope of the signal is obtained. Therefore, to retrieve the phase infor­ mation, coherent or interferometric detection is required. By using SCM, multilevel modulation is modulated on an electrical carrier which then is modulated on an optical carrier. After the optical link, the information is still carried on the subcar­ rier, and therefore the optical link could be simple and transparent. Nevertheless, chromatic dispersion will still limit the transmission distance of an SCM system if dual sideband modulation is used. It has been shown in a transmission link with no compensation for dispersion, that this can be overcome if each subcarrier in one sideband is filtered out and separately directly detected [137]. By using disper­ sion compensation, up to 20 Gb/s data throughput has been demonstrated using SCM [138]. Apart from the poor dispersion tolerance, dual sideband modulation is also optically spectral inefficient. To improve the spectral efficiency and reduce the effects of dispersion, SSB modulation in SCM system has been considered [139,140]. f1

f1 “Black box”

Data 1

Data 1

f2

f2

Data 2

Data 2

O/E

E/O

fn

Electrical input

Electrical output

fn Data n

Data n

Figure 4.4: Schematic overview of subcarrier multiplexing.

49

However, it has been shown that even though the dispersion effect is reduced, it will still cause a transmission distance limitation. Furthermore, SCM channels could be used together with wavelength division multiplexing to further increase data throughput. Nonlinearities in the optical fiber, such as XPM, FWM and SBS, can deteriorate the SCM signals, since much power of the optical signal is concentrated in the optical carrier. This sets a constraint on how high total power that could be used in the transmission in the fiber [139, 140]. Subcarrier modulation could also improve capacity on optical links which has a band-pass transmission characteristics such as the standard multimode fiber (MMF) link [141]. Using this technique, 2.5 Gbit/s transmission has been demon­ strated by using binary modulation and a 5 GHz subcarrier. The capacity is in­ creased if QPSK modulation is used and 5 Gbit/s throughput has been achieved [142]. The data throughput in a fiber could be further improved if dense WDM (DWDM) is used, with optical carriers as close as 20 GHz [142].

4.2.1 Near-Baseband Synchronous Subcarrier Modulation [Paper H] In all above-mentioned subcarrier systems, the subcarrier frequency is at least two but usually many times larger than the symbol rate. This will give poor utiliza­ tion of the spectrum close to DC, and a lower overall spectral efficiency in the optical domain, if the full bandwidth from DC to the highest carrier frequency is considered. In Paper H, near-baseband synchronous subcarrier modulation was introduced, where the subcarrier frequency is equal to the symbol rate. The band­ width utilization is half of the optical spectral efficiency, compared to baseband data modulation, if dual sideband modulation is used and comparable if SSB modulation is used. As any subcarrier system, the requirements on electrical components are relaxed with respect to the lower cut-off frequency, compared to electronics for baseband communication, since no frequency components are present near DC. Furthermore, binary PSK modulated near-baseband subcarrier will have similar appearance as the carrier-less baseband modulated Manchester code [133], but this similarity is lost when more than two states are used. In Fig. 4.5 an overview of setup for a near-baseband system is shown with schematic optical spectra of the signal is shown in the upper left corner of the figure. In the same figure, a 2.5 Gsymbols/s QPSK modulated signal is shown with the corresponding oscilloscope trace, electrical spectrum and as demodulated in a constellation diagram, as an example. The signal here has been sampled with a realtime oscilloscope and then processed with an in-house Matlab program in order to recover the carrier and demodulate the signal. In Paper H 16-PSK and 16-QAM were demonstrated with 2.5 Gsymbols/s giving a data rate of 10 Gb/s. In subsequent, yet unpublished measurements the symbol rate are increased to 5 and 10 Gsymbols/s, resulting in 20 and 40 Gb/s, respectively. It has been shown by [143] that high-bandwidth (> 10 GHz) links are possible with low-cost VCSELs and high bandwidth MMF. Using this result together with

50

near-baseband synchronous subcarrier modulation, the data throughput could be improved substantially. However, this still requires the link to behave linearly and not distorting the signal, and that the SNR is sufficient. Experiments and simulations indicate that the required signal bandwidth is about 1.5 times the symbol rate/subcarrier frequency. For example, if a 16-QAM signal was sent with this method using a symbol rate of 10 GSymbols/s, the data throughput would be 40 Gbit/s using 15 GHz bandwidth. Furthermore, if a broadband dual arm MZM is used, even higher symbol rates could then be reached by using SSB for spectral efficiency. To reach 107 Gb/s a symbol rate of 26.75 Gsymbol/s would be needed if 16-QAM where used and the required signal bandwidth would be 40 GHz. However, it still remains to be determined how this behaves over long transmission distances. Moreover, working with electrical signals pre- or post distortion could be used to improve the signal and compensate for link-induced impairments.

Electrical Power (dB)

Oscilloscope trace

Optical spectrum

10 GHz

ν

ν

0

Electrical spectrum

0 −10 −20 −30

f =2.5 GHz c

−40 −50

DATA

0

2

400 ps

Bias

Carrier recovery Demodulation

0

LD

MZM Transmitter

10

Demodulated QPSK

Near-baseband synchronous subcarrier generator

ν

4 6 8 Frequency (GHz)

DATA

PD Optical link

Receiver

Figure 4.5: Overview of the near-baseband synchronous subcarrier modulation link with the signal shown in the optical and electrical domain.

4.2.2 Binary Digital Electronic Near-Baseband Synchronous Subcarrier Signal Generation [Paper H] Digital modulation of an RF-carrier or subcarrier is usually done with an IQmodulator, with which an arbitrary signal could be generated, e.g. PSK or QAM by using digital-to-analog converters (DACs) with sufficient high sample rate and bandwidth. However, using very fast DACs, the modulated RF-subcarrier could be created directly and the need for an IQ-modulator is eliminated. To reach symbol rates beyond the bandwidths of DACs available today, one solution is to use binary digital electronics for the signal generation. This electronics have very

51

a) BPSK

Symbol

Symbol 1

0

Td

Td Ts

Ts

b) QPSK Td

Td

01 11

00

Ts

Ts

10

Td

Td

c) 8-PSK 010

011 001

110 111

Ts Td

000 100 101

Ts

d) 16-PSK 1000

1001

0000

1011

0001

1010

0011

1110

0010

1111

0110

1101

0111

1100

0101

0100

Td

Ts

Ts

Figure 4.6: Schematic of PSK symbol generation with binary digital circuits.

high bandwidth, and data patterns have been generated for bit-rates up to 165 Gb/s [144]. In Paper H we demonstrated that by programming a pulse pattern generator, M-PSK near-baseband subcarrier modulation signals could be created, which is shown in Fig. 4.6. The relationship between bandwidth of the digital electronics Bd , symbol rate Bs and number of phase statesM , is Ts = M · T d

or

Bd = M · B s

(4.3)

if the symbol rate and the carrier frequency are equal, where Td = Bd−1 is the digital bit slot and Bs = TS−1 is the symbol rate. The different symbols for M = 2, 4, 8, 16 52

(a)

2

2

(b)

2

Figure 4.7: Schematic of two possible implementations of 16-QAM near-baseband synchronous subcarrier generation based on binary digital electronics. (a) Four BPSK signals are combined as described above tow generate the 16-QAM. (b) Two in-phase QPSK signals are combined, one with double amplitude compared to the other, and the result is a 16-QAM signal. The triangles stands for amplification of the amplitude and the rings with a cross inside stand for addition of two signals.

53

are shown in Fig. 4.6, where the digital values that defines the symbol is marked with binary reflected Gray code (BRGC) [145]. It is apparent that for many phase levels the bandwidth of the binary digital electronic circuits has to be very large, but the bandwidth of the analog equipment (electrical amplifiers, modulator or directly modulated laser, photo detector) in the link only has to be the subcarrier frequency plus the symbol bandwidth. For example if 20 Gb/s data is sent with 16-PSK at 5 Gsymbols/s, the digital electronic is required to be clocked at 80 Gb/s, but only 10 GHz bandwidth of the analog electronics and the transmission link is needed. As discussed previously, 16-QAM has less requirement on SNR than 16-PSK and is therefore preferred to use. 16-QAM near-baseband synchronous subcarrier modulation could also be generated by using binary digital electronics based on either a near-baseband BPSK or QPSK signal [146]. These signals could be gen­ erated as discussed above. By adding two BPSK, signals which are in-phase with one signal amplified to the double amplitude, a four level amplitude signal is cre­ ated, which lies on the I-axis. The same principle could be used for creation of the Q-axis, but with the carrier shifted 90◦ . If these two signals then are combined, the sum of them will be a 16-QAM signal, as seen in Fig. 4.7.(a). Furthermore, the sum of two QPSK signals which are combined in-phase and with one having double amplitude compared to the other, could also be used as an alternative way to create a 16-QAM signal (Fig. 4.7.(b)).

4.3

Millimeter-Wave Photonic Links

Signal transmission from CS1 to BS1 can be performed in three ways, as RF, IF or as baseband. Since the BS should transmit a wireless mm-wave signal, IF and baseband signals must be up-converted in the BS. A large variety of different solu­ tions for transportation and up-conversion of mm-wave signals have been presented in the literature and a selection of these proposed techniques are exemplified here.

4.3.1

Remote Local Oscillator Delivery to the Base Station

The BS becomes more flexible if the LO is remotely controlled and delivered from the CS [147–149]. Furthermore, a high performance signal generator could be in the CS, which has better performance than an LO usually used in the BS. In chapter 3 several techniques for mm-wave generation were discussed which could be used for e.g. delivery of the LO signal. However, using this technique the data signal must be upconverted at the BS. Simultaneous transport of the mm-wave LO and baseband data using the same optical modulator was presented in [147] or on another wavelength [148], which is schematically shown in Fig. 4.8(a). Moreover, the remotely delivered LO could also be used for the down-conversion of the up-link signal, shown in Fig. 4.10.(a). 1 CS

and BS were defined in chapter 1, and are acronyms for central station and base station, respectively.

54

Base Station

Central Station λ-source

EOM

PD

AWG

AWG

Data λ-source

PD

EOM

(a)

Opt. LO

Base Station

Central Station λ-source

PD

EOM

(b)

LO

Data Opt.

Base Station

Central Station Dual λ source

PD

EOM

(c)

Data Opt.

Base Station

Central Station Dual λ source

PD

λ-split

(d)

EOM

Data Opt.

Base Station

Central Station Dual λ source

Single carrier modulation

PD

(e)

Data Opt.

Figure 4.8: Illustration of different methods for modulating data or IF on optical mm-wave carriers in the CS for transportation to BS.

4.3.2

Directly Modulated Millimeter-Wave Links

The full potential of a photonic mm-wave system is not utilized unless a modulated mm-wave signal is delivered to the BS. By sending a modulated mm-wave signal the BS could be simplified, since the up-conversion of data is no longer needed. However, this will require a photodetector with bandwidth in the same order as

55

the mm-wave carrier, which also is a requirement for the remote LO delivery. Wide bandwidth photo detectors are available as discussed in section 2.4.1. Photo diodes with high power capability could be directly coupled to an antenna and no electrical amplification in the BS is needed. This has been demonstrated for a UTC-PD with integrated antenna for in a 120 GHz link [150]. The data signal could be up-converted to the mm-wave frequency in the CS and modulated onto an optical carrier with an intensity modulator [151]. With this technique, the signal is propagated in the fiber with two sidebands, as shown in Fig. 4.8(b). The signal will therefore inherently be affected by dispersion in­ duced power fading, which has been discussed in previous chapters. However, the dispersion in a link could be compensated by using chirped FBGs or dispersion compensating fiber [97]. Upconversion of the data signal have also been proposed to be performed in the optical domain. The data signal can be directly modulated with an intensity modulator onto an optical mm-wave signal, which is created using e.g. carrier sup­ pression modulation or filtered components from an optical frequency comb, as seen in Fig. 4.8(c). When the combined signals are detected, a modulated mmwave signal will be created. This technique has been demonstrated using carrier suppression for a 40 GHz link with 2.5 Gb/s data and for a 120 GHz link with frequency comb generated carriers, and with a data throughput of 10 Gbit/s [152]. In this demonstration an UTC-PD was connected to a horn antenna for the wire­ less transmission. Furthermore, it has also been presented experiments where the laser source is directly modulated with the data signal and then modulated by a MZM biased for carrier suppression in order to up convert the data signal [153]. Moreover, these techniques will be less affected by dispersion than dual sideband modulation, but nevertheless the dispersion will induce penalty since both carriers are transporting the information. No power fading will occur, but since the data is on both carriers the phase of data will change with propagated distance and eventually be lost.

4.3.3 Dispersion Tolerant Millimeter-Wave Photonic Links [Papers E, F & G] The mm-wave modulation that is least affected by dispersion is SSB modulation, which has been addressed in previous chapter. SSB modulation can be achieved by using dual arm MZM or by modulating one carrier in a two carrier system. The realization of SSB modulation with a MZM was proposed by [98]. The data signal is, in conformity with double sideband modulation, up-converted in the electrical domain and modulated on the optical carrier using the dual arm MZM. This requires a broadband modulator, which bandwidth is in the order of mmwave carrier. It is known that this technique has low modulation efficiency, but can be improved by suppressing the carrier until its power is in the order of the sideband [99]. An alternative to a dual arm MZM is to use one of the techniques mentioned

56

in the previous chapter for dual wavelength generation. By only modulating one of the carriers a single sideband equivalent signal is created, as seen in Fig. 4.8(d). A carrier suppressed biased MZM modulator could for example be used as a dual wavelength source, which was utilized in Papers E, F, and G and by [154]. In Papers E and F FBGs were used to filter and split up the signals in order to modulate only one of the two carriers, and link experiments with up to a 40 GHz carrier with 2.5 Gbit/s modulation was successfully demonstrated. Another way to s3

(I)

E2 s1

E1

Amplitude modulated mm-wave output

(IIa)

s3

s2

E1 E2

E2

E1

(IIIa)

Phase modulated mm-wave output s3

(IIb)

E2 ∆τ

s1

E2 λ

E1

DGD1 s1

s2

s3

(IIIb)

E1

s2

s3

E2 E1

s1

E2 ϕ

Phase modulator

s2

(IVa)

s3

(IVb)

E2 ∆τ

s1

E1

E1

E1

s2

s3

DGD2 s1

s2

E2 λ

s1

Photo detector

E1

E1

E2

s2

E2 λ

Modulated mm-wave output

Figure 4.9: Schematic illustration of the technique presented in Paper G for modulating one sideband of a suppressed carrier modulated signal. This technique could be used in two settings to either create amplitude or phase modulation on the carrier.

modulate one carrier was presented in Paper G. Instead of splitting up the carriers in separate paths, polarization modulation together with DGD elements, and the polarization dependance of photo detection, could be used to modulate one carrier of the two carriers, as shown to the left in Fig. 4.9. The transfer characteristic of

57

this system is similar to an MZM. Alternatively, this technique could be used for phase modulation of the created mm-wave, shown to the right in Fig. 4.9.

4.3.4

Full Duplex Millimeter-Wave Systems

There are many demonstrations of full duplex implementations of mm-wave links found in the literature. In the previous sections examples of different downlink techniques were discussed. The design of the uplink is much more challenging since it is a trade-off between complexity, performance and bandwidth of components that are used. The most straightforward approach is to modulate the received RF-signal di­ rectly on an optical carrier in the BS and transport it to the CS, which is illustrated in Fig. 4.10(b). This approach seems to be simple but it needs broadband opti­ cal components, and is similar to downlinks described above [155]. Moreover, the uplink mm-wave signal could be downconverted in the optical domain by sending several wavelengths to the BS, in order to be modulated with the uplink data. The square-law detection of the photo detector in the CS will then act as a mixer and generate many mixing products, where a low frequency component with the up­ link information is one of them [156]. A nonlinear optical modulator could act as both an optical modulator and a mixer, and thereby simultaneously modulate and downconvert the uplink signal [157]. The RF-uplink signal is also suggested to be directly detected by filtering out the sidebands with bandstop-reflecting FGB [158]. This is similar to the direct sideband detection of different channels in an SCM sys­ tem [137], which was discussed earlier in this chapter. Another uplink approach is to send low frequency data signals in the uplink. This technique requires that the received RF signal is down-converted at the BS, and for that a mm-wave mixer and LO are needed. The LO could be located in the BS, illustrated in Fig. 4.10(c), similar to a downlink using upconversion of the data signals, or the LO could be distributed from the CS to the BS [147,148]. This enables many BSs to share a high performance oscillator, and the frequency can be changed remotely. The benefits of these approaches are that the bandwidth of the optical components are kept low and potentially also the cost. Furthermore, the uplink receiver in the CS becomes simple. Multifunctional components have also been proposed to be used in the BS. These components are capable of both receiving signals from CS and sending signals back to the CS. The device can be seen as a four-port with RF input and RF output, and light input and light output, which is shown in Fig. 4.10(d). These transceivers have been realized by using e.g. an EAM both as photo detector and modulator. For microwave frequencies such a device has been demonstrated without any bias, which thus makes the BS passive [159]. The electro-absorption transceiver (EAT) has also been demonstrated using different wavelengths for down- and uplink, in order to improve performance [160]. For mm-wave frequencies EAT modules operating at 60 GHz have been presented showing full duplex operation [161, 162]. Another device which has the same functions as the EAT is the asymmetric Fabry-Perot

58

Base Station

Central Station λ-source

EOM

AWG

AWG

PD

Data

PD

(a)

Opt. t.

λ-source

EOM EOM LO

Data

λ-source

PD

Opt.

Base Station

Central Station Opt.

λ-source

PD

EOM

(b)

LO

Data EOM

PD

λ-source

LO

Data Opt.

Central Station Dual λ source

Base Station Opt. PD

AWG EOM

LO

(c)

Data Data/IF

LD

PD

Opt.

Central Station

Base Station Opt.

λ-source

EOM

EAT LO

(d)

Data PD LO

Data Opt.

Figure 4.10: Illustration of different solutions of down- and uplinks for full duplex transmission.

59

modulator (AFPM). This device acts as both a modulator and a photo detector and has been shown for multi frequency use up to 6 GHz [163]. One of the advantages with the AFPM over the EAT, is that it operates by using reflection and thus only a single fiber is needed between the CS and BS.

4.3.5

Multiplexed Millimeter-Wave Signal Delivery [Paper F]

In the proposed architectures for future mm-wave system functions are centralized and concentrated to the CS which then acts as a supply node of signals for many BSs. Wavelength division multiplexing (WDM) is proposed for transport and rout­ ing of signals to specific the BS [164–169] as schematically shown in Fig. 4.11(a). Suggested topologies include star [168] or ring configurations with add/drop func­ tionalities at the BSs [164, 167]. When using WDM, each BS is addressed with a certain wavelength. Components that have been used for wavelengths demultiplex­ ing is both arrayed waveguide gratings (AWGs) [166,168,169] and FBGs [164,165]. Sending mm-waves by non-overlapping WDM is very spectrally inefficient since most of the spectrum is unused. It is therefore suggested that dense WDM (DWDM) should be used and that mm-wave frequency components should be sent interleaved with each other [166, 167, 169]. Interleaved mm-wave frequencies have been multi­ plexed/demultiplexed with DWDM AWGs with channel spacing of 25 GHz. Central station

(a) λ1

E

Base Stations

λ2 λ3

O

WDM MUX

λ4

λ2

f1, f2,f3, f4

f1

O E

λ1

WDM DEMUX UX

O E

f2

OE

f3 f4

λ3 λ4

(b) Central Station

Optical fiber

Optical Demultiplexer

MZM Laser

Data modulation of one sideband

f1/2

f2/2

f3/2

f4/2

λ

λc

f1 f2 f3 f4

λc f4

λ

λc f3

λ

λc f2

λ

λc

λ

f1

Figure 4.11: (a) Schematic illustration of WDM multiplexing and wavelength addressing of BSs. (b) Frequencies multiplexing on a single optical carrier using suppressed carrier modulation and FBGs for demultiplexing.

60

Several mm-wave channels with different frequencies could also be multiplexed on a single optical carrier. In Paper F, it was shown that several frequencies could be multiplexed on a single optical carrier using suppressed carrier modulation, see Fig. 4.11(b). The multiplexed carriers are then demultiplexed by utilizing FBGs with very good filtering and low inter-channel cross talk.

61

62

Chapter 5

Conclusion and outlook n this thesis several different techniques have been presented, where photonic

Itechnology is used in microwave and millimeter-wave applications. These tech­ niques demonstrate that microwave photonic technology is feasible to use in many

areas, especially for communication and signal generation. Future evolution of this area depends on further development of critical compo­ nents. So far, components used in telecom applications have been adopted into use in microwave photonic applications. However, the requirement on components for digital optical communication and analog RoF is somewhat different. For instance, in telecom the sensitivity at the receiver is very important, as well as good distinc­ tion between ”ones” and ”zeros” in digital intensity modulated systems. In analog systems high output power, linearity, CNR and dynamic range are of greater inter­ est. Efforts have been made to develop special components for analog applications, but nevertheless, the key to improvements are to develop the components, as well as the overall system design. For instance, further development of the UTC-PD is a key for ultra broadband, multi-band links with high linearity, dynamic range and optical power capability. To change from electrical cables to optical fiber is a paradigm shift, and it is therefore very important to consider all aspects of present systems as well as desirable enhancements in future systems. In many applications, new features can be added that reduce cost or improve the performance, which is more than just replacing the cable with fiber. As discussed in chapter 1, hybrid cables with comprising both fiber and electrical cables are available, that simplifies installation significantly and reduces the cost. This is a step towards the paradigm shift. The optical fiber is a very transparent medium in many ways. Therefore it is likely that RoF solution could be operating alongside with other services in the same optical fibers in the future. Using WDM techniques or baseband and subcarrier modulation on the same wavelengths, wired and RF signals could simultaneously be transmitted. Proof of concepts of such solutions have been presented and are of special interest in ”last mile” connections and fiber-to-the-x1 (FTTx) solutions. Microwave photonics can also find places in future niche applications, where cost 1 The

”x” can stand for curb, home, antenna etc.

63

is of less concern and ultimate performance is of higher interest. These applications can be signal generation with e.g. low phase noise OEOs, signal processing or special filters with unique features. Further investigations of broadband near-baseband synchronous subcarrier modu­ lation and its usage in datacom links are also interesting. This could improve the spectral efficiency since multilevel modulation is simple to implement with the elec­ trical input and electrical output. However, it requires high bandwidth electronics and the impact of propagation in the fiber is yet to be investigated. This can potentially be useful in short links or where the available bandwidth of the link is limited, for instance in a link with a directly modulated VCSEL and MMF. Generation and distribution of high quality LO is also a topic for future in­ vestigation. Even though numerous techniques have been presented, there is still room for improvement. In photonic TeraHertz generation the photodetector is very important. Generations of frequencies higher than 1 THz have been demonstrated with traveling-wave photo detectors [170] and UTC-PDs [171] monolithically inte­ grated with resonant antenna circuits. These experiments show that this area has a potential for further improvements. In conclusion, microwave photonics has been used in many applications and the future prospects are excellent. There is still room for improvements and more re­ search has to be invested in basic and fundamental technology, such as components, as well as overall system designs.

64

Chapter 6

Summary of papers Paper A: “Microwave Harmonic Frequency Generation Utilizing the Properties of an Optical Phase Modulator” In this paper new optical techniques for the generation of the fourth harmonic of a microwave are presented. Three different techniques are demonstrated, whose common factor is the utilization of the polarization-dependent properties of an optical phase modulator. In this way a 40-GHz mm-wave carrier is generated using only 10-GHz electronics. The signal quality of the generated frequency is mainly determined by the 10-GHz signal generator, and the suppression of undesired lower order harmonic frequencies exceeds 30 dB. The three different setups use a polarizer, a fiber loop-mirror configuration, or a Faraday-mirror together with a polarization beam splitter, respectively. Paper B: “Photonic Microwave Generator Utilizing Narrowband Bril­ louin Amplification and a Fiber-Based Oscillator” This paper is a further development of the third setup presented in Paper A, which is refined into an opto-electronic oscillator (OEO). The OEO uses a dual-loop setup with bidirectional modulation in a phase modulator and a Faraday mirror. The mirror reflects the light in the outer fiber cavity, and gives a polarization shift that enhances the stability of the oscillation. In our experiment, the oscillation is locked on 10 GHz and generates harmonics, which are extracted through narrow­ band Brillouin amplification. The output of the system is thus only dominated by two frequencies in the optical domain, separated by an arbitrary harmonic of the oscillator frequency. This technique is demonstrated for generation of up to 60 GHz millimeter-waves, limited mainly by the amount of phase-shift, that can be achieved in the modulator.

65

Paper C: “Microwave-Photonic Frequency Multiplication Utilizing Op­ tical Four-Wave Mixing and Fiber Bragg Gratings” A novel technique for optical multiplication of a microwave is presented in this paper. It utilizes optical four-wave mixing in a highly non-linear fiber, and the filtering properties of matched fiber Bragg gratings. In this experiment the multi­ plicator is driven electronically at 6.67 GHz and the created millimeter-wave has a frequency of 40 GHz. This technique is scalable, so that e.g. a 20 GHz input would generate a 120 GHz output. Furthermore, the quality of the signal was investigated by transmission of data over fiber with up to 2.5 Gbit/s using the same technique, which was introduced in Paper E. Paper D: “Tunable Terahertz Signal Generation by Chirped Pulse Photo Mixing” In this paper a method for the generation of signals in the GHz to THz range using photo-mixing of chirped optical pulses is presented. These signals can for example be used together with a broadband photodetector for THz generation. Pulses from a femtosecond fiber ring laser is temporally split up in two by using birefringence, and they are then chromatically dispersed, so that the two pulses overlap in time. Consequently, a single frequency pulse is ideally created, if the chirp is linear. The frequency of the pulse is easily changed with the amount of birefringence (differential group delay). The dispersive medium, a standard single mode fiber or a fiber Bragg grating, used in the experiment, has also higher order of dispersion and will therefore give a chirp of frequency in the pulse. Experimental evidence of frequency generation up to 725 GHz is presented. Analytical calculations agrees well with the experiment and verify that higher-order dispersion gives a frequency chirp of the generated pulse. Paper E: “Fiber Optic 40 GHz mm-Wave Link with 2.5 Gb/s Data Transmission” In Paper E a broadband millimeter-wave transmission link for downstream trans­ mission is investigated. The mm-wave carrier is created by a dual-frequency optical source based on suppressed-carrier double-side-band modulation. One of the op­ tical sidebands is modulated with baseband data up to 2.5 Gbit/s by separating the two sidebands using fiber Bragg gratings. At the base station a millimeterwave signal modulated with the data is created using heterodyne mixing of the two optical waves. This mm-wave signal is amplified and transmitted to the mobile unit, where the reception is performed using self-homodyne mixing. Error free data transmission was demonstrated for the downlink after both 22 and 44 km of optical fiber, which demonstrates that the setup is insensitive to chromatic dispersion.

66

Paper F: “Optical Demultiplexing of Millimeter-Wave Subcarriers for Wireless Channel Distribution Employing Dual Wavelength FBGs This paper focuses on sending several RF-carriers simultaneously through an opti­ cally carrier suppressed system and use the filtering properties of dual overwritten fiber Bragg gratings to optically demultiplex the signals. The different carriers are combined and modulated on a MZM, biased for carrier suppression, creating multi­ ple components in the upper and lower sidebands. To modulate data on one of the sidebands, fiber Bragg gratings are used to filter out the different components in one sideband and modulate them separately with independent data using MZMs. The signals are then propagated in the fiber to the optical demultiplexer close to the base stations, where the different carriers are demultiplexed and sent to the re­ spective base station. The concept is demonstrated using carriers at 20 and 40 GHz and the detected signals after demultiplexing shows less than -40 dB inter channel power leakage. Successful data transmission without crosstalk penalty from the other present channel, verifies the performance of the demultiplexer. Paper G: “Dispersion-Tolerant Millimeter-Wave Photonic Link Using Polarizarion Dependent Modulation” In this paper a method for modulating one sideband of a suppressed carrier, to be used in dispersion tolerant mm-wave links, is presented. The technique utilizes the polarization dependance of an optical phase modulator and the properties of photodetection to create a modulated mm-wave after transmission. This technique could be used in two settings to either create amplitude or phase modulation on the carrier. A theory which explains the technique is presented, as well as experimental verifications. Demonstrations are made for up to 2.5 Gb/s OOK data on a 40 GHz carrier transmitted over as long as 44 km optical fiber and over a short wireless link. Paper H: “Near-Baseband Synchronous Subcarrier Modulation” A new concept for subcarrier modulation, where the symbol rate and carrier fre­ quency are identical is presented in this paper. With this technique the signal is treated as a subcarrier signal with electrical modulation and electrical coherent detection, even though it is close to DC. The optical link can therefore be treated as a ”black box”. The technique is experimentally demonstrated using an arbitrary waveform generator to create symbol rates of 2.5 Gsymbols/s and modulation for­ mats such as QPSK, 16-QAM and 64-QAM. It is also proposed in the paper, that binary digital electronics can be used for creating the near-baseband signals and this is demonstrated with 2.5 Gbymbol/s for QPSK, 8-PSK and 16-PSK, which corresponds to bit rates of 5, 7.5 and 10 Gb/s, respectively.

67

Paper I: “Linearity and Noise Comparison of Uni-Traveling-Carrier­ and PIN-Photodiodes” In this paper two photo detectors are compared in the context of an analog link with a highly linear transmitter, and working at high photo currents to achieve high CNR. The compared diodes are an in-house fabricated UTC-PD and a commercial PIN-PD, with similar bandwidths. The behavior of the CNR and the linearity, in the form of the third order output intercept point and spurious-free dynamic range, is evaluated as a function of photo current. It is found that the increment of the CNR is only minor and limited by the excess noise from the optical amplifier. However, the OIP3 of the UTC-PD is substantially larger and increases with pho­ tocurrent, in contrast to the PIN-PD. Thus, even though the responsively of the UTC-PD is smaller than that of the PIN-PD, it is beneficial to use the UTC-PD in order to have superior dynamic range.

68

Appendices

Appendix A Three-wave propagation in an opti­ cal fiber The phase of a single optical frequency component propagating through a fiber is given by θ = βz, (A.1) where β is the propagation constant and z is the distance traveled. From this an expression for the phase dependance on wavelength can be derived [49], by expanding β with Taylor series around an optical frequency ν0 . Neglecting the constant phase, constant delay and higher order derivatives the phase angle θ is expressed as zλ20 D(ν − ν0 )2 , 2c where the dispersion parameter D is defined as θ=−

(A.2)

2πc (A.3) β2 , λ2 where the c is the speed of light, λ is an optical wavelength and β2 is the second derivative of the propagation constant with respect to frequency. Consider an amplitude modulated optical signal. The modulation generates upper and lower sidebands at frequency ±fm around the optical carrier, ν0 . If a simplified case is studied where only one upper and one lower frequency component is present, we will have three optical frequencies propagating in the fiber. The dis­ persion will induce a phase shift between the optical waves, which can be expressed by D=−

E

= |A0 | exp(j(2πf0 t + ϕ0 )) + |A1− | exp[j(2π(ν0 − fm )t + ϕ1−,0 )] + |A1+ | exp[j(2π(ν0 + fm )t + ϕ1+,0 )].

69

(A.4)

Evaluating the phase θ for the optical carrier and sidebands from Eq. A.2 gives � 0 θ= (A.5) zλ2 − 4πc0 D(2πfm )2 . If the optical signal (Eq. A.4) is propagated through a fiber, then the signal leaving the the fiber is E

= |A0 | exp (j2πf0 t + ϕ0 ) + |A1− | exp[j(2π(ν0 − fm )t + ϕ1− )] + |A1+ | exp[j(2π(ν0 + fm )t + ϕ1+ )],

(A.6)

where ϕ1− = ϕ1−,0 + θm and ϕ1+ = ϕ1+,0 + θm . The factor θm is the dispersion induced phase shift in the fiber and is defined in Eq. A.1. The detected light, given by Eq. A.7, is inducing a photocurrent, which is proportional the square of the optical wave (Sec.2.4), is given by � i ∝ �A0 + A1− exp [j − 2πfm t + ϕ1− ] �2 + A1+ exp [j2πfm t + ϕ1+ ]� = |A0 |2 + |A1+ |2 + |A1− |2 + 2 |A0 ||A1− | cos(2πfm t + ϕ0 − ϕ1− ) + 2 |A0 ||A1+ | cos(2πfm t − ϕ0 + ϕ1+ ) + 2 |A1− ||A1+ | cos(4πfm t − ϕ1− + ϕ1+ ).

(A.7)

The first terms are DC and correspond to the average detected optical power. The fourth and fifth terms are the cross products between carrier and sideband, i.e. ifm

∝

2 |A0 ||A1− | cos(2πfm t + ϕ0 − ϕ1−,0 − θm ) + 2 |A0 ||A1+ | cos(2πfm t − ϕ0 + ϕ1+,0 + θm ).

(A.8)

Here the dispersion induced phase shift, θm , will make the two components either to be in or out of phase depending on transmission distance. This will give rise to a periodical power fading with distance. The condition for complete extinction of the mm-wave carrier is π θ = n , where n = 1, 3, 5, . . . (A.9) 2 This condition is used with Eq. A.5 and the distance to first complete extinction is hence L1 = and the period length is ΔL =

c 2 2λ20 Dfm

c . 2 λ20 Dfm 70

(A.10)

(A.11)

The power fading can be avoided by only propagating two optical carriers in the fiber. The E-field of two optical components propagating in the optical fiber is described by E

= |A0 | exp(j(2πν0 t + ϕ0 )) + |A1 | exp[j(2π(ν1 )t + ϕ1 )].

(A.12)

The received photo current after propagation and detection is hence propor­ tional to i

� �2 ∝ �A0 exp [j2πν0 t + ϕ0 ] + A1 exp [j2πν2 t + ϕ1 ]� = |A0 |2 + |A1 |2 + 2 |A0 ||A1 | cos(2πfm t + ϕ0 − ϕ1 )

(A.13)

= |A0 |2 + |A1 |2 + 2 |A0 ||A1 | cos(2πfm t + ϕ0,0 − ϕ1,0 + θm ) where fm = ν1 −ν2 and θm is given by Eq. A.5. It is seen in the last term that there is no power fading and that the dispersion will only induce an absolute unimportant phase shift of the microwave carrier.

71

Appendix B

Power fading due to PMD

To show the impact of PMD, a derivation of the worst case scenario is exemplified below. Assume a situation with two optical frequencies separated by Δf . The polarization states of these two optical components can be described as � E1 in

= �

E2 in

=

cos θ1 sin θ1 ejφ1 cos θ2 sin θ2 ejφ2

� |A1 | exp [j2πν1 t + ϕ1 ]

(B.1)

|A2 | exp [j2πν2 t + ϕ2 ],

(B.2)

�

where θ (∈ {0, 2π}) and φ (∈ {0, π}) are angles which determine the polarization state. The polarization rotation induced by a birefringent fiber can be described by a Jones-matrix, if first order PMD is assumed, as � T(ν) =

cos α − sin α

sin α cos α

��

ej2πν Δτ /2 0

0 e−j2πν Δτ /2

��

cos α sin α

− sin α cos α

� , (B.3)

where Δτ is the DGD and the α the rotation angle of the DGD. We now examine the worst case, i.e. when the polarization of the incoming frequency components1 have an angle of π/4 compared to the birefringence axis in the fiber. The output polarization of the components can then be written, �

E1 out

1 = T(0) √ 2

E2 out

1 = T(Δf ) √ 2

1 1 �

� |A1 | exp [j2πν1 t + ϕ1 ] 1 1

(B.4)

� |A2 | exp [j2πν2 t + ϕ2 ]

(B.5)

where Δf = ν2 − ν1 . The introduced relative rotation, gives that component 1 seems to be constant while component 2 rotates. A photodetector at the end of the fiber receives the light through quadratic detection (see Sec. 2.4) and can be written as 1 The

two optical frequency components are assumed to have the same polarization at the input of the fiber.

72

I

∝

� � �E1,out + E2,out � 2

=

� � �E1,out,x + E2,out,x � 2

�2

� + �E1,out,y + E2,out,y �

=

� �2 1� j(2πν1 t+ϕ1 ) + ejπΔf Δτ |A2 | ej(2πν2 t+ϕ2 ) � 1| e 2 |A �2 � + 12 �|A1 | ej(2πν1 t+ϕ1 ) + e−jπΔf Δτ |A2 | ej(2πν2 t+ϕ2 ) �

=

|A1 |2 + |A2 |2 + |A1 ||A2 | cos[2πΔf (t + Δτ /2) + ϕ1 − ϕ2 ] + |A1 ||A2 | cos[2πΔf (t − Δτ /2) + ϕ1 − ϕ2 ]

=

|A1 |2 + |A2 |2 + 2|A1 ||A2 | cos[πΔf Δτ ] cos[2πΔf t + ϕ1 − ϕ2 ].

(B.6)

A power fading is induced as function of both Δf and Δτ , which is seen in the last term. The condition for total power extinction, i.e. perpendicular polarizations for E1,out and E2,out is n = 2Δf Δτ, where n = 1, 3, 5, . . .

73

(B.7)

Appendix C Optimum amplitude in heterodyne detection In a system with two optical frequencies mixed on a photodetector the photo current will be proportional to �2 � i ∝ �|A1 |ej(−2πν1 t+ϕ1 ) + |A2 |ej(2πν2 t+ϕ2 ) � = |A1 |2 + |A2 |2 + 2 |A1 ||A2 | cos(2πΔf t + ϕ1 − ϕ2 ),

(C.1)

where Δf = ν2 − ν1 . The following expression compares the RF-power out with the power of DC-component, K (|A1 |, |A2 |) ≡

2|A1 ||A2 | . |A1 |2 + |A2 |2

(C.2)

To maximize the RF-power power, Eq. C.2 is differentiated with respect to |A1 | and |A2 | and set equal to zero. The equation system is then solved and the result is ⎧ ⎪ ⎪ ⎨ ⎪ ⎪ ⎩

2 |A2 |(|A1 |2 +|A2 |2 )−4|A1 |2 |A2 | (|A1 |2 +|A2 |2 )2

=0

2 |A1 |(|A1 |2 +|A2 |2 )−4|A1 ||A2 |2 ∂K ∂|A2 | = (|A1 |2 +|A2 |2 )2

=0

∂K ∂|A1 |

=

⇔

|A1 | = |A2 |

(C.3)

Thus, to maximize the RF-power and get full modulation index the amplitude of the two components must be equal.

74

Appendix D Calculation of symbol-error-probability for M-ary modulation A theory for calculation of the probability of a symbol error, PM , for M-PSK, M­ PAM and M-QAM is found in [132] with additive white Gaussian noise present. The symbol error PM is usually calculated as a function of the SNR per symbol, γs , or SNR per bit, where γb = γs /log2 (M ) and M is the number of symbols. The calculation of probability of a bit error, Pb , for coherent and incoherent OOK is found in [133]. M-PSK The probability of a symbol error, PM , for M-PSK can be calculated as a function of SNR per symbol, γs , with a composite expression �

π/M

PM = 1 − −π/M

pΘr (Θr ) dΘr

(D.1)

where

� 1 −γs sin2 Θr ∞ −(V −√2γs cos Θr )2 /2 Ve e dV (D.2) 2π 0 where V and Θr are cylindrical coordinates. For high SNR per symbol (γs ), a simplified expression could be used �� π � (D.3) 2γs sin PM = 2 erfc M where the complementary error function (erfc) is defined in [132]. pΘr (Θr ) =

M-PAM The probability of a symbol error, PM , for M-PSK can be calculated as � �� � � 6γav 1 erfc PM = 2 1 − M M2 − 1

(D.4)

where γav is the average SNR per symbol. M-QAM The probability of a symbol error for M-QAM, with M = 2n , n ∈ N, is (D.5) PM = 1 − (1 − P√M )2 √ where P√M is the probability of an M -PAM with half the average power in each quadrature signal of an equivalent QAM system, and is � �� � � 3γ 1 av erfc (D.6) P√M = 2 1 − √ M −1 M

75

where γav is the average SNR per symbol. Coherent and incoherent OOK The probability of a bit error, Pb , are given [133] √ Pb = erfc ( γav ) for coherent OOK and

1 1 − 1 γav e 2 , γav > 2 4 is the average SNR per symbol.

Pb = for incoherent OOK, where γav

76

(D.7)

(D.8)

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Papers A– I

Paper A “Microwave Harmonic Frequency Generation Utilizing the Properties of an Optical Phase Modulator” P.O. Hedekvist, B-E. Olsson, and A. Wiberg J. Lightwave Technol., Vol. 22, No. 3, pp. 882–886, March 2004.

Paper B “Photonic Microwave Generator Utilizing Narrowband Brillouin Amplification and a Fiber-Based Oscillator” A. Wiberg and P.O. Hedekvist Presented at conference on Microwave and Terahertz Photonics,

Proceedings of SPIE Vol. 5466, Apr 29-30 2004, Strasbourg, France

Paper C “Microwave-Photonic Frequency Multiplication Utilizing Optical Four-Wave Mixing and Fiber Bragg Gratings” A. Wiberg, P. Per´ez-Mill´ an, M. V. Andr´es, and P.O. Hedekvist J. Lightwave Technol., Vol. 24, No. 1, pp. 329-334, January, 2006.

Paper D “Tunable Terahertz Signal Generation by Chirped Pulse Photomixing,” J. Stigwall and A. Wiberg IEEE Photon. Technol. Lett., Vol. 19, No. 12, pp. 931-933, 2007

Paper E “Fiber Optic 40 GHz mm-Wave Link with 2.5 Gb/s Data Transmission” A. Wiberg, P. Per´ez-Mill´ an, M. V. Andr´es, P. A. Andrekson and P.O. Hedekvist, IEEE Photon. Technol. Lett., Vol. 17, No. 9, pp. 1938–1940, September, 2005

Paper F “Optical Demultiplexing of Millimeter-Wave Subcarriers for Wireless Channel Distribution Employing Dual Wavelength FBGs” P. Per´ez-Mill´ an, A. Wiberg, M. V. Andr´es, P. O. Hedekvist Optics Communications, Vol. 275, No. 2, pp. 335-343, 2007.

Paper G “Dispersion-Tolerant Millimeter-Wave Photonic Link Using Polarization-Dependent Modulation” A. Wiberg, B-E. Olsson, P. O. Hedekvist and P. A. Andrekson, J. Lightwave Technol., Vol. 25, No. 10, pp. 2984-2991, October, 2007.

Paper H “Near-Baseband Synchronous Sub-Carrier Modulation” A. Wiberg, B-E. Olsson, and P. A. Andrekson submitted to IEEE Photon. Technol. Lett.

Paper I “Linearity and Noise Comparison of Uni-Traveling-Carrier- and PIN-Photodiodes” A. Wiberg, J. Vukusic, H. Sunnerud and P. A. Andrekson submitted to IEEE Photon. Technol. Lett.