Model Predictive Control With a Reduced Number of ... - IEEE Xplore

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Model Predictive Control With a Reduced Number of Considered States in a Modular Multilevel Converter for HVDC System Ji-Woo Moon, Student Member, IEEE, Jin-Su Gwon, Student Member, IEEE, Jung-Woo Park, Dae-Wook Kang, Member, IEEE, and Jang-Mok Kim, Member, IEEE

Abstract—This paper proposes a model predictive control (MPC) method for modular-multilevel-converter (MMC) high-voltage direct current (HVDC). To control the MMC-HVDC system properly, the ac current, circulating current, and submodule (SM) capacitor voltage are taken into consideration. The existing MPC methods for the MMC-HVDC system utilize weighting factors to configure the cost function in combinations of the SM capacitor voltage balancing algorithm, ac current control, and circulating current control. Because all combinations of the switch states are considered in order to minimize the cost function, their possible combinations increase geometrically according to the increase of the level of the MMC, which is a significant disadvantage. This paper proposes a new MPC method with a reduced number of states for ac current control, circulating current control, and the SM capacitor voltage-balancing algorithm. The proposed cost functions are divided into three types according to their control purposes. Each cost function determines the minimum number of states for controlling the ac current, circulating current, and SM capacitor voltage. The efficacy of the proposed controlling method is verified through simulation results using PSCAD/EMTDC. Index Terms—Circulating current, model predictive control (MPC), modular multilevel converter (MMC), unbalance voltage.

I. INTRODUCTION

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URRENTLY, investment and research on high-voltage direct-current (HVDC) systems have been actively conducted and expanded to improve the efficiency and reliability of electric power generation through large-capacity power transmission and linkage among different networks [1]–[6]. The HVDC system is divided into two types: 1) the current-source converter HVDC(CSC-HVDC) system and 2)

Manuscript received August 20, 2013; revised November 11, 2013 and December 25, 2013; accepted January 25, 2014. Paper no. TPWRD-00947-2013. J.-W. Moon and J.-S. Gwon are with the Department of Electrical Engineering, Pusan National University, Pusan 609-735, Korea, and also with Power Conversion and Control Research Center, Korea Electrotechnology Research Institute, Changwon 642-120, Korea (e-mail: [email protected]; [email protected]). J.-W. Park and D.-W. Kang are with Power Conversion and Control Research Center, Korea Electrotechnology Research institute, Changwon 642-120, Korea (e-mail: [email protected]; [email protected]). J.-M. Kim is with the Department of Electrical Engineering, Pusan National University, Pusan 609-735, Korea (e-mail: [email protected]). Color versions of one or more of the figures in this paper are available online at http://ieeexplore.ieee.org. Digital Object Identifier 10.1109/TPWRD.2014.2303172

Fig. 1. Basic structure of the MMC.

the voltage-source converter HVDC (VSC-HVDC) system. In particular, the interest in the VSC-HVDC system is increasing because of the need for active control for renewable energy systems. Compared to the CSC-HVDC system, the VSC-HVDC system can control active power and reactive power independently; it reduces the ac filter size and has a fast transient response because of the pulsewidth-modulated (PWM) method. Furthermore, there is no need to use a transformer for the commutation process [3]–[8]. A modular multilevel converter (MMC) has been widely used for the VSC-HVDC system. Despite the need for a reactor to suppress the circulating current, an MMC has the merits of extending to hundreds of voltage levels because of its simple circuit structure and ease of modularization. For that reason, many studies have been carried out on an MMC as the most suitable structure for the VSC-HVDC system [4]–[12]. Fig. 1 shows a three-phase MMC with six arms. In the case of levels, each arm is composed of half-bridge submodule (SM). An MMC possesses a voltage difference between the total SM capacitor voltage and the dc-link voltage, which causes circulating currents to flow through the arms. Thus, a control method is necessary to suppress the circulating current. And an SM voltage-balancing algorithm is also necessary [1]–[4], [22]. The MPC method is gradually adopted as a control method of power converters because of the development of processor speed and characteristics, such as fast response, easy inclusion of nonlinearities, flexibility of various goals, and the availability of simple modulation techniques. Many papers have studied a finite MPC method to control the power converter. In the MPC

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method, first, the future values are predicted on the basis of the switch states, and then the final switch state is decided in a way to minimize the cost function. The cost function is designed according to the control goal [13]–[21]. It may consist of several subsets of cost functions through the weighting factor [13]–[16], which may achieve various goals of control simultaneously. Reference [13] shows that the cost function may include a control goal, such as switching frequency reduction, common-mode voltage reduction, reactive power reduction, and current ripple reduction in order to control the power converter. The MPC method can improve total harmonic distortion (THD) and transient characteristics in comparison with a proportional-integral (PI) or proportional-resonant (PR) controller [21]. Many studies have been carried out on the application of the MPC method to multilevel converters, such as an MMC. Unlike two-level converters, multilevel converters have problems, such as the capacitor voltage unbalancing and the circulating currents. Hence, the MPC method for multilevel converters must use a cost function to solve their problems as well as the ac-side current control. Reference [9] proposed an MPC method for the cascaded H-Bridge (CHB) multilevel converters. The cost function was designed to control the ac-side current. To reduce the number of available states, the redundant voltage vectors were eliminated by choosing those that generated the minimum common-mode voltage. However, this MPC method did not consider capacitor voltage balancing. Reference [15] proposed a method of controlling an MMC using MPC. The proposed control method applied the cost function to control the ac current, circulating current, and SM capacitor voltage balancing. However, this MPC method has to consider the switching state of when the number of SMs of each arm is . This paper proposes an MPC method with a reduced number of considered states for ac-side current control, circulating current control, and capacitor voltage-balancing control for an MMC. The proposed method consists of three MPCs, and each cost function is designed for three different control goals. The first cost function is for the control of only the ac-side current without considering redundancy. The second cost function is for control of the dc-link current ripple, the transient characteristics of the unbalanced voltage condition, and the circulating current. The last cost function is for control of the capacitor voltage balancing and reducing the switching frequency of the SM. The proposed control method reduces the required states in each cost function and controls an MMC in a stable manner. In addition, the proposed control method can control zero-sequence current with a double-line frequency in the circulating current under unbalanced voltage conditions. Further, transient characteristics are enhanced through direct control of the dc-link current. The effectiveness of the proposed MPC method was verified through simulation results using PSCAD/EMTDC. II. PROPOSED MPC STRATEGY OF THE MMC As mentioned in the previous section, the proposed MPC method has three control goals, and the cost functions are designed according to them. They are as follows: ac-side current

Fig. 2. Single-phase equivalent circuit of the three-phase MMC.

control, circulating current and dc-link current control, and SM capacitor voltage balancing and switching frequency reduction. Generally, the three subsets of cost functions can be combined into one cost function through the weighting factor in the MPC method [13]–[21]. However, the proposed MPC method is divided into three stages according to the control goals, and the cost function in each stage is designed separately without the weighting factor. A. MPC Strategy for the AC-Side Current Fig. 2 shows the single-phase equivalent circuit of the threephase MMC. Here, and ( , , ) are the upper-arm and lower-arm currents, respectively, where is the inner unbalanced current; is the grid-side voltage; and and are the upper-arm and lower-arm voltages, respectively. From Fig. 2, the voltage equation of the MMC is given by (1) [1], [2], [4]. Here, is the converter output voltage, which is defined as (1) (2) The voltage equation of the MMC can be rewritten as (3), where and . The ac-side current is deduced in (4) using the Euler approximation for the current derivative [15]. Here, is the sampling period (3)

(4) If the ac-side current reference is is designed as

, the cost function (5)

Assuming that the number of arms in the SM is , and the SM capacitor nominal voltage is , the converter output voltage reference is expressed as shown in (6), depending on the number of switch states per phase. Here, and

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imply the switch states and return 0 or 1 for turn-off or turn-on, respectively

(6) In the case of levels, the combination of all switch states per phase is , but the available number of combinations of them is reduced to because the sum of turn-on switches per phase should always be [15]–[17]. Nevertheless, this method has to consider too many combinations of switch states. If the number of SMs increases, it is difficult to implement the conventional method. Therefore, a method is required to reduce this high number of switch combinations for practical use. In this paper, the value of in (6) is determined according to the output voltage level, not according to the available combinations of switch states, and is expressed as Fig. 3. Block diagram of the ac-side current MPC strategy.

(7) Therefore, the presentable voltage level is determined by (8), . and the number of statuses to be determined is reduced to The upper-arm and lower-arm voltage-level references are defined in (9) and (10) using determined by the cost function. When the MMC has levels, the presentable voltage-level reference of the upper arm and lower arm is expressed by (11). Fig. 3 shows the proposed block diagram of the ac-side current MPC strategy. However, the control method in this case needs to consider the balancing of the SM capacitor voltage and circulating current

(8)

and are determined based on the cost function in (5), (9), and (10)

(14) In (2), is determined by the difference between the lower-arm and upper-arm voltages. Thus, the same voltage level added to or subtracted from and has no significant effect on the ac current of the MMC. Hence, the voltage level can be added to (14) for inner unbalanced current control, as shown

(9) (10) (11)

B. MPC Strategy for the Circulating Current and DC-Link Current The inner unbalanced current and voltage equations are expressed as shown in (12) and (13) [4], respectively (12) (13) The inner unbalanced current is derived from the Euler approximation of the current derivative in (14), where

(15) If two voltage levels are allocated to to control the inner unbalanced current, the number of considered statuses is three, as shown in (16). The voltage level of may be expanded according to the value and the control characteristics of the circulating current (16) The inner unbalanced current of the MMC is defined in (17) [4], [10], where is a dc component of the inner unbalanced current, and is the circulating current. The circulating current has a negative-sequence component double-line frequency under the balanced voltage condition, and it needs to be removed because the circulating currents increase the arm currents and



power loss [4], [22]. Thus, the reference to control the circulating current is , and the cost function can be designed by (17) (18) However, under unbalanced voltage conditions, the circulating current involves not only the negative-sequence component, which is the double-line fundamental frequency, but also the positive- and zero-sequence components [2], [22]. Since the zero-sequence component rotates with the same phase component in each phase, a current ripple appears in the dc-link current. Thus, zero-sequence ripple is included in of the inner unbalanced current reference in (18). When the cost function is applied in (18), it is difficult to control the zero-sequence circulating current [22]. Furthermore, the unbalanced voltage causes a transient ripple in the dc-link current. Hence, the cost function is designed to control the dc-link current ripple due to transient characteristics as well as the zero-sequence component of the inner unbalance current under the unbalanced voltage conditions. Assuming the MMC active power is controlled without ripple, and there is no loss in the MMC, the MMC’s ac-side active power and dc-side active power are given as (19) and the dc-link current reference by (20) [22]. Thus, the cost function is designed by

Fig. 4. Block diagram of the circulating and dc-link current MPC strategy.

(19) (20) (21) The voltage-level references of the final upper arm and lower arm are given as (22) and (23), respectively. The presentable voltage level is expressed by (24). Fig. 4 shows the proposed MPC strategy for the circulating current and dc-link current (22) (23) (24)

C. MPC Strategy for SM Capacitor Voltage Balancing and Switching Frequency Reduction 1) Proposed Balancing and Switching Frequency Reduction Method 1: The SM capacitor voltage depends on the switch status of the SM and arm current [4], [15]. The upper and lower arm voltage-level references in (22) and (23), respectively, are equivalent to the sum of the output voltage of the SM turned on in each arm. Thus, the voltage-level reference can be rewritten in terms of the turn-on number of the SM, as shown

Fig. 5. Block diagram of the SM voltage balancing and switching frequency reduction MPC strategy 1.

The SM capacitor voltage is defined in (27) and (28), depending on whether the SM is turned on or off [15]. Here, or , with (27) (28)

(25)

The SM capacitor voltage reference is . When the cost function is designed as in (29) in [15], the number of considered statuses increases to and .

(26)

(29)


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TABLE II MAIN CIRCUIT PARAMETERS

Fig. 6. Block diagram of the SM voltage balancing and switching frequency reduction MPC strategy 2.

TABLE I NUMBER

OF CONSIDERED STATUSES OF THE 11-LEVEL-MMC CONVENTIONAL MPC AND PROPOSED MPC

Fig. 8. Gate signal simulation results for the different SM voltage-balancing algorithms: (a) proposed control method without considering the switching frequency, (b) proposed control method 1, and (c) proposed control method 2.

Fig. 7. System structure of simulations.

Equation (30) is a cost function that has been newly designed. One SM is selected at each loop, and the loop is repeated according to the number to determine turn-on status of the SM. Thus, the number of statuses to be considered is the total number of SMs multiplied by the number of SMs to be turned on and is thus expressed as and . However, the total number of SMs turned on in the upper arm and lower arm should be always [15]–[17]. When the number of SMs turned on in the upper arm is defined as , the number

Fig. 9. Mean loss of the IGBT and diode parts(ABB-5SNA 120E330100) in a-phase SM1 for the different SM voltage-balancing algorithms: (a) proposed control method without considering the switching frequency, (b) proposed control method 1, and (c) proposed control method 2.

, and that of statuses of the upper arm to be considered is . Thus, the number of considof the lower arm is ered statuses of the upper arm (30) and lower arm are reduced to

.



Fig. 10. Simulation results under the unbalanced voltage conditions: (a) grid-side voltage, (b) ac-side current, (c) – axis circulating current, (d) a-phase inner unbalanced current, (e) dc-link current, (f) ac-side active power, and (g) SM capacitor voltage. (A) Conventional control method [4] and (B) proposed MPC method.

However, in the case of (30), the cost function may be different depending on whether the SM capacitor is recharged or discharged. When it is recharged, the minimum value of the cost function is selected, and when it is discharged, the maximum value of the cost function is selected. Thus, the cost function in (30) is modified to (31) to select the turn-on status of the SM depending on the minimum value of the cost function when it is recharged or discharged. Here, has a positive value when it is recharged and a negative value when it is discharged. Hence, the turn-on status of the SM is always decided on the basis of the minimum value of the cost function (31) When only capacitor voltage balancing is considered for the SM turn-on status, the SM to be switched at every control period is changed, which results in increasing the switching frequency. So the reduced switching frequency modulation method is required [4], [25], [26]. Therefore, an additional cost function is added to (31) for reduced the switching frequency. Equation (32) is the proposed cost function to reduce the switching frequency and capacitor voltage balancing. The cost function is defined on the basis of the weighting factor in comparison to the previous switch status and the next switch status. Fig. 5 shows the block diagram of the proposed MPC strategy for SM

capacitor voltage balancing and switching frequency reduction. For the proposed MPC method for the MMC, the number of statuses to be considered includes statuses for ac current control, three statuses for circulating current and dc-link current control, and statuses for SM capacitor voltage balancing and switch frequency reduction. For a comparison of the conventional MPC method [15] with the proposed MPC method status based on an 11-level MMC, the number of statuses to be considered to control a single leg with the conventional MPC method is , whereas the proposed MPC method includes 11 statuses for ac current control, 3 for circulating current and dc-link current control, and 100 for the SM capacitor voltage-balancing method, which is only 114 statuses in total. The number of considered statuses decreases compared to the conventional MPC method

(32) 2) Proposed Balancing and Switching Frequency Reduction Method 2: Similar to the proposed balancing and switching frequency reduction method 1, the number of statuses to be considered is . Since the number of arms in the SM increases, the number of statuses to be considered increases accordingly. Thus, an improved algorithm is proposed in this section to reduce the number of considered statuses, capacitor voltage bal-


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ancing, and reduced switching frequency modulation method [4], [25]. The SMs to be turned on are selected only when there is a change in the SM turn-on state to reduce the switching frequency. The variation in the SM turn-on state is defined as

(33) Since the existing switching state needs to remain to reduce the switching frequency, the cost function in (31) is used. When , an additional SM needs to be turned on. Thus, an SM whose switching state is off is selected, and then the SM to be turned on is selected based on the cost function. Basically, the SM whose cost function is at the minimum value needs to be selected. One SM is selected at one time for the application of proposed method 1, and the loop is repeated times to determine the SM turn-on status. In the same method, when , an additional SM is turned off. However, in this case, the SM whose cost function is at the maximum value is selected. Thus, the number of statuses to be considered is reduced to and . In general, the application of a PI controller involves the change range of from 0 to 1 [4], but in the case of the MPC, whose switching status is determined on the basis of the voltage level, is changed to three in the simulation. Fig. 6 shows the block diagram of the proposed MPC strategy for SM capacitor voltage balancing and switching frequency reduction. Table I shows the number of considered statuses of the 11-level MMC conventional MPC and proposed MPC. For the conventional MPC method [15], the number of considered statuses is 184 756 in total, but when the proposed MPC method 1 is adopted, the number of considered statuses is reduced to 114. There is approximately a 99.93% reduction in the total number of considered statuses. In the proposed MPC method 2, the number of considered statuses is approximately 14–74. When the max value is applied, there is approximately a 99.96% reduction in the total number of considered statuses. III. AC-SIDE CURRENT REFERENCE Under balanced voltage conditions, ac-side – current references according to the active power and reactive power references are shown in (34) and (35). Under unbalanced voltage conditions, the active power and reactive power are separated as the positive and negative sequences, which are shown in (36) [22], [24]. Here, the total active power of a grid is represented as (37)

Fig. 11. Simulation results for active power reversal demand: (a) grid-side voltage, (b) ac-side current, (c) ac-side active power, (d) aca-side reactive power, (e) a-phase inner unbalanced current, (f) – axis circulating current, (g) dc-link current, and (h) SM capacitor voltage.

components, respectively. Subscript g denotes the grid component; 0 denotes the dc component; and denote the d-axis and q-axis components in the rotational reference frame, respectively; and and denote the double line-frequency component, respectively

(34) (36) (35) In (37), the total active power involves ripple with a doubleline frequency, where is the active power, is the reactive power, is the grid-side current, and is the grid-side voltage. The superscripts of and are positive- and negative-sequence

(37) To supply active power constantly without ripple to the grid and need to be controlled to zero. If theta side, is determined such that 0 through a phase-locked loop



(PLL) in (36), the negative-sequence current references to control and to 0 are expressed in (38) and (39), respectively. The current reference of each phase is obtained through the coordinate transformations of (34), (35), (38), and (39) [22], [24] (38) (39)

IV. SIMULATION RESULTS A simulation was carried out using PSCAD/EMTDC, and Fig. 7 shows the structure of the simulated system. The main circuit parameters and operating conditions are listed in Table II. Fig. 8 shows the a-phase SM1 PWM simulation result for the application of the SM capacitor voltage balancing algorithm. where is the average switching frequency of SM. Fig. 8(a) shows the PWM simulation results for the cost function in (31). Since the switching frequency reduction is not taken into consideration, the PWM changes depending on the control period, and the switching frequency is high. Fig. 8(b) shows the PWM simulation results with the proposed balancing and switching frequency reduction method 1. In comparison with Fig. 8(a), the switching frequency is reduced. Fig. 8(c) shows the simulation results with the proposed balancing and switching frequency reduction method 2. Although the number of statuses to be considered is reduced, the switching frequency decreases in comparison with Fig. 8(a) and (b). Fig. 9 shows the mean loss of the insulated-gate bipolar transistor (IGBT) and diode in a-phase SM1 according to the SM voltage balancing and reduced switching frequency method, where and are the upper and lower IGBT. and are the upper and lower diodes. is conduction loss, is switching loss, and is the diode reverse-recovery loss. The parameters of the IGBT modules are obtained from the datasheet of the 5SNA 1200E330100 from ABB [27]. And the losses evaluation method is used in [28]. In order to simplify the loss calculation, junction temperature is considered to be 125 and the current factor is set to 1 [4], [28]. Active power is 4 MW and reactive power is 0 MVA. The conduction loss is almost the same in three control methods. However, in the proposed control method1 and 2, there is a decrease in the IGBT switching loss and diode reverse-recovery loss due to the reduction of the switching frequency. Fig. 10 shows the simulation results under the unbalanced voltage conditions. Fig. 10(a) shows the conventional control method [4], and Fig. 10(b) shows the proposed MPC method. The SM capacitor voltage-balancing algorithm adopts proposed method 2. The ac-side positive -axis current reference is 0 A, and the positive -axis current reference is 285 A, corresponding to 4 MW at a normal voltage. To apply unbalanced voltage conditions, the a-phase line-to-ground fault is applied at 0.8 s. Fig. 10(a) and (b) shows the system voltage and ac-side current of the MMC. In [4], an unbalanced voltage is not considered, and each phase current is controlled to a three-phase equilibrium. However, ac-side negative-sequence current is injected in the proposed MPC method to reduce the active power ripple.

Thus, each phase current is unbalanced. Fig. 10(c) shows the circulating current in a rotational reference frame. In [4], the circulating current element is not completely reduced. In contrast, the proposed MPC method controls the circulating current to zero, even under the unbalanced voltage conditions. Fig. 10(d) and (e) shows the a-phase inner unbalanced current and dc-link current, respectively. At the point of applying unbalanced voltage, an extensive amount of current ripple occurs because of the transient characteristics, but the proposed MPC method stably controls it without any transient characteristics. Fig. 10(f) shows the ac-side active power of the MMC. Since [4] does not consider the unbalanced voltage condition, the MMC’s ac-side active power ripple cannot be reduced, so the double-line frequency ripple is generated under active power. In contrast, the proposed MPC method applies a negative-sequence current to reduce the active power ripple, and constant active power is supplied without ripple. However, the magnitude of the active power is reduced by a negative-sequence’s active power component. Fig. 10(g) shows the SM capacitor voltage. Under the unbalanced voltage condition, [4] involves a great deal of capacitor voltage ripple because of the inner unbalanced current transient ripple, whereas the proposed MPC method involves no transient characteristics. Fig. 11 shows the simulation results for the proposed MPC method for a rapid change of load. The proposed MPC method maintained zero power control up to the point of 0.4 s, where a load of 4 MW was admitted to the ramp. The load of 4 MW was admitted at 0.7 s to analyze the characteristics of the proposed MPC method depending on the load change. The reactive power was controlled to 0 MVA. Fig. 11(a) and (b) shows the system voltage and ac-side current of the MMC, respectively. The ac-side current is stably controlled without ripple despite a rapid change in load. Fig. 11(c) and (d) shows the ac-side active power and reactive power, respectively. The active power value increased to 4 MW depending on the load reference and then maintained a stable value of 4 MW. Fig. 11(e) shows the a-phase inner unbalanced current, and Fig. 11(f) shows the – axis circulating current. As the load fluctuated, the inner unbalanced current maintained stable control at . The – axis circulating current was controlled to 0 regardless of the load fluctuation. Fig. 11(g) shows the dc-link current. Since the load was changing rapidly, the dc-link current was likely to involve transient ripple, but the proposed MPC method controls the dc-link current without transient ripple. Fig. 11(h) shows the SM capacitor voltage of the a-phase upper arm. Regardless of the load fluctuation, the SM’s capacitor voltage balancing is stably controlled. V. CONCLUSIONS This paper proposes the MPC method with a reduced number of considered states for the MMC-HVDC system. To control the MMC, three cost functions were adopted: first, a cost function to control ac-side current; second, a cost function to control the inner unbalanced current and DC-link current; and third, a cost function for SM capacitor voltage balancing and switch frequency reduction. The proposed MPC method minimizes the number of statuses to be considered by means of the cost functions with the MMC stably controlled. The proposed


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[22] J.-W. Moon, C.-S. Kim, J.-W. Park, D.-W. Kang, and J.-M. Kim, “Circulating current control in MMC under the unbalanced voltage,” IEEE Trans. Power Del., vol. 28, no. 3, pp. 1952–1959, Jul. 2013. [23] Z. Yuebin, J. Daozhuo, G. Jie, H. Pengfei, and L. Zhiyong, “Control of modular multilevel converter based on stationary frame under unbalanced AC system,” in , Proc. 3rd Int. Conf. ICDMA, 2012, pp. 293–296. [24] L. Xu and Y. Wang, “Dynamic modeling and control of DFIG based wind turbines under unbalanced network conditions,” IEEE Trans. Power Syst., vol. 22, no. 1, pp. 314–323, Feb. 2007. [25] J. Qin and M. Saeedifard, “Reduced switching-frequency voltage-balancing strategies for modular multilevel HVDC converters,” IEEE Trans. Power Del., vol. 28, no. 4, pp. 2403–2410, Oct. 2013. [26] G. T. Son, H.-J. Lee, T. S. Nam, Y.-H. Chung, U.-H. Lee, S.-T. Baek, K. Hur, and J.-W. Park, “Design and control of a modular multilevel HVDC converter with redundant power modules for noninterruptible energy transfer,” IEEE Trans. Power Del., vol. 27, no. 3, pp. 1611–1619, Jul. 2012. [27] ABB HiPak, Data Sheet: IGBT Module 5SNA 1200E330100. 2012. [Online]. Available: http://www.abb.com [28] S. Rohner, S. Bernet, M. Hiller, and R. Sommer, “Modulation, losses, semiconductor requirements of modular multilevel converters,” IEEE Trans. Ind. Electron., vol. 57, no. 8, pp. 2633–2642, Aug. 2010.

Ji-Woo Moon (S’13) was born in Pusan, Korea, in February 1981. He received the B.S. and M.S. degrees in electrical engineering from Dong-A University, Pusan,, in 2006 and 2008, respectively, and is currently pursuing the Ph.D. degree in electrical engineering at Pusan National University, . Since 2008, he has been with KERI, Changwon, Korea. His research interests include control of HVDC, wind energy generation, and the application of power electronics to power systems.

Jin-Su Gwon (S’13) was born in Masan, Korea, in February 1983. He received the B.S. and M.S. degrees in electronic engineering from Kyungnam University, Masan, Korea, in 2009 and 2011, respectively, and is currently pursuing the Ph.D. degree in electrical engineering at Pusan National University, Pusan, Korea.. Since 2009, he has been with KERI, Changwon, Korea. His present interests include control of HVDC, multilevel converters, and renewable energy.

Jung-Woo Park was born in Chungnam, Korea, in February 1963. He received the B.S. and M.S. degrees in electronic engineering from Chungnam National University, Chungnam, Korea, in 1986 and 1988, respectively, and the Ph.D. degree in electrical engineering from Kyungpook National University, Kyungpook, Korea. Since 1998, he has been a Principal Researcher with KERI, Changwon, Korea. His present interests include control of HVDC, multilevel converters, wind energy generation, and renewable energy.

Dae-Wook Kang (S’99–M’04) received the B.S., M.S., and Ph.D. degrees in electrical engineering from Hanyang University, Seoul, Korea, in 1998, 2000, and 2004, respectively. Since 2004, he has been a Senior Researcher with KERI, Changwon, Korea. His interests include control of HVDC, multilevel converters, renewable energy, and the application of power electronics to power systems.



Jang-Mok Kim (M’01) received the B..S. degree in electrical engineering from Pusan National University (PNU), Pusan, Korea, in 1988, and the M.S. and Ph.D. degrees in electrical engineering from Seoul National University, Seoul, Korea, in 1991 and 1996, respectively. From 1997 to 2000, he was a Senior Research Engineer with Korea Electric Power Research Institute. Since 2001, he has been with the School of Electrical Engineering, PNU, where he is currently a Research Member of the Research Institute of Computer Infor-

mation and Communication, a Faculty Member, and a Head of LG Electronics Smart Control Center, Changwon, Korea. As a Visiting Scholar, he joined the Center for Advanced Power Systems (CAPS), Florida State University, Tallahassee, USA, in 2007. His current interests include the control of electric machines, electric-vehicle propulsion, and power quality.