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UNIVERSITE DE LIMOGES ECOLE DOCTORALE Sciences – Technologie - Santé

FACULTE des SCIENCES et TECHNIQUES de LIMOGES Année 2006

N° ordre : - 2006

THESE Pour obtenir le grade de DOCTEUR DE L’UNIVERSITE DE LIMOGES Discipline : Electronique des Hautes fréquences et Optoélectronique Spécialité : "Communications Optiques et Microondes"

Cristiano PALEGO Le Vendredi 19 janvier 2006

Composants MEMS RF pour les têtes de réception RF reconfigurables Thèse dirigée par Pierre BLONDY Roberto SORRENTINO

Professeur à Università di Perugia, Italie

Président

Jean-Louis CAZAUX Nathalie ROLLAND

Ingénieur à ALCATEL ALENIA Space, Toulouse Maître de Conférences à IEMN, Lille

Rapporteur Rapporteur

James C.M. HWANG Pierre BLONDY Valérie MADRANGEAS Thierry MONEDIERE Arnaud POTHIER

Professeur à Lehigh University, Bethlehem, USA Ingénieur de Recherches CNRS, XLIM – Limoges Professeur à l’Université de Limoges – XLIM Professeur à l’Université de Limoges – XLIM Chargé de recherches au CNRS, XLIM – Limoges

Examinateur Examinateur Examinateur Examinateur Examinateur

Christine ZANCHI Xavier GRISON

Ingénieur au CNES – Toulouse Ingénieur à la DGA – Paris

Invitée Invité

Acknowledgements I would like to thank my thesis advisor Pierre Blondy who greatly inspired me throughout my way, with his outstanding insight into research. A very few supervisors are talented in obtaining high quality work in a short time while keeping an extraordinarily liberal attitude and Pierre is one of them. His far-sighted advices often bordering on intellectual temerity, always excited and enriched me. I wonder if he realizes how much I have learned from him in our multiple, virtually zero time consumption discussions. Those discussions were among the factors that made a several years stay away from home worth to be lived and I will miss them inconsolably. I would like to thank the members of my dissertation committee: Roberto Sorrentino, Jean-Louis Cazaux, Nathalie Roland, James Hwang, Pierre Blondy, Valérie Madrangeas, Thierry Monedière, Arnaud Pothier, Christine Zanchi and Xavier Grison for taking the time to attend my defense. A special thanks goes to Professor Roberto Sorrentino who accepted to serve as Chair on my doctoral committee. He remains an unforgettable master of knowledge and style, and I am proud to be numbered among the members of his “offspring”. A special thanks also goes to Jean-Louis Cazaux at Alcatel Alenia Space and Nathalie Roland at IEMN, who dared to serve on the reading committee and to Professor James Hwang at Lehigh University who braved the Atlantic Ocean to attend my defense. I am greatly indebted to Arnaud Pothier for a variety of reasons: for his constant, precious and not owed support, for carefully listening and perceptively replying to any idea of mine and, most important, for giving the impression that everything I undertook was within my capabilities. My appreciation for Arnaud’s technical competences is very deep. My admiration for his human qualities is probably even deeper. I would like to acknowledge the director of Minacom department at XLIM, Professor Serge Verdeyme, for generously accepting me in the group and for his always helpful availability.

I am also very thankful to the Minacom secretary Ms Marie-Laure Guillat for her kind and effective assistance when most needed. There are many people that helped me along the way and I am particularly grateful to those of them who offered hospitality empathy and friendships when I first arrived in Limoges. Finally I would like to express my deepest gratitude to my family, to my friends right here or faraway and to everybody who contributed to make sure that this thesis was not just a thesis.

INTRODUCTION GENERALE

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Introduction générale

Les années 90 ont marqué un profond changement dans l’univers des technologies hyperfréquences qui est en partie imputable aux événements économiques et géopolitiques de la fin du dernier siècle. Notamment l’avènement de l’ère de l’information à crée à la fois un intérêt croissant et un marché à l’échelle mondiale pour les systèmes de communications et les réseaux de données multimédia. L’essor fulgurant vers les dispositifs de communication individuelle et la conséquente démocratisation des technologies électroniques ce sont en général traduites par une reconversion des systèmes de communication classiques, centralisés, encombrants et à fort consommation de puissance, vers une multitude de systèmes distribués de taille et consommation réduites. Les ressources d’énergie étant limitées, les contraintes de consommation de ces nouveaux systèmes deviennent encore plus sévères lorsque aux caractéristiques mentionnées s’ajoute la mobilité ou même la portabilité. C’est pourquoi il existe actuellement un besoin pressant de technologies hyperfréquences à faible consommation de puissance, avec un faible encombrement et capables d’atteindre une fonctionnalité supérieure par unité de volume tout en gardant des performances électriques satisfaisantes jusqu’aux fréquences millimétriques. Dans les 20 dernières années, un important effort de recherche à été mené dans le développement de circuits hyperfréquences intégrés (RFIC) sur Silicium, Silicium-Germanium et Arséniure de Gallium. Les remarquables progrès dans la fabrication de transistors bipolaires CMOS (technologie BiCMOS sur Si et SiGe) et de transistors bipolaires à hétéro-jonction (technologie HBTsur AsGa) ont ainsi abouti à une prolifération de solutions disponibles pour les RFIC avec un faible coût de réalisation. Cependant il semble actuellement difficile de déterminer une technologie hyperfréquence dominante car les contraintes varient énormément suivant les systèmes et les circuits et la plupart des technologies consolidées à l’heure actuelle offrent des capacités de portabilité et modularités modestes. La technologie MEMS (pour “Micro Electro Mechanical Systems”: systèmes micro-électromécaniques) s’est au contraire développée au cours des 3

dernières années et laisse entrevoir des potentialités prometteuses pour le développement de systèmes intégrables à la fois performants et reconfigurables. Cette technologie exploite des techniques de la microélectroniques compatibles avec la fabrication des RFIC classiques pour réaliser une nouvelle classe de microsystèmes qui combinent des propriétés mécaniques et électromagnétiques intéressantes. Ces systèmes sont également caractérisés par une faible consommation de puissance et un comportement en fréquence très linéaire. La possibilité de développer de nouvelles architectures intelligentes avec de très bonnes performances et un coût raisonnable, que ce soit en termes de puissance, de taille ou de complexité de système, a suscité l’effervescence des communautés scientifique et industrielle. Le potentiel de cette technologie à été démontré par un nombre impressionnant de prototypes dans le milieu universitaire/militaire et cela a répandu le sentiment que le saut technologique vers une production à grande échelle est désormais possible ou imminent. Néanmoins il reste encore d’importants progrès à réaliser avant d’achever cette évolution, notamment en ce qui concerne le packaging, la fiabilité à long terme et la tenue en puissance de ces composant MEMS. Le travail de thèse présenté dans ce mémoire s’inscrit dans ce contexte. Notre objectif a été de contribuer au développement de nouvelles topologies de circuits microondes reconfigurables et également d’apporter de solutions pour améliorer la fiabilité et la durée de vie de composant MEMS. L’originalité de ce travail se situe dans la superposition des plans technologiques (optimisation des performances des dispositifs) et théoriques (synthèse de systèmes reconfigurables). En effet notre effort de recherche ne se résume pas à essayer de repousser les limitations techniques des composants MEMS pour égaler (ou améliorer) les prestations et les fonctions hyperfréquences typiquement assurées par les semi-conducteurs. Il se propose aussi de montrer comme l’intégration des composant MEMS directement au niveau de la synthèse offre une nouvelle classe de fonctionnalités autrement inaccessibles. Ce manuscrit se construit de la manière suivante : Dans un premier chapitre, nous proposerons un tour d'horizon des principaux dispositifs issus de l’évolution de la technologie MEMS. Celle-ci c’est en effet avérée très attractive pour les domaines d’applications les plus variés (les accéléromètres et les capteurs en général, l’optique, le biomédical, les systèmes de stockage d’information et très largement, les télécommunications). Les dix dernières années ont connu une prolifération extraordinaire de dispositifs, systèmes ou projets intégrants des composants MEMS, par de nombreux 4

organismes de recherche et groupes industrielles. Nous montrerons ainsi quelques exemples qui ont déjà atteint une importance commerciale (Analog Devices) ou sont en état de développement avancé (NASA, IBM). Nous nous intéresserons plus particulièrement au domaine des hyperfréquences car la contribution de la technologie MEMS à ce secteur est majeure notamment grâce au développement de plusieurs micro-commutateurs très performants. Les applications des commutateurs MEMS aux dispositifs à micro-ondes et millimétriques sont très nombreuses et semblent être parvenue à un stade de maturité relative. Ceci justifie le développement d’un domaine spécifique dit des MEMS pour les radiofréquence (RF MEMS). Nous présenterons les structures les plus répandues, nous analyserons leurs modes de fonctionnement, leurs avantages, leurs inconvénients, et nous comparerons leurs performances à celles de technologies traditionnelles pour les applications hyperfréquences. Cette approche intentionnellement simple et intuitive nous permettra de situer notre travail tout en traçant une prospective pour la suite. Nous verrons enfin quelles sont les caractéristiques actuelles ou potentielles des dispositifs électroniques (tels que les filtres accordables pour les soussystèmes de réception/transmission et les déphaseurs) qui intègrent des composants développés grâce à ces technologies. Dans le second chapitre, nous présenterons la conception, la fabrication, et la caractérisation de filtres accordables en bande passante, en fréquence centrale ou en toutes les deux simultanément, pour des applications en bande L et R. Ce travail vise à dépasser l’ approche traditionnelle de conception de filtres simplement accordables intégrant des composants micro-électromécaniques, et se propose de montrer la possibilité d’une conception “programmable” de filtres reconfigurables. En effet, l’approche traditionnelle consiste à synthétiser une réponse particulière en fréquence, tout en prévoyant une variation des éléments accordables pour en moduler la largeur de bande ou la fréquence centrale. Cela donne une possibilité d’accord qui s’avère en général assez limitée. La méthode proposée au contraire est idéale pour l’implémentation d’un grand nombre de fonctions de filtrage qui atteignent une flexibilité élevée d’accord pour toutes les caractéristiques du gabarit de filtrage. Les outils de synthèse utilisés, ainsi que l’implémentation physique du filtre reconfigurable à l’aide de banques de capacités commutées seront montrées dans ce chapitre. 5

Ce travail s’avère en outre intéressant car il représente un des premiers exemples de circuits hyperfréquences reconfigurables ayant intégré les composants MEMS sur un substrat d’alumine. L’intégration de la technologie MEMS aux substrats céramiques fait actuellement l’objet d’un effort de recherche très important vu la prédominance de ces substrats notamment dans les sous-systèmes de réception sans fil à faible niveau des pertes. Le succès de cette intégration pourrait accélérer l’évolution définitive des MEMS de la dimension prototype/universitaire vers une production en grande échelle et faible coût. La conception électromagnétique des filtres qui font l’objet de ce chapitre est donc consacrée au développement de résonateurs à fort facteur de qualité en technologie micro-ruban sur alumine. Cette étude à fait appel à une approche originale à éléments localisés afin d’obtenir des résonateurs très compacts et compatibles avec la réalisation de filtres miniaturisés. Ceci à demander un effort particulier pour la mise au point d’un procédé de fabrication spécifique dont nous résumerons les étapes principales. Les résultats de mesures seront enfin présentés et discutés montrant un excellent accord avec les performances hyperfréquences visées . Le troisième chapitre sera dédié au développement d’une autre topologie de circuit hyperfréquence reconfigurable. Ce travail s’inscrit dans le cadre du projet ReRaFe (Reconfigurable Radio Front-End) soutenu et financé par le réseau européen AMICOM (Advanced MEMS for RF and Millimeter-wave Communications) pour la réalisation d’un filtre reconfigurable MEMS destiné à un sous-système de réception radio. Ce filtre est basé sur une architecture à résonateurs couplés proche de celle utilisée dans le second chapitre mais il réalise une performance de reconfiguration différente et originale. Le but du filtre ReRaFe sera en effet de reconfigurer la réponse en fréquence sur deux standards préfixés (DCS 1800 et WLAN) complètement hétérogènes. Le défi technologique à relever en ce cas consiste à réaliser un circuit capable de commuter entre deux standards très hétérogènes avec une architecture de filtre unique et compacte. Il sera ainsi montré que cela est possible au prix d’un effort relatif de conception des transformateurs d’impédance en entré et sortie du filtre, qui permettent de parvenir à des éléments LC technologiquement réalisables pour les deux standards. Ce chapitre sera donc en grande partie consacré au développement de solutions spécifiques pour réaliser des transformateurs qui combinent une architecture distribuée avec des capacités localisés et accordables par des commutateurs MEMS. L’architecture obtenue ainsi que les premiers 6

résultats de mesure pour ce filtre reconfigurable bi-standard seront ensuite présentés et discutés. Dans un quatrième et dernier chapitre nous présenterons un nouveau concept de capacité commutée MEMS spécifiquement conçue pour des applications forte puissance. Les capacités variables MEMS présentent un très fort potentiel notamment pour l’application aux filtres reconfigurables et aux systèmes de routage et déphasage électronique. Néanmoins la plupart des exemples disponibles actuellement considère cette intégration en conditions de faible puissance du signal RF injecté. En effet la fiabilité des composants MEMS soumis à un niveau de puissance RF élevé n’a été prouvée qu’en mode “cold switching ”, ce qui signifie que la puissance injectée est coupée avant toute commutation de la capacité MEMS. Il apparaît pourtant primordial d’étudier la tenue en puissance des varactors MEMS lorsque la puissance RF est continuellement injectées dans le dispositif. Ceci revient alors à un étude de fiabilité en mode “hot switching”. Le travail que nous présenterons dans ce chapitre se propose d’une part de développer une varactor de puissance pour des applications hot switching, et de l’autre de caractériser la tenue en puissance du composant réalisé. En ce qui concerne la conception, elle a d’abord fait appel à une étude théorique des principaux mécanismes de défaillance, tels que l’auto-actionnement ou l’auto-maintien, d’une micro-poutre à contact capacitif en condition de forte puissance. Cette étape a essentiellement portée sur la modélisation de la pression électrostatique qui vient s’installer sur la poutre induite par la puissance RF traversant le dispositif. Il a été ainsi montré que la région du contact capacitif est extrêmement sensible à l’influence de la puissance RF et que l’amélioration de la tenue en puissance du dispositif relève du dimensionnement approprié de son impédance capacitive par rapport à sa raideur structurale. Une topologie originale à été mise en place qui se propose de dissocier la performance capacitive du varactor de son comportement mécanique et parvient à limiter efficacement les effets de la puissance. Ensuite une méthode de synthèse/optimisation a été mise au point afin de déterminer exactement la géométrie optimale du dispositif et d’en repousser les limites de tenue en puissance. Cette méthode est basée sur l’optimisation récursive multi-physique des paramètres géométriques de la structure combinant les aspects électrostatique, mécanique et électromagnétique. Les outils de calcul et la démarche suivie dans cette étape afin d’atteindre les performances souhaitées seront décrits dans ce quatrième chapitre. 7

Finalement, les choix de conception seront validés par des caractérisations mécaniques et électromagnétiques des dispositifs réalisés. Cela fait appel à de tests de tenue de puissance maximale ainsi que de tests de durée de vie des composants soumis à divers niveaux de puissance. La fiabilité à long terme est évaluée par de mesures de caractéristiques capacitétension (courbes C(V)) qui sont répétées au bout de plusieurs sessions de cyclage sous un niveau de puissance RF constant. Ce type de tests se propose donc de détecter les éventuelles dégradations de la performance des varactors en fonction de la puissance injectée et de la durée du cycle. L’étude de tenue de puissance maximale consiste à répéter des mesures des tensions d’actionnement et de relâche des dispositifs pour des niveaux de puissance croissants. Ce type de tests se propose ainsi de déterminer le niveaux de puissance à la quelle une défaillance survient par l’auto-actionnement ou l’auto-maintien des micropoutres. Les résultats des tests de fiabilité seront présentés et discutés. Les résultats des tests de fiabilité, qui permettent de valider à la fois la topologie et la méthodologie d’optimisation proposées, seront présentés et discutés. Enfin, nous effectuerons une synthèse des résultats obtenus au cours de ce travail de thèse et nous développerons les perspectives qui en découlent dans la conclusion générale.

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CHAPTER 1

RF MEMS technology for reconfigurable microwave applications

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RF MEMS technology for reconfigurable microwave applications

The manuscript introduced with this chapter involves predominantly analysis, design and fabrication of circuits using the technology of micro-electromechanical systems (MEMS) for Radio-Frequency circuits (RF MEMS). This technology has emerged in recent years with a comparable level of interest and even more rapid development than silicon-based RF integrated circuits (RFIC’s). RF MEMS enable a new class of components and subsystems that are electrically reconfigurable, provide new system capabilities and display superior high-frequency performance relative to conventional (usually semiconductor) devices. RF MEMS technology probably represents one of the most interesting and inter-disciplinary research/industry area in microwave engineering. Indeed on one side MEMS devices enable substantial enhancement of microwave designer possibilities by providing almost ideal tunable network transfer functions. On the other side MEMS fabrication represents a challenge for technology engineer that intrinsically conjugates different areas as mechanics, electrostatics, electromagnetics, material science… To a certain extent, the author was given the chance to follow the still ongoing evolution of RF MEMS from the status of latest revolution in microelectronics into one of the major topics in microwave engineering. MEMS technology is growing so fast and in so widespread domains that an exhaustive introduction even limited to RF area would provide sufficient material for a whole Ph.D. thesis. Hence this chapter is not intended as a systematic introduction to MEMS technology. On the other hand it appeared useful to collect some of the subjects, principles and metrics that are commonly employed all over the rest of the manuscript. The main goal is not to achieve completeness in enumerating MEMS components and properties but showing the complete versatility of MEMS technology while providing an insight into MEMS impact on modern RF communication systems.

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I MEMS components

Micro-electromechanical systems include a variety of circuits and devices. These components borrow some consolidated techniques from integrated circuits fabrication processes to realize mechanically deformable devices at micron scale [1] . The most important technique for RF MEMS is surface micromachining which consists of the deposition and lithographic patterning of various thin films, usually on silicon, quartz, or other substrates. Generally, the intent is to make one or more of the sacrificial films freestanding over a selected part of the substrate, later able to undergo the mechanical deformation motion or actuation. This is done by depositing a sacrificial film (or films) underneath the one(s) to be released, which is removed in the last steps of the process using selective etchants. Bulk micromachining involves the creation of mechanical structures directly in silicon, quartz, or other substrates by selectively removing the substrate material. MEMS devices currently collect considerable efforts in multiple research/industry domains including electronics, micromechanics, optics, fluid dynamics, bioengineering and defence. Originally the term MEMS was

strictly employed to designate mechanically

deformable devices, while nowadays it rather includes the totality of devices obtained by micro-fabrication techniques. The first devices that fully demonstrated the potential of this technology were microsensors that exploit the sensibility of special materials and thin film to pressure, acceleration or mechanical deformation. This lead to a variety of monolithically integrated systems where packaged MEMS sensors are integrated to traditional electronic components. A well known example is probably the Analog Devices AD-XL50 automotive airbag accelerometer [2-3]. This is essentially a monolithic surface micromachined accelerometer with capacitive position detection. Beyond crash sensing for airbag control MEMS devices offer interesting solutions for vehicle dynamic control, rollover detection, antitheft systems and electronic parking brake systems. Several examples of these navigation systems are based on MEMS angular-rate sensing gyros.

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Figure 1.1 Analog Devices AD-XL50 automotive airbag accelerometer.

Optics also exploits MEMS technology to realise adaptive micromachined mirrors based on thin reflecting membranes [4] whose position and inclination can be electronically controlled resulting in reconfigurable optic networks. Deformable mirrors were successfully integrated to fibre laser, to control and modulate pulse laser emission (active Q-switched fiber laser), [5].

(a)

(b)

Figure 1.2 Reconfiguralbe mirrors realized at TU Delft (a), and Xlim (b).

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MEMS technology is showing increasing potential for fluidic (inkjet nozzles) and biomedical applications. Indeed miniaturized size exhibited by micromachined devices is strongly attractive for the implementation of non invasive systems for diagnostic or, in more advanced applications, local treatment. In particular, there is a growing trend towards development of complex microsystems (lab on chip) integrating micro-probes, micro-pumps at cell-scale and capable to deal with minimum clusters of cells for on side detection of physiological parameters or micro-injection of reagents.

(a)

(b)

Figure 1.3 (a) Microfabricated silicon neural probe array [6]. (b) Cell-based biosensors with micro-electrode array [7].

Spectroscopy and astronomy take advantage of MEMS technology as well. Microshutter arrays (see Figure 1.4, [8]) are being developed at NASA Goddard Space Flight Center for use in the Near Infrared Spectrograph (NIRSpec) on the James Webb Space Telescope (JWST), which is expected to measure the number and density of young galaxies. The microshutter arrays are designed for the simultaneous observation of a large number of sources in the sky and the transmission of light to the NIRSpec detector with high contrast. Individual shutter cells can be actuated to a fully open position, allowing light to completely pass through the device and reach the NIRSpec detector. A dedicated actuation mechanism allows selecting and controlling single shutters in a matrix of more than 200 objects (Figure1.4 (b)). Shutters are opened by a magnetic field (that allows large displacement actuation), latched by an electrostatic force, and selected to be open or close by addressing the electrostatic force. This allows an arbitrary pattern to be generated as shown in Figure 1.4 (c). 14

(a)

(b)

(c)

Figure 1.4 Frontside (a) and backside (b) of the microshutter array. Arbitrary pattern generation (c).

IBM is also testing a “millipede” high density data storage system based on MEMS components (see Figure 1.5, [9]). This exploits tiny depressions created with an atomic force microscopy (AFM) tip in a polymer medium to write data bits that can be read back and erased by the same tip. This thermodynamical storage technique is capable of achieving data densities exceeding 1 Tb/in2. well beyond the expected limit of magnetic recording. High data rates can also be achieved by making use of massive parallelism.

Cantile ver

Figure 1.5 IBM MEMS millipede high density data storage system.

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Apparently the domains of application of MEMS technology are expanding very quickly. This is mainly due to compatibility with miniaturized size applications and potential for low-cost mass production. However to date there is little doubt that one of the application field that took huge advantage from advent of MEMS era is that of RF communication systems. This justifies the establishment of a branch named RF MEMS and to some extent can be attributed to the success encountered by MEMS switches which are still considered as the paradigm of RF MEMS devices [10]. The tremendous impact that MEMS technology had on microwave systems in the last ten years can be explained by thinking that probably since the dawn of GaAs IC’s no other RF technology had shown so much potential to improve system performance and affordability at the same time.

II RF MEMS

MEMS fabrication techniques developed and differentiated in the last 15 years and result today in a multitude of devices for microwave applications. A survey of RF MEMS current research/industry activities leads to the following main areas [11]. •

RF MEMS switches: based on controlled movement of mechanical small parts to perform almost ideal RF operations from DC-120 GHz.

•

Variable capacitances: as switches move several micrometers when actuated for operations from 0.1 up to100 GHz.

•

Micromachined inductors: relatively high-Q (Q=20-60) inductors (L=2-15 nH) whose parasitic capacitance is reduced by suspending the inductor high above the substrate [12]. They are static and do not move.

•

Micromachined transmission lines, resonators, filters and antennas: they are generally integrated on thin dielectric membranes or use bulk micromachining of silicon to reduce dielectric loss for application from 10 up to 200 GHz. [1316]. They are static.

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•

FBAR (Film Bulk Acoustic Resonators) devices: they use acoustic vibrations in thin piezoelectric films to achieve high-Q (Q>2000) up to 3 GHz with excellent loss and power handling behaviour. The acoustic wave is excited in the vertical plane of the piezoelectric film resulting in extremely compact resonators [17-18].

•

RF micromechanical resonators and filters: they use mechanical vibration of extremely small beams to achieve high-Q (Q>8000) resonance for 0.01-200 MHz operations. They can be used for reference clock circuits [19-20]

RF MEMS switches and variable capacitances are introduced in some more detail since they were crucial for the development of this thesis work.

II.1 RF MEMS switches

RF MEMS switches are miniature devices that use mechanical movement to achieve a short circuit or an open circuit on a RF transmission line [11]. Although they present a wide variety of structure and layout configurations most MEMS switches share large surface to volume ratio so that surface effects such electrostatics, wetting or air damping dominate volume effects such as inertial or gravitational forces. In order to produce mechanical movement and to achieve switching (“actuation”), several kind of forces can be applied as electrostatic, magnetostatic, piezoelectric or thermal. Electrostatic actuation (see Figure 1.6) is the most common technique in use to date and is discussed in some detail because it concerns all MEMS devices presented in this work. This mechanism is easy to implement since it just requires a fixed (“pull-down”) electrode, a movable electrode which is integrated or corresponds to the mechanically deformable part of the switch, and a bias line to apply voltages between the electrodes. Fixed and movable electrode constitute an approximately parallel-plate capacitor and when a voltage is applied between them an electrostatic force is induced on the movable electrode. This force provokes

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deflection of the movable electrode on the pull-down electrode and results in rapid (1-200 µs) actuation of the switch if the applied voltage exceeds a critical Vp “pull-down” voltage.

Non actuated switches Movable electrode V=0

Movable electrode g=g0

g=g0

V=0

Actuated switches RF INPUT

V=Vp

Pull-down eletrode

g=0

Contact eletrode

g=0

V=Vp

RF OUTPUT Pull-down+contact eletrode

Figure 1.6 Electrostatic actuation for different kinds of RF MEMS switches. Therefore electrostatic actuation occurs through the capacitance formed by pull-down and movable electrode and a very small current (~µA) flows into the bias line during the time necessary for this capacitance to charge/discharge. On the other hand, the current is zero in static conditions which means that power is consumed only during commutation from non actuated to actuated state (or vice versa). Hence electrostatic actuation provides very low power consume, small electrode size, relatively short switching time and contact force which is adequate for most applications. However electrostatic actuated switches require actuation voltages from 10-100 V which makes necessary to amplify the typical 3-5 V source voltages. Furthermore due to non linear dependence of electrostatic force with the actuation gap, controlled deflection of the movable electrode gets impossible as it approaches actuation (see section II.1.1). Thus the system collapses in the actuated state rather than gradually achieving it. Such an instable behaviour especially affects MEMS varactors performance because analog controlled variation of capacitance can be only obtained over a limited range. Magnetostatic actuation provides high contact force with no instability phenomena, but requires considerable size and currents to induce a permanent magnetic field. Thermal actuation results in high contact force but requires relatively high switching time and power consumption. Piezoelectric actuation has virtually no power consume and allows control on the release mechanism of the movable membrane, but requires complex and high temperature 18

fabrication process. Finally different actuation mechanisms can be coupled together (ex. thermal actuation with electrostatic hold) resulting in virtually zero power consumption when the switch is actuated. However in practice only electrostatic actuation was tested in a large frequency range and with high reliability. We now summarize some of the characteristics and configurations that allow dividing MEMS switches in different categories. This classification does not intend to be exhaustive and is not the only possible however it turns out to be practical for further developments. Then switches are categorized by the following three characteristics: 1. Mechanical structure 2. RF circuit configuration 3. Form of contact As for the mechanical structure the switches are either fabricated using a fixed-fixed membrane that results in a air bridge (Figure 1.7 (a)), or a floating cantilever (Figure 1.7 (b)).

(a)

(b)

Figure 1.7 Fixed-fixed beam (a )[21] , and floating cantilever (b)[22] configurations.

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As for the RF circuit configuration it should be pointed out that following electrical convention the number of poles is the number of input terminals or ports to the switch, while the number of throws is the number of output terminals or ports. Then the two common circuit configurations are single pole single throw (SPST) shunt (Figure 1.8 (a)) or series (Figure 1.8 (b))connected [10].

(a)

(b)

Figure 1.8 Shunt (a), and series (b) configurations [23].

The two common contact forms are the capacitive (metal-insulator-metal) contact, and the resistive (metal to metal) contact. The air bridge switch in Figure 1.7 (a) also represents an example of capacitive contact while the cantilever switch in Figure 1.7 (b) illustrates a typical metal to metal contact. However it should be noticed that there are several examples of bridge switches with resistive contact [24] as well as cantilever switches with capacitive contact (see Chapter 4). The resistive contact switches use metal-to-metal direct to achieve an Ohmic contact between two electrodes. In the actuated (On) state the switch presents an essentially resistive impedance ROn which allows signal transmission from DC up to very high frequency. The ROn value and evolution with time and usage strongly depend on the metal material used. The reliability of Ohmic switches is limited by contact degradation and hardening due to the impact force or deposit of contaminants and typically allows lifetime in the order of 1012 cycles (in low power conditions) [11,25]. For capacitive contact switches the contact electrode is covered by an insulating dielectric layer. In the On state the metal beam gets in contact with this layer and the switch presents an essentially capacitive COn impedance, which allows high frequency signal transmission across the capacitance through displacement current. Large COn capacitances are 20

required for proper transmission of RF signal. The ratio of capacitance in the actuated state COn to capacitance in the non actuated (Off) state is COn/COff and is a fundamental parameter for capacitive contact switches. Capacitive contact switches experience smaller contact degradation but suffer from “stiction” between the bridge and bottom contact due to dielectric charging [26-27], humidity or contaminants, and are intrinsically unfit for low frequency applications. Capacitive switches have been tested up to 1011 cycles in low-power conditions [28-30]. From an electrical point of view it is easy to realize how MEMS ohmic contact switches approach an ideal 2-state digital behaviour. On the other hand, although capacitive switches are often employed to perform two 2-state operations (see section II.2 and Chapter 4), they can also be used to provide continuous variation of the capacitance, to some extent. An ideal RF switch is a lossless device for which 2 only transmission states are possible: an Off state where the signal is blocked and the output port is disconnected from the input port, and an On state where the two ports are connected and the signal is transmitted with no loss from input to output. An RF MEMS ohmic contact switch is a real device that can be described with good approximation by the 2-state simple circuit model in Figure 1.9.

COff

RF Input

RF Output ROn

Figure 1.9 Two state circuit model for an RF MEMS ohmic contact switch.

In the Off state the MEMS switch is comparable to a small capacitance COFF and the component presents a high impedance condition so that output is practically isolated from input. In the On state the impedance of the switch reduces to a small resistance ROn that connects input and output and allows transmission of the signal with small loss.

21

A generally used figure of merit for such a component is the cut-off frequency fc that is defined as the frequency such that the ratio of the modulus of impedance in the On state to the modulus of impedance in the Off state is 1.

fc =

1 2π COff ROn

(1)

This limits the range of application of the switch since at f= fc it performs no more impedance commutation. The higher the cut-off frequency the better the switch performance as compared to the ideal behaviour. For instance a MEMS switch with typical values of COFF = 10 fF, RON = 1Ω and has an fc =16 THz, which is about 10 times as higher as typical p-i-n switches cutoff frequencies. Switching time is defined as the time necessary for going from Off to On state (and vice versa) and is also an important parameter when defining the switch performances and range of applications. For example typical switching times range between 1-300 µs and are higher than p-i-n diodes and FET transistors typical switching times (1-100 ns). Finally a few conventional RF metrics commonly apply to MEMS switches and are extensively used all along this work. These are all linked to the On/Off ratio and the cut-off frequency parameters introduced above. 1. The insertion loss, i.e. the 1/|S21| parameter in the On state 2. The isolation, i.e. the 1/|S21| parameter in the Off state 3. The return loss i.e. the 1/|S11| parameter in both states

II.1.1 Overview of electromechanical behaviour for electro-statically actuated switches.

Structural analysis of a cantilever beam submitted to electrostatic actuation will be carried out in some detail in Chapter 4; however the following intuitive description is freely summarized from [31] and introduces some general use parameters basing on the analogy 22

with a simple mechanical system. The switches are modelled as mechanical springs with an equivalent spring constant k [N/m] which depends on the geometrical dimensions of the movable electrode and on the Young’s modulus of the material used. k is 5-40 N/m for most switch designs. Also, the switches have very low mass m (≤10-10 Kg) which makes them pretty insensitive to acceleration forces. By assuming an electrostatic actuation force and referring to the structure parameters as reported by Figure 1.7 (a) the force between the top and the bottom electrodes is:

CV 2 F=  t 2 g + d εr 

  

=

ε 0 AV 2  t 2 g + d εr 

  

2

(2)

Where V, g, and C are the applied voltage, gap distance and capacitance between the lower (pull-down) and upper electrodes, respectively, and A is the area of the pull-down electrode. The latter is often covered by a dielectric layer with a thickness td and a relative dielectric constant εr to prevent a short circuit between the movable and fixed plates. For a switch with A=100*100 µm2, V= 50 V, g= 3 µm the initial pull-down force is 12 µN which is pretty low. However this force is sufficient to actuate the switch because as the top electrode starts to deflect the gap is reduced and the pull-down force on the beam increases. On the other hand, a pull-up restoring force also applies to the beam due to the spring constant of the switch, so that an equilibrium is achieved when pull-down and pull-up forces are the same:

F=

ε 0 AV 2  t 2 g + d εr 

  

2

= k(g − g0 )

(3)

where g0 is the beam initial height. This equation predicts a stable position of the beam up to ~ g0/3, after what the actuation force is no more balanced by the spring force and the switch collapses to the down state position. The pull-down voltage Vp that causes this collapse is:

Vp ≈

8kg 03 27ε 0 A

(4) 23

For k=10 N/m, and the values above Vp =30 V. The typically applied voltage is 1.21.4Vp so that actuation is achieved rapidly. Once the switch is pulled in the down state, in capacitive switches an intimate contact is achieved between the beam and the dielectric layer, while in resistive switches a 0.3-0.6 m air gap layer can be present between the beam and the pull-down electrode. In both cases the electrostatic voltage can be reduced to a “release” voltage, Vr =8-20 V, while still keeping the switch actuated. This allows reducing the electric field in the dielectric and the possibility of dielectric breakdown or charge injection into the dielectric. Such an hysteretic behaviour is typical of all MEMS switches and is represented in

3

Stability limit

2.5

1

Release

2

0.5

Vr

1.5

0 0

Actuation

Top electrode height (µm)

Figure 1.10.

Vp Hold-down 10

20

30

40

Applied Voltage (V) Figure 1.10 Beam height versus applied voltage.

It should be observed that while the initial force is 12 µN the force at contact increases to hundreds of µN up to 1.5 mN depending on switch configuration and design. This is essentially what allows metal-to-metal contact switches to achieve low contact resistances. Once the bias voltage is removed the pull-up force is approximately given by F=kg0 since the displacement of the electrode is g0. This results in a pull-up force of 30-60 µN for most switches and is quite small. For this reason MEMS switches are very sensitive to surface chemistry (humidity, contaminants, dielectric charging…) and must be packaged in cleanroom conditions [32-33].

24

MEMS switches also follow classical laws of motion and the d’Alembert-Lagrange principle [34]:

m

d 2g dg +b + k ( g − g 0 ) = Fe 2 dt dt

(5)

where m and b are the mass and damping factor of the top electrode and Fe is the electrical force from (2). This is a second order system with a resonant frequency:

ω0 =

k ω 0 = km / b m

(6)

A switch with m= 10-11 Kg and k = 10 N/m results in a resonant frequency of 50 KHz. The damping factor is seen to be inversely proportional to the mechanical quality factor (Q) defined as Q =

km /b which is 0.2-5 for most switch designs. Low Q systems result in slow

switches while high Q systems result in long settling times for both the actuation and the release operations. Switches that operate in vacuum can have a Q as high as 10-1000, while for atmospheric pressure operations it can be shown that a proper condition is reached for Q~1 which assures an appropriate damping factor. An approximated equation that accurately predicts the switching time can be derived [35] by setting b=0 (no damping) and taking the electrical force as the initial force with g=g0.

t ≈ 3.67

Vp Vω 0

(7)

where V is the applied voltage and the accuracy is within 10% for Q>1 and V>1.3Vp. For a switch with ω0=40 KHz and V=1.4Vp the switching time is 10 µs. (7) shows that it is hard to get very low switching times. For instance a switching time lower than 1 µs requires a resonant frequency ω0=190 KHz and a ratio Vp/V ≤3. Such a high resonant frequency could only be achieved using a high spring constant (and a low mass), so that the associated pulldown voltage would be high and the applied voltage V would be really high! [36]. 25

II.2 MEMS variable capacitors

MEMS tunable capacitances are one of the most promising area of RF MEMS technology although they have not progressed at the speed of MEMS switches. This is probably because silicon and GaAs varactors result in excellent performance especially below 5 GHz and do not have special packaging requirements. However MEMS varactors have the potential of very high-Q operations (Q=100-400), they withstand large RF voltage swings and result in high third order intercept point (IP3) tunable networks, they consume low current even in high power application and can be inexpensively fabricated on glass, ceramics or high-resistivity silicon substrates. The capacitive micro-switches based on a parallel plate architecture are intrinsically suitable for realizing MEMS tunable capacitances. Nevertheless it has been observed how difficult is to achieve analog capacitance variation over a large range for this kind of device. Indeed the previous section described the phenomenon of mechanical instability of parallel plate capacitors under an electrostatic force. It was found that the top plate can be moved to a gap height of 2 g0/3 before it collapses on the bottom plate. The maximum capacitance ratio that can be achieved over the stable range of capacitive variation is then:

ε0 A

+Cf C Max C (2 g 0 / 3) (2 g 0 / 3) = = Cr = ε0 A COff C(g 0 ) +Cf g0

(8)

where Cf is the parasitic fringing field capacitance associated with the beam and typically results in 20-40% increase of capacitance in both cases. This leads to degradation of the capacitance ratio which ranges between 1.4 and 1.3 for most designs. Indeed higher capacitance ratios are typically obtained by a pseudo-digital operation which consist of exploiting the capacitance value in the down state. The latter is typically much higher than the CMax value in (8) due to dielectric layer that covers the contact electrode and has relative dielectric constant 4-10 which is higher than the air’s one. This kind of functioning is more

26

compatible with the instable behaviour of electrostatically actuated MEMS capacitances and is widely employed for the varactor presented in Chapter 4. Alternatively MEMS tunable capacitors can be designed with more sophisticated configurations or different kind of switches. Therefore it is essentially possible to distinguish three different kinds of MEMS varactor. The first one is based on the parallel-plate (vertical) approach where the variable capacitance is achieved by changing the gap between the capacitors plates (Figure 1.11). This suitable for capacitances of few pF applications from 150 GHz.

Electroplated membrane

Springs Anchors

m ov

em en

t

Figure 1.11 Two examples of parallel plate analog MEMS varactors [37-38].

Figure 1.12 An example of interdigital analog MEMS varactor [39]. 27

The second approach is based on an interdigital design and a variable capacitance is obtained by exploiting mechanical horizontal movement to change the common interfaced area of two comb-shaped electrodes (Figure 1.12). This allows obtaining large capacitance values (4-10 pF) and ratio (5-10), and is suitable for applications from 0.1 up to 6 GHz . Nevertheless the size and the design complexity are often considerable. The third approach consists of building a fixed capacitance bank and using MEMS switches to select the required total capacitance (Figure 1.13). This approach allows achieving very high capacitance value (up to 35 pF) and large capacitance ratios, but the Q factor can be limited by the series switch contact resistance or by the metal-insulator-metal (MIM) capacitors used.

RF CPW input MEMS Ohmic switch Bias line

Bit2

Variable C bank

MEMS switch1

BIT1

MEMS switch2

BIT2

MEMS switch2

BIT3

Bit1

Bit3

MIM capacitor

Figure 1.13 An example of MEMS switched capacitor [40].

II.3 RF MEMS from a system perspective

Sections II.1-2 have illustrated several intrinsic properties of RF MEMS technology from a device point of view. This section resumes the introduced characteristics in order to evaluate the impact of RF MEMS on current or imminent RF communication systems. For this purpose two specific examples are considered which still are relevant for a variety of RF systems and subsystems. This necessarily implies a comparison with traditional semiconductor technologies that currently dominate the RF applications. 28

Miniaturized size: RF MEMS devices allow the execution of complex functions on a size scale orders-of-magnitude lower and at far less power than discrete circuits. At microwave frequency most MEMS components exhibit very small size to wavelength ratio and make possible short-circuit/opencircuit/tunable-capacitor operations that approach the lumped elements ideal behaviour. The potential for evolution into nano-scale is currently being showed and is compatible with the main trend in electronics. Virtually zero power consumption: most of the RF MEMS devices developed to date achieves electromechanical actuation electrostatically through air (or vacuum). Hence, the power consumption comes from dynamic current flowing to the MEMS only when actuation is occurring (10-100 nJ per switching cycle). Electromechanical isolation: the RF circuit of MEMS component does not leak or couple significantly to the actuation circuit. Excellent RF performances: currently fabricated micro-switches present low contact resistance in the On state, which results in very low insertion loss (2nH) in order to obtain compact LC resonators with feasible capacitors (C>2cm at 1.84 GHz on a 525µm thick quartz substrate and the resulting circuit, which combines two transformers with the filter core, would be unfit for the 77

integration to the ReRaFe front-end demonstrator. This shows the necessity for more compact solutions. Alternatively a hybrid lumped/distributed network could perform a quarter wave transformer, on narrowband basis, with a shorter length. Indeed by capacitively loading a transmission line it is possible to synthesize a sufficiently compact equivalent transformer, provided the line impedance and the capacitive load are properly chosen. These can be determined analytically, as summarized below, by computing and comparing the ABCD matrix for both the normal quarter wave transformer and the hybrid network to synthesize. Then, at 1.84 GHz, a Z1 characteristic impedance transmission line performs a quarter wave transformer with n=5 if the electrical length θ1=π/2 and Z1=Z0/n (Fig. 3.13). L1 and W1 are the line physical length and width.

θ1 Zo

Z1

ZL =

Z0 n2

L1

w1

Z1

Z1 = Z 0 Z L =

Z0 n

Figure 3.13 Transmission line quarter wave transformer: n=5 .

The associated ABCD [T1] matrix is known from the transmission line theory and is reported below:

 0 [T1 ] =  1 j  Z1

jZ1   0  

(1)

78

As about the equivalent hybrid transformer, the Z2 impedance transmission line capacitively loaded by a normalized b susceptance, is visible on Fig. 3.12. L2 and W2 are the line physical length and width, while θ2 is the electrical length at 1.84 GHz.

θ2 Zo

b =ωCZ2

Z2

ZL =

Z0 n2

w2

L2

Z2

b

Figure 3.14 Lumped/distributed equivalent transformer .

The related [T2] transmission matrix can be calculated, by multiplying the ABCD matrices of the 3 two-ports networks featuring in the cascade connection (Figure 3.14). These include two external θ2/2 lines, and a shunt C capacitance in the middle. All matrices are known from transmission line theory and [T2] is a function of the unknown Z2, θ2 , b:

θ2   cos 2 [T2 ] =  1 θ j sin 2 2  Z 2   =  j 1  Z 2

θ2  

1 2 ⋅ θ   cos 2   jωC 2  

jZ 2 sin

b cosθ 2 − sin θ 2 2 b b  sin θ 2 + 2 sin θ 2 + 2   

θ   0   cos 2 2 ⋅ 1 θ sin 2 1  j 2   Z 2

b b   jZ 2 sin θ 2 + sin θ 2 −   2 2   b  cosθ 2 − sin θ 2  2

jZ 2 sin

θ2 

2= θ  cos 2  2 

(2)

79

For this network to behave like a quarter wave transformer around 1.84 GHz, [T2] must be equated to [T1] , which leads to the three non linear equation system below, for the unknown θ2, b and Z2 :

b cosθ 2 − sin θ 2 = 0 2 b b  Z 2 sin θ 2 + cosθ 2 −  = Z1 2 2 

(3)

b b 1  sin θ 2 + 2 cosθ 2 + 2  = Z   1

1 Z2

This can be rearranged in a two equation system upon linearly combining the second and the third equation:

b = 2 cot g (θ 2 ) b=

(4)

Z 2 Z1 − Z1 Z 2

From the latter

Z 22 1 2 b ± b2 + 4 − Z1 ⇒ Z 2 − bZ 2 − Z1 = 0 ⇒ Z 2 = Z1 2 Z1 Z1

Z 2b =

100

30 20

Z2

50

b

10 0

0

b

-10 -20

Line Impedance: Z2 (Ω)

Succeptance: b (Ω-1 )

40

(5)

-50

-30 -40

00

0.5

1 π/2

1.5

π2

2.5

3 3π/2

Elec. Length: θ2 (rad)

3.5

2π4

-100 -10

-5

0

5

10

Susceptance: b (Ω-1 )

Figure 3.15 Graphical solution for the equation system (3) . 80

Then from (3), Z2 can be directly expressed as a function of Z1 and θ2 (Figure 3.15):

[

]

 Z1 cos 2 (θ 2 ) + sin 2 (θ 2 )  2 Z2 = cot(θ 2 ) ± 4 cot (θ 2 ) + 1 = Z1 cot(θ 2 ) ± = 2 sin 2 (θ 2 )   =

Z1 [cos(θ 2 ) ± 1] sin(θ 2 )

⇒ Z2 =

(6)

Z1 [1 ± cos(θ 2 )] sin(θ 2 )

Actually from a practical point of view θ2 cannot be considered as completely arbitrary and is more conveniently fixed by technological considerations. Since the upper standard central frequency is about 3 times as bigger as the lower standard one, there is significant advantage in choosing

1 π π = 3 2 6

θ2 = ⋅

(7)

which leads to L2 ~7500µm. This allows synthesizing a very compact hybrid network which is equivalent to the quarter-wave transformer in the lower band with Z1Low=Z0/nLow = 10Ω.

b π  = 8.02 pF bLow = 2 cot g   = 3.464 Ω −1 ⇒ C Low = ωZ 2 6 Z1Low =

Z0 = 10 Ω n Low

Z 2low =

Z1 π  sin  6

  π  1 + cos 6  = 37.32 Ω   

(8)

81

80

Z2 (θ2 ) b (θ2 )

60

Z2 =37.32 Ω

40

Z2 (Ω)

20 0

-20 -40

θ2 =

-60 -80

0

0.5 π/2

π 6 π1

1.5 3π/2

2

Elec. Length: θ2 (rad) Figure 3.16 Lumped/distributed equivalent transformer .

Now the microstrip structure visible in Figure 3.17 can be optimised such that it performs a hybrid transformer with nLow=5, Z2Low=37.3 Ω , bLow=3.5 Ω-1 and CLow=8 pF according to (8). This provides the required impedance matching for the DCS1800 standard. The equivalence to a quarter-wave transformer in the lower band can be appreciated from Figure 3.18.

W2 Low

L2

Z2 Low

bLow

Figure 3.17 Hybrid transformer for the lower (DCS1800) standard.

82

0

)) -20 1, 1( S( B d -40

S11 (dB)

)) 3, 3( S( B d

S11 λ/4 transf. S11 hybrid transf.

-60 1.0

1.2

1.4

1.6

1.8

2.0

2.2

2.4

2.6

Frequency (GHz) freq, GHz

Figure 3.18 Comparison between quarter wave transformer matching, and hybrid transformer matching .

On the other hand the same network turns out to be 3θ2=π/2 long in the upper band. Then a classical quarter wave transformer could be obtained for the WLAN standard if the CLow capacitive load could be disconnected, and the line impedance switched to Z2up=Z0/nup=27 Ω. For example, the hybrid network in Figure 3.19 presents a very small capacitive load connected to the line and can be optimised to behave as a simple Z2up=27 Ω transmission line with good approximation in the upper band. Therefore it performs a transformer with nup=1.85, Z2up= Z1up =27 Ω . This provides the required impedance matching for the upper WLAN standard.

W2 Low

W2 up

L2

Z2 Low

bLow

bup

Z2 up Figure 3.19 Hybrid transformer for the upper (WLAN) standard.

83

II.2.1 Double stadard MEMS transformers Now MEMS Ohmic switches can be easily employed to implement a control system that allows commuting from the transformer configuration in Figure 3.17 to the one in Figure 3.19, while keeping a single transformer architecture. The resulting double standard transformer can be seen in Figure 3.20 .

L2

Figure 3.20 MEMS double standard transformer .

For the MEMS switches in the ON-state (Figure 3.21), the capacitive load is connected to the transmission line and the system behaves as the lower standard equivalent transformer in Figure 3.17 with WLow =1750µm (Z2Low=37.32 Ω).

CON=8 pF

WON

ON state ZON

Figure 3.21 MEMS switches in the ON state: DCS standard transformer .

84

For the MEMS switches in the OFF-state (load disconnected), the line effective width is Wup =2715 µm (Z2up=27 Ω), the equivalent capacitive load is very low and the system performs a quarter wave transformer in the upper band (Figure 3.22).

CUP