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Silicon-Filled Rectangular Waveguides and Frequency Scanning Antennas for mm-Wave Integrated Systems Gennaro Gentile, Student Member, IEEE, Vladimir Jovanović, Member, IEEE, Marco J. Pelk, Student Member, IEEE, Lai Jiang, Ronald Dekker, P. de Graaf, Behzad Rejaei, Leo C. N. de Vreede, Senior Member, IEEE, Lis K. Nanver, Member, IEEE, and Marco Spirito, Member, IEEE
Abstract—We present a technology for the manufacturing of silicon-filled integrated waveguides enabling the realization of lowloss high-performance millimeter-wave passive components and high gain array antennas, thus facilitating the realization of highly integrated millimeter-wave systems. The proposed technology employs deep reactive-ion-etching (DRIE) techniques with aluminum metallization steps to integrate rectangular waveguides with high geometrical accuracy and continuous metallic side walls. Measurement results of integrated rectangular waveguides are reported exhibiting losses of 0.15 dB/ at 105 GHz. Moreover, ultra-wideband coplanar to waveguide transitions with 0.6 dB insertion loss at 105 GHz and return loss better than 15 dB from 80 to 110 GHz are described and characterized. The design, integration and measured performance of a frequency scanning slotted-waveguide array antenna is reported, achieving a measured beam steering capability of 82 within a band of 23 GHz and a half-power beam-width (HPBW) of 8.5 at 96 GHz. Finally, to showcase the capability of this technology to facilitate low-cost mm-wave system level integration, a frequency modulated continuous wave (FMCW) transmitreceive IC for imaging radar applications is flip-chip mounted directly on the integrated array and experimentally characterized. Index Terms—Flip-chip, frequency scanning array, integration, mm-wave interconnect, mm-wave system, radar, substrate integrated waveguide (SIW), W-band, waveguide.
I. INTRODUCTION
T
HE development of integrated millimeter-wave (mm-wave) systems has recently triggered the interest of a wide research community. The reasons for this are to be found in the large worldwide license-free bandwidths
Manuscript received March 01, 2013; revised August 01, 2013; accepted September 06, 2013. Date of publication September 11, 2013; date of current version November 25, 2013. This work was supported by the Memphis Smart Mix project. G. Gentile, V. Jovanović, M. J. Pelk, L. Jiang, L. C. N. de Vreede, L. K. Nanver, and M. Spirito are with the Delft Institute of Microsystems and Nanoelectronics (DIMES), Delft University of Technology, 2628 CT, Delft, The Netherlands (e-mail:
[email protected];
[email protected];
[email protected];
[email protected];
[email protected];
[email protected];
[email protected]). R. Dekker is with the Philips Research Laboratories, 5656 AA Eindhoven, The Netherlands, and also with the DIMES Institute, Delft University of Technology, Delft 2628 CT, The Netherlands (e-mail:
[email protected]). P. de Graaf is with the Philips Research Laboratories, 5656 AA Eindhoven, The Netherlands (e-mail:
[email protected]). B. Rejaei is with the Electric Engineering Department, Sharif University of Technology, Azadi Street, Tehran, Iran (e-mail:
[email protected]). Color versions of one or more of the figures in this paper are available online at http://ieeexplore.ieee.org. Digital Object Identifier 10.1109/TAP.2013.2281518
that are available in the mm-wave region allowing to target high performance systems for industrial, scientific and medical (ISM) applications, including broadband communication at 60 GHz and 122 GHz, imaging radar at 96 GHz and 140 GHz, and radio location at 120 GHz [1]. Nevertheless, mm-wave smart systems (i.e., employing multiple antennas and with beam steering capabilities) are hampered by the losses of the transmission lines required to distribute the signal to the multiple antenna elements and by the cross-talk between neighboring lines and radiating elements [2], [3]. Traditionally, bulk rectangular waveguides (RWGs) have been used in high-performance mm-wave systems to reduce losses, but are not suitable for integrated systems due to their large dimensions, and have poor manufacturing tolerances. To overcome these limitations, laminate-based waveguides structures, also referred to as substrate integrated waveguides (SIWs), have been introduced [4]. These structures use fenced via side walls, where the diameter and spacing define of the vias with respect to the waveguide width the losses and maximum operating frequency (i.e., nonleakage regime) of these structures [5], [6]. Most of the SIWs reported are based on multilayer printed circuit boards (PCBs) [7], [8] or low-temperature co-fired ceramics (LTCCs) [9], and have been used to design linear [10] or planar [11]–[14] leaky-wave arrays up to 140 GHz [15], [16]. These technologies present limitations for frequency up-scaling, that hinder the use of SIWs for deep mm-wave systems, due to: — Process tolerances: as example, the minimum metal opening on a PCB SIW which is in the order of 75 m would limit the directivity of a Taylor array [10] at mm-wave frequencies since the power is mostly radiated by the first slots of the array. — Fenced side wall implementation: optimal field confineand ment was proven to be achieved when [6]; these ratios were just above optimal values for SIWs on PCB at 94 GHz [10] and on LTCC at 140 GHz [15], [16]. The possibility of realizing a waveguide with continuous metallic walls operating in the mm-wave region was shown by [17], [18] by means of a SU-8 micro-machining technique, by [19] with a stacked silicon substrate, and by [20], [21] with a photo-imageable film as substrate. However, the reported waveguide electrical performances are limited by the losses
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due to low alignment accuracy (use of shim spacers, screws or pins to assemble the waveguide blocks) or lossy substrates. Recently the authors presented a manufacturing process to realize silicon integrated waveguides [22], [23] employing high resistivity wafers [24] and potassium hydroxide (KOH) etching [25], resulting in waveguides with a trapezoidal cross-section and photo-lithographic accuracy in the patterning of metal. Although low insertion losses in the mm-wave range were achieved, the poor geometrical accuracy resulting from the use of a wet etching process would complicate the interfacing with monolithic microwave integrated circuits (MMIC). In this work, we present a new integration platform for the realization of integrated rectangular waveguides and slotted antennas, with continuous vertical side walls, based on through wafer deep reactive-ion-etching. The manufacturing technology and its related process flow is presented in Section II. The designs of a coplanar to waveguide transition (i.e., to interface with planar components) and of a slotted waveguide frequency scanning antenna are given in Section III. Experimental results of a Dolph-Tschebyscheff array are presented in Section IV in the form of conducted (i.e., S-parameters) and radiated (i.e., employing a near-field antenna setup) measurements. Finally, a complete mm-wave system is presented in Section V, where the proposed technology platform is used to integrate, by means of a flip-chip assembly, a Bi-CMOS FMCW transmit-receive chip with the developed travelling wave antenna.
Fig. 1. Technology steps: deposition and patterning of the (thin) top metal, and deposition of the back metal (a), gluing of the wafer to a mechanical carrier (e.g., silicon dummy wafer) and realization of the deep trenches (b), second (thick) metal layer deposition and patterning (c).
II. TECHNOLOGY
Fig. 2. Photograph of the silicon-filled integrated waveguide. The waveguide is at the center, surrounded by the etched trench. On the top face, metal opening (in black) forming the transition (see Section III) can be observed.
A. Process Considerations The proposed manufacturing process is based on deep reactive ion etching, which facilitates the implementation of silicon-filled waveguides with rectangular cross section. Nevertheless, the metallization of perfectly vertical waveguide side walls poses a technological challenge when high aspect ratio trenches are considered. The process presented here targets a trench aspect ratio of 1.5, which ensures a metal coverage on the bottom of the waveguide side walls of at least two skin depths for mm-wave operation (e.g., 0.5 m at 100 GHz for sputtered aluminum). The process requires a limited number of masks, making it economically attractive. The metallization is performed in two steps: a (thin) first layer, deposited before the DRIE, is used for high resolution planar structures (transitions and slot antennas), while a (thick) second layer, deposited after the DRIE, ensures metal coverage on the side and bottom of the waveguide trench. The developed process also includes the possibility to integrate resistors and capacitors on the RWG using the first metal deposition and highly doped implanted regions. The integration of such components is crucial when targeting the realization of travelling wave antennas (resistors) and to minimize the area of on-chip decoupling capacitors. B. Process Flow Fig. 1 sketches the proposed process flow. Starting from an substrate ( cm), the first metal deposition (1 m of Al) is sputtered on the top side of the wafer and patterned using mask 1. The backside of the wafer is then fully metalized
Fig. 3. Schematic cross section of the integrated diffused resistor (a) and capacitor (b).
[Fig. 1(a)] and glued to a silicon wafer for mechanical support [Fig. 1(b)] [24]. The DRIE is then performed on the trench areas: the plasma beam etches through the whole silicon substrate and lands on the oxide layer. A second layer of thicker metal (4 m of Al) is sputtered on the top face including the waveguide walls [Fig. 1(c)] making contact to the first metal layer. A metal coverage on the side walls of 1.9 m near the top face and 0.5 m near the bottom face is obtained; the measured tilt angle of the trench side walls is 0.36 . Fig. 2 shows the final metalized waveguide including the coplanar to waveguide transition, which is described in Section III. The layer stacks of a diffused resistor and a metal-insulator-metal (MIM) capacitor are reported in Fig. 3(a) and (b), respectively. A resistance of 6.8 and a capacitance of 0.86 fF/ m were experimentally measured. III. INTEGRATED WAVEGUIDE COMPONENTS A. Folded Slot Transition Conventional rectangular waveguides employ a height-towidth ratio of 1:2 to realize full-band single-mode (i.e., )
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Fig. 5. HFSS simulation: relation between efficiency, offset (normalized to the waveguide half-width) and length of a slot resonating at 94 GHz. TABLE I DESIGN PARAMETERS OF THE DOLPH-TSCHEBYSCHEFF ARRAY
Fig. 4. Proposed transition: picture (a), dimensions (b), and HFSS simulated field distribution (c). In (a), on the top metallic face of the waveguide (in light grey), the shape of the integrated slot transition (dark grey) can be seen. Three planes of interest are depicted in (c).
structures for any specific frequency of operation. The waveguides implemented in this work have a height of 280 m, a width of 560 m, and a cut-off frequency of 77 GHz. In order to interface the proposed technology with a CPW environment (e.g., during testing phase or when connecting with active devices) we designed a low-loss coplanar to waveguide transition making use of a folded slot antenna coupling, as shown in Fig. 4. The transition operation can be described by dividing the structure into three sections. The transition starts with a coplanar waveguide (CPW) at plane 1: the signal line width and signal to ground spacing (i.e., 70 m and 12 m, respectively) are designed to provide a 50 impedance in order to interface with conventional coplanar systems (e.g., wafer probes, flip-chip or bond-wires assemblies). At plane 2, the CPW feeds a transverse coplanar slot antenna which couples energy inside the waveguide. In [26] a similar transition was reported, but the relative matching bandwidth, smaller than 2%, was explained by the reduced length of the transverse slot. To solve this, [27] increased the waveguide width around the transition, hence introducing a step discontinuity and placing extra constraints to avoid the generation of disturbing parasitic modes. In the transition layout we propose, two parasitic longitudinal slots (i.e., folded) are added at plane 2 in order to provide the proper loading terminations for the transverse slots and achieve a broadband impedance matching, without requiring discontinuities in the waveguide width. At plane 3 the field has become vertically uniform in the waveguide cross-section, representing a proper mode. The CPW lateral ground planes extend for 148 m from the CPW gaps to the longitudinal slots; therefore, the transition can be measured using wafer probes with a pitch of 100, 125 or 150 m.
B. Dolph-Tschebyscheff Array To demonstrate the capabilities of the proposed technology to realize high performance antennas, a travelling wave frequency scanning array [28], [29] was implemented using a slotted waveguide. The slots were designed to realize a Dolph-Tschebyscheff distribution of the radiated power [30] in order to reduce the levels of the secondary lobes in the radiation pattern. A constant slot width of 10 m was used to achieve a fine radiated power resolution: wider slots would have limited the minimum radiated power, while narrower slots would have been affected by manufacturing tolerances. Since the exact slot resonance frequency varies with its length and offset, both these parameters are optimized for each slot according to the radiated power and the center frequency desired. The relations between the radiated power, length and offset of the slots at the design center frequency are computed using a 3-D finite element model (FEM) solver, i.e., Ansoft HFSS. The results of this analysis is in Fig. 5. In order to obtain an accurate Dolph-Tschebyscheff distribution of the radiated powers and to properly compensate for the perturbation introduced by the resonant longitudinal slots to the superficial current flowing, a numerical optimizer within the Agilent Advanced Design System (ADS) environment was employed. The design parameters of the implemented DolphTschebyscheff array are summarized in Table I; as it can be seen by the dimensions reported in the table, the capability of realizing a Dolph-Tschebyscheff array strongly relies on the accuracy of the manufacturing process.
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Fig. 6. S-parameters of a waveguide segment (635 m) with transitions. A vertical line is drawn in correspondence of the maximum measured frequency (i.e., 110 GHz).
Fig. 7. Waveguide loss and dispersion diagram: simulations versus measurechoosing mm. ments. Losses are expressed as The simulation waveguide cut-off was shifted to account for the slightly reduced waveguide width due to manufacturing (i.e., from 77 to 79.5 GHz).
IV. MEASUREMENT RESULTS A. Transition The integrated waveguides are measured in the W-band with an Agilent PNA-X platform using mm-wave module extenders (e.g., WR 10) from OML Inc. The waveguides are contacted using Cascade Microtech infinity probes. Fig. 6 shows the measured S-parameters of a 635 m long waveguide section, including the input and output CPW to waveguide transitions. The measured is below dB in the whole 80–110 GHz frequency range. Although measurements could not be carried out above 110 GHz due to instrument limitations, HFSS simulations predict a dB matching bandwidth from 80 to 129 GHz (relative bandwidth 48%); above this value, impedance mismatching occurs due to the large amount of reactive energy stored inside the waveguide by the mode approaching its cut-off frequency, predicted by simulations, at 159 GHz [31]. Using an de-embedding algorithm as described in [23], an insertion loss for one single transition of 0.6 dB at 105 GHz was extracted. These performances in the W-band well correlate with the existing literature at lower frequency: [32] reported a coplanar to waveguide transition working in Ka-band with a matching bandwidth of 43% and an insertion loss of 0.4 dB, and [27] a U-band design with 45% relative bandwidth and 0.5 dB insertion loss. B. Waveguide The extracted propagation constant of the waveguide is reported in Fig. 7 and compared with HFSS simulations where the silicon substrate bulk resistivity is set to 1250 cm. Both the attenuation and the phase constant show an excellent agreement with simulation results. At 105 GHz, losses as low as 0.12 dB/mm are measured, corresponding to 0.15 dB/ , when normalized to the guided wavelength. A shift of 2.5 GHz was observed between the designed (77 GHz) and measured (79.5 GHz) cut-off frequency, which could be explained by manufacturing tolerances (in the order of few microns) that caused a reduction of the waveguide width, and was corrected for at simulation level. Table II shows a comparison of losses with conventional transmission lines (microstrip and CPW)
Fig. 8. Delft University of Technology near-field antenna system showing: feeding waveguide wafer probe and open ended waveguide probe (WR-10). The long metal lines visible on the wafer are the Dolph-Tschebyscheff antenna arrays presented in this work.
and with other waveguide structures reported in literature: the proposed process for the integration of silicon-filled rectangular waveguides achieves comparable losses per wavelength as the best results reported so far for an air-filled waveguide [19]. Given the high dielectric constant of the waveguide substrate material (silicon, ), the guided wavelength at 100 GHz is 2.1 times smaller than in air, which is beneficial in the design of compact traveling wave antennas. C. Array The radiation performance of the travelling wave array described in Section IV was evaluated making use of a custom near-field antenna system developed at the Delft University of Technology [34] (see Fig. 8). Waveguide probing (i.e., infinity waveguide WR-10) was used to feed the antenna, while an open ended WR-10 waveguide probe was used to sense the field above the structure. The waveguide end of the travelling
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TABLE II COMPARISON OF LOSSES IN TRANSMISSION LINES
Fig. 9. Simulated and measured radiation pattern for the Dolph-Tschebyscheff GHz). array in Table II at broadside (
wave antenna was terminated by an integrated 50 load (as described in Section II) in the waveguide to CPW transition. The near-field scan was performed over a 40 mm 20 mm plane at a distance of 5 mm from the antenna. Fig. 9 reports the normalized radiation pattern of the integrated Dolph-Tschebyscheff array in the longitudinal plane at broadside frequency (i.e., 96 GHz). The reconstructed main beam in the far field pattern closely resembles the simulation results; the half-power beam-width is 8.5 . The measured side lobe level (SLL) is below dB in the whole plane. Note that the proximity of the wafer (feeding) probe limits the range of validity [35] of the near-field measurements, reducing the accuracy of the measured side lobe levels close to the end-fire directions. In Fig. 10 the measured scanning angle of the array (i.e., the direction of broadside radiation) versus frequency is reported, showing a good agreement between measurement results, theory [28] and simulations. Considering FMCW applications where a side lobe level of at least 10 dB is required [36], the scanning range of the implemented array goes from at 87 GHz to at 110 GHz, as shown in Fig. 10. Although the directivity of the array could not be directly measured due to limitation in the characterization test-bench, closed form equations for a Dolph-Tschebyscheff power distribution can be used to relate the measure half-power beam-width to the directivity [30], [37]: the extracted value is 12.4, which is in close agreement to the 13.4 dB obtained in simulation. A summary of the overall performances of the array is reported in Table III.
Fig. 10. Theoretical, simulated and measured scanning angle as a function of frequency. TABLE III ARRAY PERFORMANCES
V. SYSTEM INTEGRATION A. Integration Technology Platform The waveguide manufacturing process proposed in Section II can be used as technology platform for the integration of mm-wave systems. In order to interface in a broadband fashion the waveguide structures with high-speed ICs, flip-chip interconnects are employed. Flip-chip transitions are often preferred to bond-wire or ribbon-cord bonding techniques since they present lower insertion losses and larger bandwidths at millimeter-waves. To motivate this consider that the of a series inductor in a 50 system can be expressed by [38] (1)
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At 94 GHz a flip-chip interconnection with 60 m high solder bumps (self-inductance of 56 pH [39]) produces an insertion loss of 0.45 dB, while an interconnection based on a 0.5–1.0 mm long bond-wire (wire diameter 25 m, self-inductance 0.4–0.8 nH [40]) causes 8.2–13.7 dB of loss. Compensating capacitors can be added at the output but their values is in the order of few femtofarads at mm-wave, which is too small for practical implementations. Moreover, the compensating technique is only effective in a narrow bandwidth around the resonant frequency of the so-formed LC resonator: as example, using a MIM capacitor and a 0.4 nH bond-wire with quality factors of 20 and 50, respectively, at 94 GHz [41], a 3-dB bandwidth of 14 GHz (relative bandwidth 15%) is obtained. Therefore, bond-wire interconnections at 94 GHz would limit the bandwidth of the developed broadband waveguide components, and a flip-chip assembly is preferred. B. Mm-Wave Radar Transmitter In this subsection we present a highly integrated FMCW front-end system using the proposed waveguide technology. The active chip, manufactured in 130 nm Bi-CMOS technology, provides an eight-time multiplication to up-convert an FMCW signal from 11.25–12.5 GHz to 90–100 GHz. Flip-chip assembly allows to interface this IC directly to the CPW to waveguide transition described in Section III. Using this approach a mm-wave chirp signal can be delivered, in a broadband fashion, to the antenna presented in Section V. A photo of the assembled radar frontend is shown in Fig. 11: the FMCW chip (in black) is connected to a pair of waveguide arrays using gold stud bumps with a height of 60–70 m after thermo-compression, as shown in the inset of Fig. 11. Such bumps are also used to connect dc biasing, RF input and IF signal lines to the FMCW chip. The IC losses after flip-chip assembly were characterized with on-wafer measurements on dummy silicon substrates (i.e., without antenna): feeding the circuit with an input signal of dBm at 12 GHz, an output power of dBm at 96 GHz was measured. After the system assembly (i.e., IC with antenna), the output of the chip cannot be accessed by wafer probes anymore, and a radiation measurement was carried out to verify the up-converter performances. A 10 dB standard gain horn antenna was placed above the center of the array at a distance of 5 cm, as shown in Fig. 12, and connected to a spectrum analyzer, which measured a received power of dBm at 96 GHz. The power transmitted at the input of the antenna was then calculated using the Friss transmission equation [30] (all the terms are expressed in decibels)
Fig. 11. Photos of the radar transmitter: top (a) and side (b) view. The flip-chip (in black) lies on top of two Dolph-Tschebyscheff waveguide arrays (in white). The coplanar to waveguide transition used for interfacing the antenna to the chip is visible at the end of the two unconnected array at the bottom of (a).
Fig. 12. Measurement setup of the radar transmitter front-end at broadside. The power level and the frequency of the signal are also reported.
(2) where the distance between the two antennas (i.e., 5 cm) was increased of to account for the phase center shift of the horn antenna [30], the transmitter antenna gain is computed with HFSS simulations, the receiver antenna gain is 10 dB, the transition insertion loss is extracted from measurements, and is the insertion loss of the waveguide sections used to connect the horn antenna to the spectrum analyzer (waveguide attenuation constant 2.73 dB/m from component
specifications). As a result, a transmitted power of dBm was obtained at the output of the IC, which very well agrees with the direct power measurement after flip-chip. Afterwards, the horn antenna is replaced by an open-ended waveguide probe (as shown in Fig. 8), and the radiation pattern of the radar transmitter is measured. Fig. 13 shows a good agreement of the main beam in the far field pattern at broadside of the Dolph-Tschebyscheff array when standing alone and embedded in the system. The boresight direction is shifted of less than 1 in
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VI. CONCLUSION
Fig. 13. Measured radiation pattern at 96 GHz of the stand-alone (before flipchip) and in-system (after flip-chip) antenna.
A new technology platform has been presented for the realization of mm-wave silicon filled rectangular waveguides and high performance frequency scanning antennas. Fully metalized silicon-filled waveguide components, with low-losses (0.15 dB/ at 105 GHz) have been reported. Using these waveguides, a frequency scanning slotted-waveguide array with a Dolph-Tschebyscheff distribution was designed and tested, providing a broadside HPBW of 8.5 . The measured results show excellent agreement with simulation predictions in terms of scanning angle, beam width and side lobe level, confirming the high accuracy of the chosen approach. To interface the fabricated waveguide components, a low-loss (0.6 dB at 105 GHz) wideband coplanar to waveguide transition was introduced to facilitate direct flip-chip interfacing with surface mountable ICs and MMICs. This approach was demonstrated by the successful assembly of an integrated imaging radar system front-end. Using the technology platform presented in this work, high-performance mm-wave systems featuring multiple antennas and/or smart antenna operations can be manufactured with high reproducibility and cost competitive level. ACKNOWLEDGMENT
Fig. 14. Measured radiation pattern at different frequencies (-stand-alone antenna; - - - antenna in-system).
the whole range, which corresponds to an alignment error [42] of the center of the near field scanning plane with respect to the antenna under test of less than 200 m (the alignment was performed only by visual inspection). The side lobe around to the left of the main lobe is practically unchanged, while the side lobe on the right increases from dB to dB due to radiation from the flip-chip substrate. Note that the coplanar to waveguide transition that interfaces the active chip to the substrate is based on a resonant slot design, which normally radiates a small amount of energy (approximately [43], i.e., 2.4%) in free space rather than inside the waveguide. HFSS simulations confirm that this fraction significantly increases when a component with a high dielectric constant (i.e., the active chip) is mounted on top of the transition (the soldier bumps height is only 60–70 m). Since the power radiated by each array slot is quite small (see Table I), the transition contributes to the near field of the array by creating an unwanted side lobe around . Therefore, the reconstructed far field of the radar transmitter is not accurate toward the end-fire direction, and only the angular range between and is shown in Fig. 13. Changing the frequency of the synthesizer from 11.25 to 12.5, the direction of broadside radiation was steered as illustrated in Fig. 14. The main beam scans from at 90 GHz to 14 at 100 GHz, for a total scanning capability of 40 using a frequency bandwidth of 11% (i.e., 10 GHz).
The authors would like to thank the staff of the DIMES group, A. Akhnoukh, and W. Straver, Delft University of Technology, The Netherlands, for their assistance and R. Jackson, University of Massachusetts, and M. Simeoni, Delft University of Technology, The Netherlands, for many useful discussions. REFERENCES [1] D. Liu and U. Pfeiffer, Advanced Millimeter-Wave Technologies. Hoboken, NJ, USA: Wiley, 2009. [2] N. Llombart, A. Neto, G. Gerini, and P. de Maagt, “Planar circularly symmetric EBG structures for reducing surface waves in printed anternnas,” IEEE Trans. Antennas Propag., vol. 53, no. 10, pp. 3210–3218, Oct. 2005. [3] R. W. Jackson, “Mode conversion at discontinuities in finite-width conductor-backed coplanar waveguide,” IEEE Trans. Microw. Theory Tech., vol. 37, no. 10, pp. 1582–1589, Oct. 1989. [4] H. Uchimura, T. Takenoshita, and M. Fujii, “Development of the “laminated waveguide”,” IEEE Trans. Microw. Theory Tech., vol. 46, no. 12, pp. 2438–2443, Dec. 1998. [5] D. Deslandes and K. Wu, “Accurate modeling, wave mechanisms, and design considerations of a substrate integrated waveguide,” IEEE Trans. Microw. Theory Tech., vol. 54, no. 6, pp. 2516–2526, Jun. 2006. [6] F. Xu and K. Wu, “Guided-wave and leakage characteristics of substrate integrated waveguide,” IEEE Trans. Microw. Theory Tech., vol. 53, no. 1, pp. 66–73, Jan. 2005. [7] L. Yan, W. Hong, G. Hua, J. Chen, K. Wu, and T. J. Cui, “Simulation and experiment on SIW slot array antennas,” IEEE Microw. Wireless Compon. Lett., vol. 14, no. 9, pp. 446–448, Sep. 2004. [8] E. Moldovan, R. G. Bosisio, and K. Wu, “W-band multiport substrateintegrated waveguide circuits,” IEEE Trans. Microw. Theory Tech., vol. 54, no. 2, pp. 625–632, Feb. 2006. [9] X. Wang and A. Stelzer, “A 79-GHz LTCC laminated waveguide to metallic rectangular waveguide transition using high permittivity material,” in Proc. European Microwave Conf. (EuMC), Sep. 28–30, 2010, pp. 664–667. [10] Y.-J. Cheng, W. Hong, and K. Wu, “94 GHz substrate integrated monopulse antenna array,” IEEE Trans. Antennas Propag., vol. 60, no. 1, pp. 121–129, Jan. 2012. [11] M. Ettorre, A. Neto, G. Gerini, and S. Maci, “Leaky-wave slot array antenna fed by a dual reflector system,” IEEE Trans. Antennas Propag., vol. 56, no. 10, pp. 3143–3149, Oct. 2008.
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[34] M. J. Pelk, “Near field characterization of integrated antenna’s at (sub)/mm-wave frequencies.,” presented at the Workshop WFS06 (EuMC)/EuMIC) Silicon Characterization From MHz to THz, 2010. [35] A. G. Repjar, A. C. Newell, and M. H. Francis, “Accurate determination of planar near-field correction parameters for linearly polarized probes,” IEEE Trans. Antennas Propag., vol. 36, no. 6, pp. 855–868, Jun. 1988. [36] F. N. M. Isa, M. Ash, and P. V. Brennan, “Preliminary antenna system design for FMCW avalanche radar,” in Proc. XXXth URSI General Assembly and Scientific Symp., Aug. 13–20, 2011, pp. 1–4. [37] R. S. Elliott, Antenna Theory and Design, ser. Series on Electromagnetic Wave Theory. New York, NY, USA: IEEE Press, 2003, pp. 148–157. [38] P. A. Rizzi, A Microwave Engineering: Passive Circuits. Upper Saddle River, NJ, USA: Prentice-Hall, 1988, pp. 541–548. [39] Y. Arai, M. Sato, H. T. Yamada, T. Hamada, K. Nagai, and H. Fujishiro, “60-GHz flip-chip assembled MIC design considering chip substrate effect,” IEEE Trans. Microw. Theory Tech., vol. 45, no. 12, pp. 2261–2266, Dec. 1997. [40] T. H. Lee, The Design of CMOS Radio-Frequency Integrated Circuits. Cambridge, U.K>: Cambridge Univ. Press, pp. 144–146. [41] H.-Y. Lee, “Wideband characterization of a typical bonding wire for microwave and millimeter-wave integrated circuits,” IEEE Trans. Microw. Theory Tech., vol. 43, no. 1, pp. 63–68, Jan. 1995. [42] A. C. Newell, “Error analysis techniques for planar near-field measurements,” IEEE Trans. Antennas Propag., vol. 36, no. 6, pp. 754–768, Jun. 1988. [43] N. G. Alexopoulos, P. B. Katehi, and D. B. Rutledge, “Substrate optimization for integrated circuit antennas,” IEEE Trans. Microw. Theory Tech., vol. 31, no. 7, pp. 550–557, Jul. 1983.
Gennaro Gentile (S’10) received the B.Sc. (cum laude) and M.Sc. degrees (cum laude) in electrical engineering from the University of Naples “Federico II,” Naples, Italy, in 2004 and 2006, respectively. He joined the Delft Institute of Microsystems and Nanoelectronics (DIMES), Delft University of Technology, the Netherlands, in 2006 to pursue the Ph.D. degree to do postdoctoral research in 2012. From 2008 to 2012, he worked on the project “Merging Electronics and Micro & Nano-Photonics in Integrated Systems”, where he was involved in the development of a technology platform for the integration of mm-wave systems using metallic waveguides. Since September 2012, he has been working on near-field characterization of antennas for novel communication links. His current research interests include the modeling, design, manufacturing, and measurement of millimeter-wave integrated phased array antennas. Mr. Gentile is the recipient in 2012 of the Research Excellence Grant from the European Association of National Metrology Institutes (EURAMET) and of the Gallium Arsenide (GAAS) Association Student Fellowship.
Vladimir Jovanović (S’98–M’09) received the B.Sc., M.Sc., and Ph.D. degrees from the Faculty of Electrical Engineering and Computing, University of Zagreb, Zagreb, Croatia, in 1999, 2004, and 2008, respectively. In 1999, he joined the Department of Electronics, Microelectronics, Computer, and Intelligent Systems of the Faculty of Electrical Engineering and Computing, University of Zagreb, where he became an Assistant Professor in 2010. From December 2005 to September 2007, he was a visiting researcher at the Delft Institute of Microsystems and Nanoelectronics (DIMES), Delft University of Technology, Delft, The Netherlands. Since March 2010, he has been a postdoctoral researcher at DIMES. His research involves novel CMOS devices, in particular, FinFETs and other fully-depleted devices, and the integration of silicon devices for RF and microwave applications.
GENTILE et al.: SILICON-FILLED RECTANGULAR WAVEGUIDES AND FREQUENCY SCANNING ANTENNAS
Marco J. Pelk (S’06) was born in Rotterdam, The Netherlands, in 1976. He received the B.Sc. degree in electrical engineering from The Hague Polytechnic, The Hague, The Netherlands, in 2000, and is currently pursuing the Ph.D. degree at the Delft Institute of Microsystems and Nanoelectronics (DIMES), Delft University of Technology, Delft, The Netherlands. In 2000, he joined DIMES. From 2000 to 2002, he was involved in the implementation of compact and mixed-level device models for circuit simulation. Beginning in 2002, he was also involved with the development of a novel active harmonic load-pull system, as well as the design and practical realization of highly efficient power amplifier concepts together with a custom-made measurement setup to characterize its performance. He has authored or coauthored over 20 technical papers. He holds several patents. His current research interests are microwave circuit design, nonlinear device characterization, near-field antenna measurement techniques, and radar systems. Mr. Pelk was a recipient of the IEEE MTT-S 2008 Microwave Prize.
Lai Jiang graduated from Tongji University, Shanghai, China, in electrical engineering in 2006. From 2007 to 2009, he was enrolled in the microelectronics M.S. program at the Delft University of Technology, Delft, The Netherlands. He was as a research assistant at the Delft University of Technology for 9 months on a project related to his thesis. Afterward, he went to pursue a career in the field of applied mathematics.
Ronald Dekker received his M.Sc. degree from the Technical University of Eindhoven, Eindhoven, The Netherlands, on the topic of GaAs mesfet modeling (cum laude) and the Ph.D. degree from the Delft University of Technology, Delft, The Netherlands, on the topic of substrate transfer (cum laude) in 2004. He joined Philips Research in 1988 where he worked on the development of RF technologies for mobile communication. In this context, he developed the concept of substrate transfer. From 2000 onward, his research focus has been on flexible electronics and MEMS technology. In 2007, he was appointed part-time Professor in the Electronic Components Technology and Materials (ECTM) Group of the Electrical Engineering faculty. He has published in leading journals and conferences and holds 45 patents. Dr. Dekker he received the Philips Research “Gilest Holst” peer award in 2000.
P. de Graaf, photograph and biography not available at the time of publication.
Behzad Rejaei received the M.Sc. degree in electrical engineering from the Delft University of Technology, Delft, the Netherlands, in 1990, and the Ph.D. degree in theoretical condensed matter physics from the University of Leiden, Leiden, the Netherlands, in 1994. From 1995 to 1997, he served as a member of the Physics faculty at the Delft University of Technology, where he carried out research on mesoscopic chargedensity-wave systems. Between 1997 and 2010, he was with the Department of Electrical Engineering, Mathematics, and Computer Science, Delft University of Technology. He is currently an Associate Professor with the Department of Electrical Engineering, Sharif University of Technology, Tehran, Iran. His research interests are in the area of electromagnetic modeling of integrated passive components, microwave magnetic devices, and magnetic metamaterials.
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Leo C. N. de Vreede (M’01–SM’04) was born in Delft, the Netherlands, in 1965. He received the B.S. degree (cum laude) in electrical engineering from the Hague Polytechnic, The Hague, The Netherlands, in 1988, and the Ph.D. degree (cum laude) from the Delft University of Technology, Delft, the Netherlands, in 1996. In 1988, he joined the Laboratory of Telecommunication and Remote Sensing Technology, Department of Electrical Engineering, Delft University of Technology. In 1996, he was appointed as Assistant Professor at the Delft University of Technology working on the nonlinear distortion behavior of bipolar transistors at the Delft Institute of Microelectronics and Submicron Technology (DIMES). In the winter season of 1998–1999, he was a guest of the High Speed Device Group at the University of San Diego, California, La Jolla, CA, USA. In 1999, he became an Associate Professor, responsible for the Microwave Components Group of the Delft University of Technology. Since that time, he has worked on RF solutions for improved linearity and RF performance at the device, circuit, and system level. He is co-founder/advisor of Anteverta-mw. He has (co)authored more than 100 IEEE refereed conference and journal papers and holds several patents. His current interest includes RF measurement systems, technology optimization and circuit concepts for wireless systems. Dr. de Vreede was (co)recipient of the IEEE Microwave prize in 2008, mentor of the Else Kooi prize awarded Ph.D. work in 2010, and mentor of the Dow Energy dissertation prize awarded Ph.D. work in 2011. Lis K. Nanver (S’80–M’83) received the M.Sc. degree in physics from the University of Aarhus, Aarhus, Denmark, in 1979; the Dr.ing. degree from the Ecole Nationale Supeure des Télécommunications, Paris, France, in 1982, where she worked on the simulation of charge-coupled device structures; and the Ph.D. degree from the Delft University of Technology, Delft, The Netherlands, in 1987, where she developed a medium-frequency bipolar field-effect transistor process. In 1988, she joined the Delft Institute for Microsystems and Nanoelectronics (DIMES) IC Process Research Sector as Bipolar Process Research Manager. She became an Associate Professor and later Professor with the Faculty of Electrical Engineering, Mathematics, and Computer Science, Delft University of Technology, working at the DIMES Technology Center in 1994 and 2001, respectively. As head of the Silicon Device Integration group of the Microelectronics Department , she manages research on the integration of silicon devices, mainly for RF/microwave applications and photodiode detectors. This research involves several advanced technologies for ultrashallow junction formation by chemical-vapor or metal-induced solid-phase deposition and excimer laser processing, as well as substrate transfer technologies for two-sided device contacting. Prof. Nanver has served on the committees of the European Solid-State Device Research Conference and Bipolar/BiCMOS Circuits and Technology Meeting. Marco Spirito (S’01–M’08) received the M.Sc. degree (cum laude) in electrical engineering from the University of Naples “Federico II,” Naples, Italy, in 2000, and the Ph.D. degree from the Delft University of Technology, Delft, The Netherlands, in 2006. In August 2000, he joined the Faculty of Electrical Engineering, Mathematics and Computer Science, Laboratory of High-Frequency Technology and Components, Delft University of Technology, where he was involved in the design, optimization, and characterization of high-performance and linear power amplifiers. From 2000 to 2001, he was a guest at Infineon Technologies, Munich, Germany. In 2006, he joined the department of Electronics and Telecommunications Engineering, University of Naples “Federico II.” Since April 2008, he has been an Assistant Professor with the Electronics Research Laboratory, Delft University of Technology. His research interests include the characterization of highly efficient and linear power amplifiers, the development of advanced characterization setups for millimeter and sub-millimeter waves, and the integration of mm-wave sensing systems. Dr. Spirito was the recipient of the Best Student Paper Award for his contribution to the 2002 IEEE Bipolar/BiCMOS Circuits and Technology Meeting (BCTM), he received the IEEE MTT Society Microwave Prize in 2008, and was a co-recipient of the best student paper award at IEEE RFIC 2011.
