Simple Power Control for Sensorless Induction Motor ... - IEEE Xplore

IEEE TRANSACTIONS ON ENERGY CONVERSION, VOL. 23, NO. 3, SEPTEMBER 2008

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Simple Power Control for Sensorless Induction Motor Drives Fed by a Matrix Converter Kyo-Beum Lee, Member, IEEE, and Frede Blaabjerg, Fellow, IEEE

Abstract—This paper presents a new and simple method for sensorless control of matrix converter drives using a power flowing to the motor. The proposed control algorithm is based on controlling the instantaneous real and imaginary powers into the induction motor. To improve low-speed sensorless performance, the nonlinearities of a matrix converter drive such as commutation delays, turn-ON and turn-OFF times of switching devices, and ONstate switching device voltage drop are modeled using a PQ power transformation and compensated using a reference power control scheme. The proposed sensorless control method is applied for the induction motor drive using a 3 kW matrix converter system. Experimental results are shown to illustrate the feasibility of the proposed strategy. Index Terms—AC motor drive, ac–ac power conversion, observers.

I. INTRODUCTION NDUCTION motor drives fed by a matrix converter have been developed for the last decades [1]. The matrix converter drive has recently attracted the industry application and the technical development has been further accelerated because of increasing importance of power quality and energy efficiency issues. The induction motor drive fed by matrix converter is superior to the conventional inverter because of the lack of the bulky dc-link capacitors with limited life time, the bidirectional power flow capability, the sinusoidal input/output currents, and adjustable input power factor. Furthermore, because of the high integration capability and the higher reliability of the semiconductor structures, the matrix converter topology is recommended for extreme temperatures and critical volume/weight applications. However, a few of the practical matrix converters have been applied to induction motor drive system because the implementation of switch devices in the matrix converter is difficult. Also, modulation technique and commutation control are more complicated than the conventional pulsewidth modulation (PWM) inverter [1]–[8]. The instantaneous power control (IPC) scheme for induction motor drives was initially presented in [13]. It is a relatively new high-performance control scheme for induction motors. It is based on controlling the instantaneous real and imaginary powers into the induction motor. It is motivated by the fact that

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Manuscript received September 21, 2007; revised September 21, 2007. Paper no. TEC-00407-2006. K. B. Lee is with the School of Electrical and Computer Engineering, Ajou University, Suwon 443-749, Korea (e-mail: [email protected]). F. Blaabjerg is with the Institute of Energy Technology, Aalborg University, Aalborg DK-9220, Denmark (e-mail: [email protected]). Color versions of one or more of the figures in this paper are available online at http://ieeexplore.ieee.org. Digital Object Identifier 10.1109/TEC.2008.921472

real power controls the torque produced by the motor and the imaginary power controls the flux. The advantages of the IPC are simple algorithm implementation, fundamental physical-lawsbased algorithm, robustness and limited dependence on detailed knowledge of parameters, and flexibility applicable to all sinewave ac machine. The IPC algorithm conceptually lies somewhere between field oriented control (FOC) and direct torque control (DTC). It has the switching frequency predictability of FOC and the stationary reference frame implementation simplicity of DTC. In this paper, the synchronous reference-framebased IPC is presented for a high-performance matrix converter drive. In order to build a speed sensorless control algorithm for the induction motor drives fed by a matrix converter, more accurate information on the motor speed and more robust against motor parameters are required. In particular, in the low-speed operation, the machine-model-based observer does not produce fairly satisfactory performance without a parameter adaptation scheme when the motor parameters vary in operation. Since the fundamental output voltage cannot be detected directly from the matrix converter output terminal in a matrix converter drive, the calculated average voltage is normally used instead of the actual one. However, the calculated average voltage does not agree with the actual fundamental output voltage due to nonlinear characteristic of the matrix converter, such as commutation delay, turn-ON and turn-OFF time of switching device, and ONstate switching devices voltage drop. In order to compensate this problem, especially important at low speed, the authors have made an attempt to compensate the nonlinear matrix converter effects with current sign and off-line manners [6], [7]. However, it is difficult to determine the current sign when the phase currents are close to zero. If the current sign is misjudged, the nonlinearity model of a matrix converter operates improperly. It is also difficult to compensate the nonlinearity effects perfectly by off-line manners because the switching times and voltage drops of the power devices are varied with the operating conditions [8], [10]. In this paper, a new sensorless power control method for matrix converter drives using a power theory is proposed. The whole control block diagram of a sensorless power-controlled matrix converter drive is shown in Fig. 1, which consists of a power reference generation, speed observer and nonlinearity compensation, the detection of input voltage angle, and indirect space vector modulation. The proposed sensorless technique detects the speed of the rotor of an induction motor using the constant air gap flux and the imaginary power flowing to the motor. The paper calculates the nonlinearity of matrix converter drives using the power transformation, and the reference power

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Fig. 1.

Proposed sensorless vector control for matrix converter drives.

control method applies to make the compensation currents in order to improve speed sensorless operation in the low-speed region. Experimental results derived from a test drive system are presented to show the performance of the proposed observer. II. POWER CONTROL OF A MATRIX CONVERTER DRIVE The IPC is based on controlling the instantaneous real and imaginary powers into the induction motor. It is motivated by the fact that real power controls the torque produced by the motor and the imaginary power controls the flux. The proposed topology of the power control is shown in Fig. 1. The controller contains real and imaginary power PI-regulators. It receives as inputs the torque reference, the real and imaginary powers and generates the inverter’s command voltages in a synchronous reference frame. A. Instantaneous Active and Reactive Power In this section, instantaneous active and reactive powers are defined by the scalar and vector product of the voltage and the current space vectors [12]. The three phase voltages and currents in abc-coordinates can be transformed in αβ-coordinates as   1 1 1 √ √ √      2 2 2    va vo   1 1    vα  = 2  1 −   vb 3 2 2   vc vβ √ √  3 3 − 0 2 2 and   1 1 1 √ √ √    2   2 2    ia io   2 1 1  1  iα  =   (1) −  −  ib . 3 2 2   iβ i √ √ c  3 3 − 0 2 2 Active and reactive powers are defined as ¯iα + ˜iα P P¯ P˜ vα vβ = ¯ + ˜ = (2) ¯iβ + ˜iβ Q Q Q −vβ vα

¯ are the average active and reactive power dc where P¯ and Q term originated from the symmetrical fundamental component of the load current (¯iα and ¯iβ ), P¯ is the average active power ¯ is the average reactive power circulating delivered to the load, Q ˜ are the active and reactive ac ripple between the phases, P˜ and Q term originating from harmonic and asymmetrical fundamental component of the load current (˜iα and ˜iβ ), P˜ is the ripple of active power that corresponds to power oscillations between ˜ is the reactive power’s ripple that the source and load, and Q corresponds to power oscillations between the phases. As it can be seen from (2), the instantaneous powers P and Q comprise two parts, one is a dc component and the other one is an ac component. The instantaneous powers can be separated into a dc part and an ac part by filters without energy-storage element since the average values of these are zero. The ac parts ˜ are useless powers which come from various nonP˜ and Q ideal circuit conditions such as unbalanced voltage and loads, distorted system voltages, and nonlinear loads [12]. B. Power Reference Generation If The instantaneous real power P into the motor can be separated into three terms such as the power dissipated in the stator and rotor resistance, the time rate of change of the magnetic energy stored in the inductances in the motor, and the power for energy conversion [11]. This means some power is lost in the stator and rotor resistance and iron loss of the motor. While the stator resistance power and iron losses can be ignored, the power in the rotor resistance cannot be ignored, especially under heavy load conditions when the slip of the motor can be large. In this paper, a simple compensation method for those losses will be proposed later. The instantaneous imaginary power Q into the motor is related to the instantaneous rate of magnitude change or vector acceleration of the flux vector in the motor. The space vector diagram for power control can be seen as Fig. 2. The Q-axis imaginary power reference considered the slip frequency of the motor that can be shown as Q∗ = im vEM F = im ωe λm = im (P ωm + ωsl ) λm

(3)

where im is the magnetising current, vEM F is the back EMF voltage, ωe is the electrical frequency, λm is the flux magnitude,

LEE AND BLAABJERG: SIMPLE POWER CONTROL FOR SENSORLESS INDUCTION MOTOR DRIVES

Fig. 2.

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Space vector diagram for power control.

P is the motor pole pairs, ωm is the mechanical frequency, and ωsl is the slip frequency. The slip frequency of the motor is intimately related to the torque. The logical extension to this is that the desired slip frequency is related to the desired torque [13]. The slip equation with the torque expression can be written as ωsl =

Te ip = . 2 Tr |im | (3/2) (Lm /Lr ) |im |2 Tr

Fig. 3.

PQ equivalent circuit of an induction motor.

the synchronous angular frequency, and ωr is the rotor angular frequency. The flux equations are given as

(4)

λQ s = λQ m + Lls iQ s λP s = λP m + Lls iP s

By using (3) and (4), the slip can be expressed as s=

ωr Lm i2m ωsl =1+ . ωr + ωsl Q∗

λQ r = λQ m + Llr iQ r

(5)

λP r = λP m + Llr iP r

The real power reference can be written from (5) as Q∗ T ∗ Pm ech =− P∗ = . 1−s pp Lm i2m

λQ m = Lm iQ m λP m = Lm iP m

(6)

Assuming that the leakage inductance of the motor is very small related to the Lr , the Q-axis imaginary power reference can be written as 2T ∗ . (7) Q∗ = Lm i2m ωr + 3P Tr

where λP m and λQ m are the P - and Q-axis air gap fluxes, Lls and Llr are the stator and rotor leakage inductances, and Lm is the magnetizing inductance. The electromagnetic torque is in terms of d- and q-axis components Te =

III. SPEED OBSERVER USING POWER THEORY A. Constant Air Gap Flux Operation An induction motor can be described by the following voltage equations. The corresponding equivalent circuit in the rotating PQ reference frame is shown in Fig. 3 d λQ s − ωe λD s dt d vP s = Rs iP s + λP s + ωe λQ s dt d 0 = Rr iQ r + λQ r − (ωe − ωr ) λP r dt d 0 = Rr iP r + λP r + (ωe − ωr ) λQ r (8) dt where vP s and vQ s are the stator P - and Q-axis voltages, iP s and iQ s are the stator P - and Q-axis currents, iP r and iQ r are the rotor P - and Q-axis currents, λP s and λQ s are the P - and Q-axis stator fluxes, λP r and λQ r are the P - and Q-axis rotor fluxes, Rr is the stator resistance, Rr is the rotor resistance, ωe is

(9)

3p (iQ s λP m + iP s λQ m ) . 22

(10)

The stator voltage equation in (8) can be rewritten using (9) as d (λQ m + Lls iQ s ) − ωe (λP m + Lls iP s ) dt d (λP m + Lls iP s ) + ωe (λQ m + Lls iQ s ) . + dt (11)

vQ s = Rs iQ s + vP s = Rs iP s

vQ s = Rs iQ s +

If the power control is fulfilled such that the time derivative of P - and Q-axis stator currents in the rotating frame can be constant during a control sampling period, the electromagnetic torque is controlled only by P -axis current from (10), and (11) can be rewritten as d λQ m − ωe (λP m + Lls iP s ) dt d + λP m + ωe (λQ m + Lls iQ s ) dt

vQ s = Rs iQ s + vP s = Rs iP s Te =

3p iP s λQ m . 22

(12) (13)

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Ignoring the terms related to the rate of change of the air gap fluxes λQ m andλP m , (12) can be rearranged as vQ s = Rs iQ s − ωe (λP m + Lls iP s ) vP s = Rs iP s + ωe (λQ m + Lls iQ s ) .

(14)

If the voltage drop across the stator resistance is negligible in comparison with |λm | ωe >> |(Rs + jωe Lls ) is |, then |vs | ≈ |vEM F | ≈ |λm | ωe = Lm |im | ωe

(15)

where vEM F is the magnitude of the back EMF voltage. If the variations of the air gap fluxes and stator currents in the rotating frame can be constant, ωe can be calculated from the value of magnetizing current and supply voltage. In order to build a high-performance speed sensorless control algorithm for the induction motor drives fed by a matrix converter, more accurate information on the motor speed and more robust against motor parameters are required. In particular, in the low-speed operation, the machine-model-based observer [7], [9] does not produce fairly satisfactory performance without a parameter adaptation scheme when motor parameters vary in operation. New observer using the constant flux air gap and the imaginary power flowing into the motor is proposed to overcome this problem. The proposed speed observer based on the use of the imaginary power would be insensitive to the knowledge of the resistance because the imaginary power is directly related to flux in the motor and frequency, and not power flow due to the consumption of real power in resistance. The proposed algorithm is also very simple and easy to apply to the real industrial applications.

Fig. 4.

Speed observer and power calculation with nonlinearity compensation.

Fig. 5. Speed response of the proposed speed observer: speed reference and speed, speed error, and Q (simulation result).

B. Design of Speed Observer The instantaneous real power P is effectively the real power being consumed in the slip modified rotor resistance of the motor plus any power stored in the field and the imaginary powers Q is related to the instantaneous rate of magnitude change or vector acceleration of the flux vector in the motor. In steady state, Q becomes equal to the conventional reactive power in a motor [12], [13]. Fig. 2 illustrates a vector diagram for an induction motor, showing the relationship between the currents, voltages, and fluxes. The imaginary power is the voltage in one axis of the motor multiplied by the current in the other axis. The instantaneous imaginary power can be defined as

(17)

The proposed scheme is shown in Fig. 4, which consists of a speed observer, a PQ power transformation, and a reference power control. Fig. 5 shows the simulated waveforms using the ideal matrix converter model for the proposed speed observer. It is shown in Fig. 5 that the estimated speed converges to the actual value by the proposed observer. It can be also seen by this simulation result that the waveform-shape of imaginary power Q is almost the same as the estimated speed. If the matrix converter is no longer considered ideal, the nonlinearity of matrix converter can cause considerable distortion in the voltages, especially at low speed [6]. Therefore, the distortion of the Q would also occur due to this nonlinearity. Fig. 6 shows the speed observer response at the low-speed region. From this simulation result, the influence of the nonlinearity of the matrix converter on the speed estimation has to be investigated and compensated for these influences.

where ωe = ωr + ωsl . Given that Q can be determined from (2), then (17) can be rewritten so that ωr is the subject of the expression:

C. Nonlinearity Compensation for Low-Speed Operation Power Reference Generation

Q ≈ im |vEM F | .

(16)

Equation (16) can be rewritten by using (15) Q = Lm i2m (ωr + ωsl )

ωr =

vα iβ − vβ iα − ωsl Lm i2m

where ωsl = ip /τr im and τr = Lr /Rr .

(18)

As mentioned in the previous section, the voltages due to the nonlinearity of matrix converter drives cause the matrix converter output voltage distortion, and results in the imaginary power distortion and speed ripple. The disturbance voltages are


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Fig. 7. Simulation results for the presented nonlinearity compensation at 100 r/min: real speed and phase currents (the compensation begins to operate after 1 s).

Fig. 6. Speed response of the proposed sensorless observer in the low speed operation (a) with 0.5 s commutation delay and (b) with 2 s commutation delay: speed, P , and Q (simulation result).

a function of a commutation delay, turn-ON and turn-OFF time of the switching device, ON-state switching device voltage drop, and a current polarity [6]. However, the commutation delay, ON-state voltage drop of the switching devices, and a current polarity are varying with the operating conditions. Since it is very difficult to measure the commutation delays and ON-state voltage drops as well as to determine the current sign when the phase currents are closed to zero, it is not easy to compensate the nonlinearity of matrix converter drives in an off-line manner [8], [10]. To alleviate these problems, online nonlinearity compensation method using a power transformation is employed to improve speed sensorless performance. Fig. 6 shows some cases of the real and imaginary powers. In Fig. 6(a), real power P and imaginary power Q comprise almost a dc component. In Fig. 6(b), powers are distorted by the nonlinearities of a matrix converter. The powers P and Q contain an ac component.

As it is shown in the simulation results (see Fig. 6), if the matrix converter currents are balanced and sinusoidal, the P and Q comprise only the dc component. When the currents are unbalanced and distorted by nonlinearity of the matrix converter, the unbalanced and harmonic component of the current exists in the P and Q. To eliminate these unbalanced and distorted currents due to current sensing unbalance and distortion by nonlinearity of matrix converter drives, a reference power control method can be introduced. To make the compensation powers, simple first-order low-pass filters are used to separate the calculated variables into the ac and dc components (see Fig. 6). The classified ac and dc components are used to calculate the compensation current for the current controller. The total powers P and Q are calculated by the PQ transformation, as shown in (2). The instantaneous real power P consists of an ac component P˜ and a dc component P¯ . The Q ˜ and a dc component Q. ¯ The also consists of an ac component Q ˜ ˜ ac components P and Q are related with nonlinear conditions such as harmonics. Therefore, to get the output currents symmetrical and sinusoidal, P and Q have to be controlled to a dc value. The compensation powers can be represented as Pcom p = P˜

and

˜ Qcom p = Q.

(19)

Fig. 7 shows the simulation waveforms for the nonlinearity compensation of matrix converter drive. Before starting the compensation, the nonlinearity causes undesired speed pulsations of about six times the electrical frequency. The speed estimation error is almost eliminated after starting the compensation. IV. EXPERIMENTS To confirm the validity of the proposed control algorithm, experiments are carried out. The low-speed operation is especially in focus during experiments. The hardware consists of a three-phase, 380 V, 50 Hz, four-pole, 3 kW induction motor, and

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Fig. 8. Dynamic response of speed sensorless control with 40% rated load: speed and phase current.

Fig. 9. Dynamic response of speed sensorless control with 40% rated load: speed and phase current.

power circuit with a matrix converter. The induction motor has the following parameter values: Rs = 1.79 Ω, Rr = 1.8 Ω,Ls = 167 mH, Lr = 174.4 mH, and Lm = 160 mH. A dual controller system consisting of a 32 bit DSP (ADSP 21062) and a 16 bit microcontroller (80C167), in conjunction with a 12 bit A/D converter board is used to control the matrix-converterbased induction motor drive. The modulation used a doublesided support vector machine (SVM) algorithm with minimized number of commutations and the commutation of the bidirectional switches was performed in a four-step way depending on the output current sign [2]. Fig. 8 shows the steady-state characteristics of the estimated speed, real and imaginary powers at 100 r/min in the experiments. The nonlinearities cause undesired speed pulsations of about six times the electrical frequency. Fig. 9 shows the speed and phase current responses of the proposed sensorless power control in various speed references. It is noted here that a very satisfactory dynamic speed response is obtained. Fig. 10 shows the speed and phase current responses under parameter-mismatched condition. The controller parameter selected from the incorrect rotor time constant value. It is noted that the relatively big overshoot seems to occur due to

Fig. 10. Dynamic response of speed sensorless control with 40% rated load: speed and phase current (with mismatched motor parameters). (a) T r = 0.6 × T r . (b) T r = 2 × T r .

Fig. 11. Speed sensorless control at 50 r/min with 10% rated load: speed and phase currents.

mismatched parameter. However, it can be said from this results that the dynamic performance of the proposed sensorless power control is insensitive to the parameter variation, especially the rotor time constant.


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In Fig. 11, a very good speed estimation and a sinusoidal phase current waveform were achieved at the low-speed region down to 50 r/min with 10% rated load. Fig. 12 shows the speed estimation performance and phase current at the high-speed region with 30% rated load. Fig 13 shows the steady-state characteristics of the phase currents at 100 r/min. The nonlinearities cause undesired distortions of about six times the electrical frequency without nonlinearites compensation, as shown in Fig. 13(a). However, the current pulsations and their fifth and seventh harmonics were remarkably reduced with the proposed compensation method using the power theory shown in Fig. 13(b).

V. CONCLUSION Fig. 12. Speed sensorless control at 900 r/min with 30% rated load: speed and phase currents.

A new and simple sensorless power control for induction motor drives fed by matrix converter using the constant air gap flux and the imaginary power flowing to the motor has been proposed in this paper. The proposed sensorless scheme was independent of the value of the motor parameter. In addition, a nonlinearity compensation strategy using a power theory has been employed to make the speed control performance better in the low-speed region. The experimental results have shown that the satisfactory speed sensorless performance in various speed operations has been obtained.

REFERENCES

Fig. 13. Output currents and harmonic analysis [fast Fourier Transform (FFT)] at 100 r/min with 20% rated load. (a) Without nonlinearity compensation. (b) With nonlinearity compensation.

[1] M. P. Kazmierkowski, R. Krishnan, and F. Blaabjerg, Control in Power Electronics—Selected Problems. New York: Academic, 2002, ch. 3 (ISBN 0-12-402772-5). [2] P. Nielsen, F. Blaabjerg, and J. K. Pedersen, “New protection issues of a matrix converter: Design considerations for adjustable-speed drives,” IEEE Trans. Ind. Appl., vol. 35, no. 5, pp. 1150–1161, Sep./Oct. 1999. [3] P. W. Wheeler, J. Rodriguez, J. C. Clare, L. Empringham, and A. Weinstein, “Matrix converter: A technology review,” IEEE Trans. Ind. Electron., vol. 49, no. 2, pp. 276–288, Apr. 2002. [4] L. Huber and D. Borojevic, “Space vector modulated three-phase to threephase matrix converter with input power correction,” IEEE Trans. Ind. Appl., vol. 31, no. 6, pp. 1234–1246, Nov./Dec. 1995. [5] C. Klumpner, P. Niesen, I. Boldea, and F. Blaabjerg, “A new matrix converter-motor (MCM) for industry applications,” IEEE Trans. Ind. Electron., vol. 49, no. 2, pp. 325–335, Apr. 2002. [6] K. B. Lee and F. Blaabjerg, “A nonlinearity compensation method for a matrix converter drive,” IEEE Power Electron. Lett., vol. 3, no. 1, pp. 19–23, Mar. 2005. [7] K. B. Lee and F. Blaabjerg, “Reduced order extended luenberger observer based sensorless vector control driven by matrix converter with non-linearity compensation,” IEEE Trans. Ind. Electron., vol. 53, no. 1, pp. 66–75, Feb. 2006. [8] K. B. Lee and F. Blaabjerg, “Performance improvement of sensorless vector control for matrix converter drives using PQR power theory,” Electr. Eng., vol. 89, no. 8, pp. 607–616, Sep. 2007. [9] K. B. Lee and F. Blaabjerg, “Sensorless DTC-SVM for induction motor driven by a matrix converter using a parameter estimation strategy,” IEEE Trans. Ind. Electron., vol. 55, no. 2, pp. 512–521, Feb. 2008. [10] R. E. Betz and T. Summers, “Instantaneous power control—An alternative to vector and direct torque control?,” in Proc. IAS 2000, pp. 1640– 1647. [11] N. Urasaki, T. Senjyu, K. Uezato, and T. Funabashi, “A dead-time compensation strategy for permanent magnet synchronous motor drive suppressing current distortion,” in Proc. IEEE IECON 2003, pp. 1255– 1260.

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[12] F. Z. Peng, G. W. Ott, Jr., and D. J. Adams, “Harmonic and reactive power compensation based on the generalized instantaneous reactive power theory for three-phase four-wire systems,” IEEE Trans. Power Electron., vol. 13, no. 6, pp. 1174–1181, Nov. 1998. [13] D. W. Novotny and T. A. Lipo, Vector Control and Dynamics of AC Drives. Oxford, U.K.: Oxford Univ. Press, 1997 (ISBN 0-19-856439-2). [14] J. Holtz, “Sensorless control of induction motor drives,” Proc. IEEE, vol. 90, no. 8, pp. 1359–1394, Aug. 2002.

Kyo-Beum Lee (S’02–M’04) was born in Seoul, Korea, in 1972. He received the B.S. and M.S. degrees in electrical and electronic engineering from Ajou University, Suwon, Korea, in 1997 and 1999, respectively, and the Ph.D. degree in electrical engineering from Korea University, Seoul, Korea, in 2003. From 2003 to 2006, he was with the Institute of Energy Technology, Aalborg University, Aalborg, Denmark. From 2006 to 2007, he was with the Division of Electronics and Information Engineering, Chonbuk National University, Jeonju, Korea. In 2007, he joined the School of Electrical and Computer Engineering, Ajou University. His current research interests include electric machine drives and wind power generations. Dr. Lee is an Associate Editor of the IEEE TRANSACTIONS ON POWER ELECTRONICS. He is the recipient of the two IEEE Prize Paper Awards.

Frede Blaabjerg (S’86–M’88–SM’97–F’03) received the M.Sc. degree in electrical and electronic engineering from Aalborg University, Aalborg, Denmark, in 1987, and the Ph.D. degree in electrical engineering from the Institute of Energy Technology, Aalborg University, in 1995. From 1987 to 1988, he was with ABB-Scandia, Randers, Denmark. In 1992, he became an Assistant Professor at Aalborg University, where in 1996, he became an Associate Professor, in 1998, a Full Professor in power electronics and drives, and in 2006, the Dean of the Faculty of Engineering and Science. His current research interests include power electronics, static power converters, ac drives, switched reluctance drives, modeling, characterization of power semiconductor devices and simulation, wind turbines, and green power inverter. He is the author or coauthor of more than 300 publications including the book Control in Power Electronics (Academic, 2002). He has held a number of chairman positions in research policy and research funding bodies in Denmark. He is an Associate Editor of the Journal of Power Electronics and the Danish journal Elteknik. Dr. Blaabjerg is an Associate Editor of the IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS and the IEEE TRANSACTIONS ON POWER ELECTRONICS. In 2006, he became an Editor-in-Chief of the IEEE TRANSACTIONS ON POWER ELECTRONICS. He is the recipient of the 1995 Angelos Award for his contribution in modulation technique and control of electric drives, an Annual Teacher Prize at Aalborg University in 1995, and the Outstanding Young Power Electronics Engineer Award from the IEEE Power Electronics Society in 1998. He has also received five IEEE Prize Paper Awards during the last six years, and the C.Y. O’Connor Fellowship 2002 from Perth, Australia, the Statoil Prize in 2003 for his contributions in power electronics, and the Grundfos Prize in 2004 for his contributions in power electronics and drives.