Hexagonal cavities and their applications to multilayer substrate integrated waveguide (SIW) filters are presented. The hexagonal SIW cavity which can combine.
2013 International Workshop
on Microwave and Millimeter Wave Circuits and System Technology
Substrate Integrated Waveguide Cross-Coupling Filter with Multilayer Hexagonal Cavity Wu Bo, Xu Ziqiang, Liu Hao, Xu Meijuan, and Liao Jiaxuan Research Institute of Electronic Science and Technology ofUESTC, Chengdu, Sichuan, 611731, China
of circular cavities, and their applications to SIW filters are proposed. Three types of experimental filter configuration at the same central frequency of 10 GHz are fabricated using low temperature co-fired ceramic ( LTCC) technology. Design details are described, and both simulated and experimental results are presented and discussed.
Abstract Hexagonal cavities and their applications to multilayer substrate integrated waveguide (SIW) filters are presented. The hexagonal SIW cavity which can combine -
flexibility of rectangular one and performance of circular one is convenient for bandpass filter's design. Three types of experimental configuration with the same central frequency of 10 GHz and bandwidth of 6%, including third-order and four order cross-coupling topologies, are constructed and fabricated based on low temperature co-fired ceramic (LTCC) technology.
II. FTLTER ANALYSTS AND DESIGN
Both theoretical and experimental results are presented.
Index Terms
Cross-coupling, hexagonal cavity, low temperature co-fired ceramic (LTCC), multilayer, substrate -
integrated waveguide
A. Hexagonal Resonant Cavity
(SIW).
I.
The configurations and electric field distributions of the fundamental mode of a hexagonal cavity resonator are shown in Fig. I. Its electric field distributions and hence resonant characteristics are similar with circular ones.
INTRODUCTION
Waveguide filters are widely used in various microwave and millimeter-wave communication systems, especially airborne platforms, communication satellites, earth stations, and wireless base-stations, due to their high quality factor (Q-factor) and high power capability. However, they are bulky and not suitable for high-density integration, which greatly increases the cost of the entire system [1]-[2]. Recently substrate integrated waveguide (SIW) which is synthesized in a planar substrate with arrays of metallic via, provides a low-profile, low-weight, and low-cost solution while maintaining high performance [3]-[4]. In the past a few years, much attention has been paid to various types of SIW filters. It is known that conventional structures of SIW filters are predominantly based on rectangular and circular cavities [5]-[6]. Rectangular cavity is superior for its flexible structure while circular cavity is for its high performance [7]-[9]. To combine their predominance together, novel SIW filters with hexagonal resonators are proposed in [10]. Since any of the six sides of a hexagonal cavity can be utilized for coupling, it is flexible in the design of SIW filter. However, their physical sizes are not compact enough due to their planar and single-layer structure. Besides, it is usually not easy to implement negative coupling in planar structure owing to similar coupling manner between the adjacent resonators in the same plane. In this paper, in regard to compact size and stringent frequency selectivity, multilayer hexagonal cavities which combine flexibility of rectangular cavities and performance
978-1-4673-5504-9/13/$31.00 ©2013 IEEE
d
p
Fig. 1.
The configurations and electric field distributions of the
fundamental mode of a hexagonal cavity resonator.
Here, L, d and p stand for length of regular hexagonal cavity, diameter of metallized via holes and pitch between them respectively. There is no accurate formula between geometrical parameters and eigen frequencies of a hexagonal resonator yet, but paper [10] proposed an approximate expression as L=
c
·
-
-F
;./
·
--
2:rh
(1)
Here, c, Gr and f, stand for speed of light, relative permittivity of dielectric substrate and eigen frequencies of SIW hexagonal resonator respectively. 11'=2.75 is the modified root coefficient based on Bessel function. B.
221
Analysis of the Proposed Filters
and capacitance between which stand for coupling relationships. Hence, coupling topological structures of Filters A, Band C are shown in Fig. 2.
Taking three-order and four-order filters as examples, three types of novel multilayer SIW hexagonal cavity filters are proposed, with design goals illustrated in Table I. DESIGN
TABLE I GOALS OF THREE FILTERS
Center
Fractional
Return Transmission
Frequency Bandwidth
Loss
Zeros
Filter A
10 GHz
6%
20 dB
10.3 GHz
Filter B
10 GHz
6%
20 dB
9.6 GHz
Filter C
10 GHz
6%
20 dB
9.6, 10.5 GHz
Filter A Fig. 2.
Abstracted coupling coefficients determine coupling patterns and initial geometrical dimensions, while calculating normalized frequency of prototype lowpass filter in advance as
For Filters A and C, their coupling matrixes have negative elements on secondary diagonal, unlike direct coupling. Magnetic coupling pattern contributes to achieve cross coupling in multilayer structure. For Filter A, S1l remains and S21 inverses while inversing M/3, M.u, Mn and M21 simultaneously. For Filter C, SI1 remains and S21 inverses while inversing M14, M4h Mn and M21 simultaneously. Transformed coupling matrixes and their Q-factors are shown as following.
Ms
Mc
[ =
=
=
[
[
0.0341
0.0082
0.0546
0.0546
-0.0325 0.0546
0.0341
0.0546
QA
=
]
15.3857
-0.0082
0.0546
0.0341
0.0546
0.0325
-0.0546
0.0341
-0.0546
Qs
15.3857
0
=
Parameters of the Proposed Filters
External Q-factors determined by input and output structures play a crucial role in suppressing return loss within pass band. Input and output adopt electric current probe form, which consist of GCPW and 50 n microstrip line. Width of GCPW's gap is assigned as 0.25 mm for preventing power leakage and machining tolerance. Effect of GCPW's length on return loss in Filter A is shown in Fig. 3. Loaded Q-factor varies due to length's changing, causing variation of return loss within pass band. Hence, it is important for ports' dimensions to return loss within pass band. 0 ,--------..
(4)
co
0
0.0384
0
0.0384
0
0.0085
0
0.0435
Q)
u
::E
�rl
·10
u
c c:n ro
-0.0435
19.5396
Coupling topological structures of Filters A, B and C.
�
0
=
]
-0.0082
0.M35
Qc
C.
(3)
0.0082
FilterC
For Filter A, signal passed primary channel obtains opposite phase against that passed cross-coupling channel, forms a transmission zero beyond pass band. Similarly, Filter B forms a transmission zero underneath pass band while Filter C forms two at both outsides of pass band. Magnetic coupling achieves by slotting at the area with maximum magnetic field intensity on common surface between cavities, while electric coupling with maximum electric field intensity.
(2)
MA
Filter B
.,
Lu·20
\;' ;:
\
·30
·······
·40
(5)
L u= 3 . 4mm
Lu=3.2mm ........... Lu=3.0mm --
8
9
10
11
12
Frequency(Ghz) Fig. 3.
Effect of GCPW's length on return loss in Filter A.
Transmission zero comes from cross-coupling when signal passed primary channel obtains 1800 phase shift against that passed cross-coupling channel at corresponding frequency point. Therefore, cross-coupling has a great impact on transmission zero. Effect of coupling window's length on transmission zero in Filter B is shown in Fig. 4. Coupling intensity between two resonant cavities enhances
Normally, coupling coefficient is positive correlated with size of coupling window, which can be preliminary determined by their relational curve graph. We defme negative coupling coefficient as electric coupling and positive coupling coefficient as magnetic coupling. Numbered circle stands for resonator, inductance
222
as length elongating, causing transmission zero underneath pass band moves downward.
holes having diameter of 0.17 mm and center-to-center pitch of 0.45 mrn. Fig. 6 shows the photograph of fabricated Filters A, Band C. The external shapes of fabricated filters are 22 mrnx1l.5 mrnxO.8 mrn, 22 mrnx1l.5 mrnxO.8 mrn, and 22 mrnx12 mrnxO.8 mrn respectively. An Agilent E8363B vector network analyzer is used for measurement. The simulated results and measured frequency responses are illustrated in Fig. 7.
o
co "0
-20
Q)
-L23-
"0
� c:
-40
Ol co
::2
-L23=1.9mm ---- L23=2.1mm
-60
········ _._._--
-60 9.0
9.5
10.0
.
.
.
L23=2.Smm
10 5 .
co
11.0
'-.....
-10
s"
"0
Q) "0 2 'c
Frequency(GHz) Fig. 4.
.. ..' ---- -- - --- -- -- ----------....-:--:----------- __ ;#-----
o
L23 = 2 3mm
Ol co
Effect of coupling window's length on transmission zero
::2
in filter B.
-20
s"
-30 - Measurement ------ Simulation
-40
After theory analysis, simulating and optlmlzmg, [mal dimensions and photographs of Filters A, B and C are shown in Fig. 5 and Fig. 6 respectively.
6
10
9
11
12
Frequency(GHz)
s"
-10
CO "0
Q) "0
-20
.�
-30
::2
-40
c: Ol co
"
"
..-
"
- Measurement ------ Simulation
-50
6
9
10
11
12
Frequency(d8) o
------.--:--- -- -- ---- - --':' - - - - - -, ,--------------- ------- -
I
-10
CO "0 Q) Fig. 5.
"0
Final dimensions of Filters A, B and C. (unit mm)
.�
§, co
811
-20
-30
::2 -40
"
FilterC
-50 +--�--'--.-_---,,-_--.--_--, 10 11 12 6 9
Frequency(d8) Fig, 7,
Simulated results and measured frequency responses of
filter A, B and C.
Fig. 6.
As can be seen, central frequency locates at 9.9 GHz for Filter A, 10.2 GHz for Filter Band 10.12 GHz for Filter C, while transmission zero locates at 10.37 GHz, 9.21 GHz and 9.77 GHz, 1O.5GHz respectively. The proposed filters have minimum insertion loss of 2.01 dB for Filter A, l.8 dB for Filter B and l.99 dB for Filter C, along with return loss larger than 15.1 dB, 17 dB and 14 dB within pass band respectively. Comparing the simulated and measured results, frequency shift and discrepancy are observed to a certain extent. The frequency shift and discrepancy are due to the fabrication
Photographs of fabricated Filters A, B and C.
III. FABRICATION AND EXPERIMENTAL RESULTS The proposed three types of filters were implemented on a substrate with relative permittivity of 5.9, loss tangent of 0.0015 and thickness of 0.8 mm. Each resonant cavity, up and down surfaces of which covering with gold, consists of four substrate layers within linear arrays of metallization via
223
tolerance and error between the actual and nominal values of the relative dielectric constant. In addition, influence of the transition between SMA connector and microstrip line will also result in discrepancy of insertion losses between simulated and measured results within pass band.
[2] Z. Q. Xu, Y. Shi, C. Y. Xu, and P. Wang, "A novel dual mode substrate integrated waveguide filter with mixed source-load coupling (MSLC)," Prog. Electromagn. Res., vol. 136, pp. 595-606,Jan. 2013. [3] Z. Q. Xu, P. Wang, J. X. Liao, and
Y. Shi, "Substrate
integrated waveguide filter with mixed coupled modified trisections," Electron. Lett., vol. 49, no. 7, pp. Mar. 2013. [4] D. D. Carlo and S. Tringali, "Automatic design of circular
IV. CONCLUSION
SIW resonators by a hybrid approach based on polynomial fitting and SVRMS," J of Electromagn. Waves and Appl., vol.
Three types of novel SIW cross-coupling filters with multilayer hexagonal cavity based on L TCC technology are proposed and fabricated. N-2 transmission zeros in proposed n-order filers are achieved by introducing cross-coupling in multilayer substrate. The proposed filters have advantages of low insertion loss and good suppression at out band due to the flexible construction and relative high Q-factor of hexagonal resonant cavity, which has potential for high density microwave communication applications.
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mUltiple
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ACKNOWLEDGEMENT
[8] X. P. Chen and K. Wu, "Self-packaged millimeter-wave
This work was supported by the National Natural Science Foundation of China (Grant Nos. 61201001 and 51172034).
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waveguide
filter
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