www.ietdl.org Published in IET Microwaves, Antennas & Propagation Received on 23rd August 2010 Revised on 3rd December 2010 doi: 10.1049/iet-map.2010.0412
Special Issue on RF/Microwave Communication Subsystems for Emerging Wireless Technologies ISSN 1751-8725
Compact coplanar waveguide filter for integration in ultra wideband medical sensors E. Pancera1 A. Ajami2 T. Zwick1 W. Wiesbeck1 1
Institut fu¨r Hochfrequenztechnik und Elektronik (IHE), Karlsruhe Institute of Technology (KIT), Karlsruhe, Germany Institut fu¨r Hochfrequenztechnik, RWTH Aachen, Aachen, Germany E-mail:
[email protected] 2
Abstract: In this study a novel compact filter structure for ultra-wideband (UWB) applications is presented. The filter is realised with coplanar waveguide technology and it is optimised for the European UWB regulation. It presents a transfer function with sharp transitional bands and low variation of the group delay time in the passband. The filter also has good time-domain behaviour and it introduces only low distortion on the transmitted pulse. The compactness of the presented filter structure and its good time-domain and frequency-domain properties allow its integration in small-size UWB sensors for medical applications, where low size and low distortion are required.
1
Introduction
Since the regulation of the spectrum occupation of the ultrawideband (UWB) technology approved by the FCC in the USA in 2002 [1] and by the European Commission in Europe in 2007 [2], many UWB hardware components, in particular antennas and filters, have been presented in the literature [3]. The availability of a very huge bandwidth for UWB applications (at least 500 MHz instantaneously and up to 7.5 GHz) has shown the UWB technology as a powerful solution in many fields. This is because the huge bandwidth allows obtaining high data rates for communication applications, high resolution for Radar purposes and permits having high user density in sensor networks. Besides, new application areas for the UWB technology are being investigated. Along with the classical communication scenario, interest has grown also for Radar applications, both for localisation and sensing purposes. In particular, in recent times there has been a deep investigation regarding the applicability of the UWB Radar technology in the medical field, both for monitoring some human activities through sensors (respirations, heart beating, water accumulation etc.) [4 – 7] and for medical diagnostic purposes (detection of breast tumour etc.) [8, 9]. It has to be pointed out that the UWB technology in the medical field for diagnostic applications is not to be considered as a substitute of classical well-performing techniques such as computerised tomography etc. Rather, since the UWB is a low-power, low-cost technology, it would permit to realise low-cost devices, which have small size, and hence could be easily adopted and transported in ambulances or directly used by general practitioners. These UWB devices, thanks to the high resolution they could achieve and their transportability, would permit to directly assess the presence of a particular IET Microw. Antennas Propag., 2011, Vol. 5, Iss. 8, pp. 877– 881 doi: 10.1049/iet-map.2010.0412
disease in emergency situations (e.g. traumatic situations, monitoring of human activities etc.), where big-size, highcost classical devices (e.g. computerised tomography) are not available, and would give readily an answer about the pathology to allow the general practitioner to directly treat the patient. For these newly explored medical applications of the UWB technology, intended wearable sensors have to be developed. Critical elements in UWB sensors for medical applications are RF components (antennas, filters etc.). In [7], an investigation on the influence of UWB antennas behaviour for medical applications purposes has been performed. Along with the investigation of the antennas’ influence, also an analysis of the filter impact in UWB medical applications is necessary. Filters are critical elements since they have not only to select the desired frequency spectrum, but also to show good time-domain behaviour, to minimise their distortion on the signal. This is a crucial problem in medical applications, where the recovered signal contains the body information and a distortion of the recovered signal provokes a wrong detection. In this contribution firstly the requirements for UWB filters to be integrated in UWB sensors for medical applications are shown. Then, a new compact coplanar waveguide (CPW) UWB filter is presented, developed to fulfil the derived requirements, which shows very low dimensions, good time-domain and frequency-domain behaviour and introduces very low distortion on the signal. The paper is organised as follows: in the next section, the UWB medical scenario is presented and the design requirements of UWB filters for medical applications are shown. Then, the design of the proposed UWB filter for medical applications is presented in Section 3. Measurement results on a fabricated filter prototype are shown and 877
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2
UWB filters for medical applications
In order to define the requirements of UWB filters for medical applications, firstly the UWB medical scenario is presented. In the UWB medical scenario, the UWB signal impinging on the human body is partially reflected by the air– skin interface and partially penetrates in the human body and is back reflected by the interfaces between tissues with different reflection characteristics. Besides, travelling through the layers, the signal is attenuated according to the layers’ properties [10, 11]. Hence, UWB sensors for medical applications have to be able to detect the signal back-reflected and attenuated by the different layers and to distinguish between its different back-reflected components. Consequently, for UWB medical applications, filters are required, which do not distort the recovered signal. According to this medical scenario, the filter has to show constancy in its passband mask and flatness in its group delay time (GDT). This is because a variation of the filter mask or a non-flatness of the filter GDT results in a worsening of the filter impulse response (high ringing, low peak value etc.), which causes a distortion of the recovered signal [12]. This can be easily quantified recalling the relationship, derived by Parseval’s Theorem [13], between the statistical power M|h| of the absolute value of the filter impulse response |h(t)|, which quantifies the dispersion of h(t), and the filter transfer function amplitude |H(v)| and GDT tg(v), namely 1 +1 d|H(v)| 2 = t |h(t)| dt = dv 2p −1 dv −1 1 +1 + |H(v)|2 (tg (v))2 dv 2p −1 +1
M|h|
2
2
(1)
where v ¼ 2pf. Hence, the presence of GDT variations and suboptimal filter masks have to be avoided not to further increase the distortion and the attenuation on the signal, which is already attenuated by travelling through the body. In the literature different methods for realising UWB filters are reported. One classical technique consists in realising filters with resonators at the UWB centre frequency [14 – 16]. Even if this method is consolidated with a large number of references [17, 18], the time-domain behaviour of this kind of filters is suboptimal (increase of ringing in the filter impulse response) because of energy storage [17], and hence they are not ideal for medical applications, because the ringing can overlap and shadow the back-reflected signal
components to be detected. Another method consists in realising filters through microstrip-to-CPW or microstrip-toslotline transitions, with back-to-back structures [19 – 21]. These filters permit reducing the occupied space, but suffer from a perfect matching of the transitions themselves. In this contribution, an UWB CPW bandpass filter is presented, which is realised through the cascade of a lowpass and a highpass section. The filter is implemented with the CPW technology and is suitable for UWB operations according to the European regulation [2]. The choice of the CPW technology is owing to its several advantages with respect to classical microstrip technology, which makes it suitable for microwave-integrated circuit applications, allowing for easily mounting series as well as shunt active and passive devices, not requiring via holes and showing low radiation losses. The usage of air bridges is often necessary to eliminate unwanted modes. Besides, in recent times new classes of CPW resonators have been introduced, allowing designing of CPW filters with enhanced behaviour and reduced dimensions [22 –25].
3
Filter design
The aim is to realise a filter to be suitable with the EU UWB regulation [2]. In compliance with this regulation, the target filter passband lies in the 6– 8.5 GHz frequency interval. Moreover, the filter stop-band rejection has to be higher than 228.7 dB in the lower stop-band (higher than 243.7 dB up to 3.4 GHz) and higher than 223.7 dB in the upper stop-band [2]. The selected technique for realising the proposed filter consists in the cascade of a lowpass and a highpass section [26, 27]. According to this design method, these two sections can be realised and optimised separately and hence this can improve the filter performance. In the following, these two sections are separately regarded. 3.1
Highpass section design
The highpass section is composed by two basic CPW elements, the open-end series stub and the short-circuited shunt stub [28], which are shown in Figs. 1a and 1b. The choice of the usage of these two different elements for realising the highpass section has been done to increase the sharpness of the transitional band at the cutoff frequency without highly increasing the occupied space, since they can be combined in a compact structure and consequently save space by simultaneously obtaining the desired sharpness. The first element (Fig. 1a) realises an open-end CPW series stub, whose series capacitance is a function of the finger dimensions [28, 29]. The second element (Fig. 1b) is composed by a pair of straight short-circuited
Fig. 1 Highpass filter section a Open-end series stub b Short-circuited shunt stubs c Their integration 878 & The Institution of Engineering and Technology 2011
IET Microw. Antennas Propag., 2011, Vol. 5, Iss. 8, pp. 877 –881 doi: 10.1049/iet-map.2010.0412
www.ietdl.org CPW shunt stubs [28, 30]. Air bridges have been added to connect the ground planes for suppression of the coupled slotline mode [30]. The stubs are realised in the CPW ground plane to permit integration of the previous structure with this one. The structure can be modelled as in [28, 30] with a p-network of three reactances, two of them taking into account the CPW mode excited in the CPW and one owing to the coupled stripline mode of the CPW. Moreover, the presence of the air bridges causes parasitic capacitances and inductance in parallel to the reactances. In order to decrease the required space, these two elements have been put together to form a compact section, as shown in Fig. 1c. The length of the shunt stubs and of the open-end series stubs has been set to quarter guided wavelength corresponding to fL ¼ 6 GHz. Moreover, to further increase the sharpness of the cutoff frequency, a cascade of two sointegrated elements has been realised. 3.2
Fig. 3 Structure of the realised filter
Lowpass section design
The lowpass section has been implemented through a step change in the width of the central conductor. This CPW discontinuity can be modelled as a shunt capacitance Cs [28] (see Fig. 2a), whose value is determined by the step dimensions [31]. In order to highly suppress the upper band and make the transitional band sharper, and not to further increase the filter dimensions, additional slots have been added in the CPW ground plane, at both sides of the CPW, realising a defected ground plane structure (Fig. 2b) [28]. Each slot can be modelled, in first approximation, as series inductance, whose value depends on the slot dimensions. This defected ground plane structure behaves as a bandstop filter, with the centre frequency of the stop-band directly proportional to the slot length. These slots have been bended to save space. The length of these slots has been set to the guided quarter wavelength lgH/4 corresponding to the upper cutoff frequency fH ¼ 8.5 GHz. In order to increase the band rejection capability and the transitional band sharpness, the complete lowpass section is composed by three stepchanges, each one with length lgH/4, and three symmetric slots in the CPW ground plane. The proposed filter structure (see Fig. 3) has been simulated and optimised through computer simulations (CST Microwave Studio). In Fig. 4 the simulated filter behaviour is shown. From the simulation results the filter presents good frequency-domain behaviour with sharp transitional bands and low ringing in the filter passband (the maximum simulated value of the |S21| in the passband is 20.5 dB, the maximum simulated value of the |S11| parameter in the passband is 211.4 dB).
Fig. 4 Absolute value of the simulated S11 (dotted line) and S21 (solid line) parameters of the filter
4
Measurement results and discussion
A prototype of the optimised filter structure has been built on a Rogers substrate RO4003 with 1r ¼ 3.38, thickness d ¼ 1.57 mm (Fig. 5). The total size of the fabricated filter is (inner dimensions) 3 × 1 cm2.
Fig. 5 Fabricated filter
Fig. 2 Lowpass section a Step change in width b Defected ground plane c Their integration IET Microw. Antennas Propag., 2011, Vol. 5, Iss. 8, pp. 877– 881 doi: 10.1049/iet-map.2010.0412
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www.ietdl.org 4.1
Frequency-domain analysis
The behaviour of the fabricated filter prototype has been analysed in the frequency domain, taking into consideration the filter transfer function H( f ) [17, 18]. Let Uin( f ) and Uout ( f ) be the voltage at the filter input and output, respectively, in the frequency domain. The filter transfer function is then defined as H(f ) =
Uout (f ) = |H(f )| · ej/H(f ) Uin (f )
(2)
where |H( f )| is the amplitude and /H( f ) the phase of H( f ). According to its definition, it can be shown that the filter transfer function coincides with the S21 parameter of the filter scattering matrix. From the filter transfer function the GDT tg( f ) is recovered, namely
tg (f ) = −
1 ∂/H(f ) 2p ∂f
(3)
which gives an estimation of the distortion of the signal, introduced by phase differences for various frequencies. The obtained measurement results on the fabricated filter prototype are plotted in Fig. 6. The filter presents good frequency behaviour. It has a 3 dB bandwidth B3dB ¼ 2.4 GHz (6.24 GHz , f , 8.64 GHz). As expected, both the transitional bands are quite sharp, owing to the integration of the different highpass elements and to the used step structure integrated with the defected ground plane. Moreover, the filter presents also good rejection levels in the stop-bands: the |S21| parameter is lower than 237 dB in the lower stop-band up to 5 GHz and lower than 222 dB in the upper stop-band. Analysing the filter passband, it can be seen that the maximum value of the |S21| in the passband is 21.4 dB, whereas the maximum value of the |S11| parameter in the passband is 28.52 dB. The lowering of |S21| in the passband is mainly owing to tolerances involved in the implementation of air bridges. In Fig. 7, the measured GDT tg( f ) is shown. In the considered EU UWB frequency interval it has low oscillations (maximum passband variation Dtg ¼ 460 ps), indicating good linearity of the developed bandpass filter in the EU UWB frequency band. This denotes that the realised filter structure introduces very low distortion on the signal.
Fig. 7 Measured GDT tg(f) of the fabricated filter prototype
4.2
Time-domain analysis
Together with the frequency-domain analysis, also an investigation of the filter time-domain behaviour has been performed. From a time-domain analysis it is possible to directly quantify the filter pulse preserving capability, which is a critical point in UWB medical applications. The filter time-domain analysis can be performed starting from the filter impulse response h(t) [12], which represents the filter output for a very short pulse excitation, in the ideal case a Dirac-function. It can be obtained starting from the filter analytic transfer function, defined as H + (f ) =
2H(f ) 0
f .0 f ≤0
(4)
and then taking the real part
