Review on Field Oriented Control of Induction Motor - ijrest

INTERNATIONAL JOURNAL FOR RESEARCH IN EMERGING SCIENCE AND TECHNOLOGY, VOLUME-2, ISSUE-7, JULY-2015

E-ISSN: 2349-7610

Review on Field Oriented Control of Induction Motor Ayman Y. Yousef 1 and S.M. Abdelmaksoud2 1

Electrical Engineering Department, Shoubra Faculty of Engineering, Benha University, Cairo, Egypt. [email protected] 2 Electrical Engineering Department, Shoubra Faculty of Engineering, Benha University, Cairo, Egypt. [email protected]

ABSTRACT The field oriented control (FOC) of induction motor provide one of the most suitable and popular speed control technique presently used. The principle of field oriented control is based on the control of both the magnitude and the angle of each phase current and voltage. The control of torque is normally achieved by controlling the armature with constant field current. The Field weakening is employed to increase the speed beyond a base peed. The simplicity and flexibility of control of dc motors have made them suitable for variable speed drive applications. In the field oriented control, a d-q coordinates reference frame locked to the rotor flux vector is used to achieve decoupling between the motor flux and torque. They can be thus separately controlled by stator direct-axis current and quadrature-axis current respectively, like in a dc motor. In this paper, the principle, classification, and the most common of the field oriented control are presented. Keywords — Flux Vector, Reference Frame, FOC, Decoupling Control.

1. INTRODUCTION

change the stator frequency [3]. Since the speed is close to

The induction motor is a rugged, reliable, and less expensive

synchronous speed the operating slip is small, and slip power

ac machine. It has been the economical workhorse for use in

loss in the rotor circuit is small. However, this will require a

ac motor drive application. It has been used for both low

frequency converter, which is expensive.

performance as well as high performance drive application. It can be operate in the most adverse circumstances and giving

In drive systems, it is desired that the machine flux is

excellent service with little maintenance.

regulated to provide better utilization of the machine. A requirement for maximum possible transient dynamics is to

The closed loop control of the induction motor is normally required to satisfy the steady state and transient performance specification of ac drives. The control strategy can be implemented as follows [1]:

operate the motor at its rated flux level. Indirect flux regulation schemes such as the volt/hertz control and the slip current control use variable frequency control and have been extensively used in industry [4]. Both the volt/Hertz and

1- Scalar control: where the control variables are dc quantities and only their magnitudes are controlled. 2- Vector control: where both magnitude and phase of the control variables are controlled.

current slip frequency control provide satisfactory steady state performance. The volt/hertz control scheme is quite simple to implement. On the other hand, the current slip frequency control scheme require closed loop current regulation as well

A simple and economic method of induction motor control is to vary the stator voltage at supply frequency. This method of control is characterized by poor dynamic and static performance. Although it is inefficient because of high slip power loss [2], it is used in fans, pump, and blower drives. An

as accurate speed measurement and, therefore, is somewhat complicated to implement. However, both these methods fail to provide satisfactory transient performance. This control strategy is called "Scalar control" which employs general non vector controlled drive schemes. These include simple voltage-

efficient method of speed control for induction motors is to VOLUME-2, ISSUE-7, JULY-2015

COPYRIGHT © 2015 IJREST, ALL RIGHT RESERVED

5


E-ISSN: 2349-7610

fed and current-fed inverters. Scalar control relates to the

Resolution

1 : 1000

1 : 1000

magnitude confrol of a variable only [5]. High performance

Non-Linearity

± 12%

± 12%

drives, such as robotics, rolling mills, and machine tools

Repeatability

± 12%

± 12%

require fast and precise torque response. To achieve this, the

(Tref → Tact) 150 ms

10 to 20 ms

Resolution

1 : 20000

1: 2000

Speed Range

1 : 40

1 : 1000

± 0.01%

± 0.01%

3% sec

0.3% sec

dynamic structure of the machine has to be taken into account. The induction machine is a nonlinear multivariable highly coupled device. Several methods have been proposed to obtain fast torque response with flux regulation [6, 7]. However, the emerging consensus is to use field-oriented control (FOC).

Torque Step Rise Time Speed Control:

(Mmin / Nmax) Static Accuracy (Mact / Nref)

2. FIELD ORIENTED CONTROL The concept of field oriented control (FOC) which was introduced by Siemens company several years ago is recently receiving wide attention. This is also known as transvector control [2], because the control implementation is based on vector transformation from rotating to stationary reference frame and vice versa. The field oriented control (FOC) method permits optimum transient response of the drive and it has a good dynamic response. On the other hand, field orientation is a technique that provides a method of decomposing the stator current into magnetizing component, IM which producing the flux and

Dynamic Accuracy

Table-1: Deference between Scalar and Vector Control The performance figures listed in the table depend on the proportional gain and integration action time constant (PI time constant) set by means of parameters according to the application or process requirements. This table show that, the vector control strategy has much better performance than the scalar control method. Also several literatures such as [5, 9] show that, the vector control methods for induction motor drives allow a better dynamic performance than the scalar control method.

torque component, IT which producing the torque. These components are then decoupled and controlled individually. In

3. PRINCIPLE OF FIELD ORIENTATION

general, the magnetizing component, IM varies slowly and is

The stator phase currents are controlled in a fictitious

kept constant for fast response [8]. On the other words, the

synchronously rotating reference frame (aligned with the flux

other component, IT varies rapidly. It may be concluded that,

vector) and are transformed back to the stator frame to feed the

IM varies with the magnetizing inductance and IT with transient

machine as shown in Fig.1.

inductance of the machine. Therefore, it provides independent control of torque and flux, which is similar to a separately exited dc motor. The magnitude and phase (vector control) of the stator currents are controlled in such a way that the flux and the torque components of current remain decoupled during dynamic and static condition [4]. In order to indicate the deference between the scalar and vector control, a typical performance for torque and speed control of the system described in reference [9] shown in table-1.

Fig-1: Stator current components in d-q Frame. Field orientation can be achieved by aligning the rotor flux

TABLE-I Control Method Required by the Application or Process Torque Control: VOLUME-2, ISSUE-7, JULY-2015

linkage vector along the d axis of the reference frame. With

Motor Scalar

Motor Vector

this arrangement, the control dynamics of the highly coupled

Control

Control

nonlinear structure of the induction motor becomes linearized and decoupled. Finally, the induction motor is controlled like


6


E-ISSN: 2349-7610

an armature controlled dc motor, with Isq analogous to the

controllers [29]. This scheme is simpler to implement than the

armature current and Isd analogous to the field excitation.

direct method of (FOC). Hence, there is an increasing popularity towards the indirect method of (FOC).

4. CLASSIFICATION OF FIELD ORIENTATION The field oriented control (FOC) system schemes can be classified into two groups: 1- Direct method of field orientation. 2- Indirect method of field orientation. Fig-3: Indirect flux sensing of induction motor vector control

4.1. Direct Field Orientation

system

The direct method of field oriented control, which was originally proposed by BLASCHKE [10] is shown in Fig. 2. This technique utilizes direct sensing of the air gap flux vector by using one of the measurement techniques which explained briefly in section (5).

5. DETERMINATION OF FLUX VECTOR POSITION It is decisive for field oriented control (FOC) to have up-todate information [12] on the instantaneous magnitude  and angular position of the flux wave

 . This is the first problem

of the field orientation. The solution is discussed in the following subsections which described some of flux sensing schemes.

5.1. Air Gap Hall Sensors Hall-effect sensors are mounted in the air gap at the stator surface. There are also known as direct field sensitive devices Fig-2: Direct flux sensing of induction motor vector control system The measured air gap flux signal is feedback to the control and used to decouple the torque producing component of stator current from the producing component. Since this method uses feedback control and direct sensing of the regulated variable, it is essentially insensitive to variations.

[13]. These sensors are produce signals representing the local flux density at the measuring points. One of the (FOC) systems depending on the air gap hall sensors method is described in reference [10]. These signals are highly distorted by effect of the rotor slots and the thermal and mechanical stresses on the sensing devices [14]. Generally, the accuracy of a torque signal computed from this information is likely to be poor [15].

4.2. Indirect Field Orientation The indirect method of field oriented control, was originally proposed by HASSE [11] is shown in fig.( 1.3). This method avoids the requirement of flux acquisition (sensing devices) by using known motor parameters to compute the appropriate motor slip frequency to obtain the desired flux position [1]. On the other hands, the rotor flux is estimated from the stator current vector, voltage vector and or rotor speed, and then this estimate is, in effect, fed forward to the flux and torque VOLUME-2, ISSUE-7, JULY-2015

5.2. Sensing Coils Sensing coils having width of full pole pitch have been installed in the stator, in order to avoid the effect of rotor slots by filter out the undesirable slot harmonics [13]. This produces emf signals proportional to flux change which, after integration, taken as representing the main flux of the motor. By adding suitable components of the stator currents, signals for the rotor flux are obtained [15]. The disadvantage that,


7


E-ISSN: 2349-7610

when the drift of the integrators become too large; this would

The equivalent circuit of an induction motor is shown in Fig.

prevent the application to drives with position control such as

4, where the stator, mutual and rotor fluxes are identified.

machine tool feed drives [13]. The complete implementation of the control system depending on this method is described in work [16].

5.3. Stator Winding as Sensing Coils The last mentioned drawback is avoid by measuring the terminal voltages i.e. using the stator windings as a sensing coils. The difficulties arise from the resistance voltage drop which dominates at low frequency [12]. This disadvantage is offset by compensating the IR-drop prior to integration [15]. Since, the stator resistance is temperature dependent, the scheme in this method becomes quite difficult. Some of the field orientation control systems depending on this method are

Fig-4: Equivalent circuit of induction motor. The equations for a balanced symmetrical three-phase induction motor can be written as:

U s  Rs i s  d  s / dt

(1)

U r  Rr i r  d  r / dt

(2)

Te  (3 / 2) Lm [is (ir e j )* ]

(3)

discussed in references [17]. Where: dθ/dt = ω, which is the rotor speed.

5.4. Machine Model: This method is remove the limitations in the low speed range, it was made possible because in the meantime microprocessors had advanced sufficiently to permit a computational solution. There are two types of machine (flux) model; voltage and current machine models. The principle of this method depends upon computing the magnetizing current Imr and the flux position φ, from the measured signals (voltage & current or current & speed) depending on the type of the model.

6. REFERENCE FRAMES FOR FIELD

6.1. Orientation Based on Stator Flux Reference Frame The vector diagram of stator current and stator flux is shown in Fig .5, and the stator flux is given by;

 s  Ls i s  Lm i r  Lm i ms

(4)

Where

ims  (1   s )i s  i r

(5)

 s  ( Ls / Lm )  1

(6)

ORIENTATION The control of induction motor using the principle of fieldorientation gives control characteristics similar to that of a separately exited dc motor. The torque component and the flux component of current are identified and control is exercised on them. The control variables can be decoupled by processing them appropriately, or a natural decoupling can be achieved by

Fig-5: Stator Flux and Current components

properly orienting the current vector [18]. The magnitude and

with Stator Reference Frame.

phase of the stator current determines the flux produced and the torque generated in the machine. These two parameters can

If the stator current is projected in phase and quadrature to the

be controlled by properly orienting the stator current and it can

stator flux, then:

be done with respect to one of the three specific synchronous reference frames [19]. These specific reference frames are the

is  isd  isq

(7)

stator flux, the air gap (mutual) flux, and the rotor flux vectors.

VOLUME-2, ISSUE-7, JULY-2015


8


E-ISSN: 2349-7610

Substituting from equations (4) to Eq. (7) in equations (1) to Eq. (3) and separating out the real and imaginary parts;

Rs isd  Lm (dims / dt)  U sd

(8)

Rs isq  Lm1ims  U sq

(9)

isd  Tr (disd / dt)  ( s / Ls )

(10)

 s1Tr isq  (Tr / Ls )[ds / dt]

s1  [isq  Tr (disq / dt) /[(Tr / Ls )s

Fig-6 : Induction motor model with stator field orientation. (11)

 Tr isd ]

6.2. Orientation Based on Air Gap Flux Reference

Te  (3 / 2) Lm [ims isq )  (3 / 2) s .isd ]

(12)

UΦsd and UΦsq are the voltage components, ωs1 is the slip angular frequency, σ is the total leakage factor, T r is the total rotor time constant, where:

The mutual flux is defined by;

 m  Lm (i s  i r )  Lm i m

 s1  1  

(13)

  1  [1 /(1   s )(1   r )]

(14)

Tr  Lr / Rr

Frame

Where

im  i s  i r

(15)

stator voltages and currents, Eq. (10) defines the build-up of flux and Eq. (12) defines the electromagnetic torque produced.

the stator voltage and currents, flux build-up, and the torque produced are given by:

imd  Ts1 (dimd / dt)  (U nd / Rs )

It is seen from Eq. (10), that the flux built-up not only depends

 s1Ts1isq  (1 / Rs )[dm / dt]

on iΦsd, but also on iΦsq and thus a coupling exists between the

imq  Ts1 (dimq / dt)  (U nq / Rs )

currents in the two axes. The equations (10), Eq. (11) and Eq.

 s1Ts1isd  (1 / Rs )m

(12) representing the structure of the induction motor in case of stator field orientation. These equations can be simplified to;

 Tr isd ]

Te  (3 / 2) s .isq

imd  Tr1 (dimd / dt)  (Tr / Lm )(dm / dt)  ( m / Lm )  s1Tr1isq

s1  [imq  Tr1 (disq / dt)] /[(Tr / Lm )s

(1  Tr D)isd  Trs1isq

s1  (1  Tr D)isq /[(Tr / Ls )s

(20)

It can be shown in a similar way that the relationships between

Equation (8) and Eq. (9) define the relation ship between the

 (1 / Ls )(1  Tr D)s

(19)

(16)

(22)

(23)

(24)

 Tr1imd ]

Te  (3 / 2) Lm (im imq )  (3 / 2) m .imq (17)

(21)

(25)

Where

Tr1  ( Lr  Lm ) / Rr

(26)

(18) From Eq. (23), it can be inferred that the flux dynamics

Equation (16), Eq. (17) and Eq. (18) are used to construct the

depends on both the current components of stator current iΦmd

block diagram shown in Fig. 6 which describe the complexity

and iΦmq the coupling is exist. Also equation (23), Eq. (24) and

of the control with stator field orientation due to the coupling

Eq. (25) can be simplified to:

between iΦsd and iΦsq



9


(1  Tr1 D)imd  Tr1s1imq

(27)

 (1 / Lm )(1  Tr D)m

for the flux and the torque are completely decoupled. Equ. 30 gives:

imr  r / Lm

s1  (1  Tr1 D)imd /[(Tr / Lm )m

E-ISSN: 2349-7610

(38)

(28)

 Tr1imd ]

Substituting from Eq. (37) in Eq. (34) and from Equ. 31 in Eq.

Te  (3 / 2) m .imq

(29)

(36)

From these equations the block diagram describe the control with air gap field orientation can be constructed as shown in

ird  (1  Tr D)(m / Lm )

(39)

Te  (3 / 2)(Lm / Lr )(r .irq

(40)

and

Fig. 7.

Then, from Eq. (39) and Eq. (40), the model of the induction motor with rotor field orientation can be drawn in the simple block diagram shown in Fig. 8.

Fig-7: Induction Motor Model with Air Gap Field Orientation.

6.2. Orientation Based on Rotor Flux Reference Frame

Fig-8: Induction Motor Model with Rotor Field Orientation.

The rotor flux is defined by;

 r  Lr i r  Lm i s  Lm i mr

Finally, the rotor flux machine model shown in Fig. 8 contains (30)

flux through a first order time delay which is the rotor circuit

Where

i mr  (1   r )i r  i s

two major paths. The control input iΦrd is related to the rotor

(31)

time constant Tr. If the flux is kept constant, only the quadrature component of stator current i Φrq is used to control

And

the torque. This relationship is analogous to a separately

 r  ( Lr / Lm )  1

(32)

current if the flux is held constant. Unlike the rotor flux case,

ird  Ts (dird / dt)  (U md / Rs )  1Ts imq  (1   )Ts [dimr / dt] irq  Ts (dirq / dt)  (U rq / Rs )

(33)

Fig. 6 and Fig. 7, show that the flux path is also affected by the quadrature component of stator current. This intercoupling is due to the inherent rotor leakage inductance Lr in the case of

(34)

 1Ts ird  (1   )Ts1imr

the air gap flux model and both the stator and rotor leakage (Ls & Lr) in the case of stator flux model [19]. Consequently, only

Tr1 (dimr / dt)  i rd  imr

(35)

1  irq / Tr imr

(36)

Te  (3 / 2)[ Lm /(1   r )]im r irq

(37)

for the rotor flux orientation and for a constant rotor flux there is no coupling between the flux and torque in the sense that the rotor flux is produced by a constant direct component of stator current (field current) whereas the torque is produced by the

 (3 / 2)[ r /(1   r )].irq

quadrature component of stator current (torque component) which is varied as desired [15]. A detail comparison between

From Eq. (35) and Eq. (37), it is evident that the flux dynamics are very simple and the stator current components responsible


excited dc machine in which torque is proportional to armature

the three method of field orientation is discussed in reference [5].


10


E-ISSN: 2349-7610

7. INDUCTION MOTOR DECOUPLING CONTROL The complex nonlinear coupling structure of induction motor can be decoupled by controlling the machine with a suitable stator mmf vector. This can be done by feeding the machine with unique sets of currents and frequencies generated by decoupling control schemes [19]. The decoupling controllers are obtained by taking the inverse of the machine flux models

Fig-11: Decoupling controllers with current fed induction

shown in Fig. 6, Fig. 7 and Fig. 8. The result of this inversion

motor for stator, air gap, and rotor flux orientation

is called a decoupling network. This decoupling circuit is

These decoupling control schemes utilize an estimated slip

necessary to obtain an independent control of the flux and

angle plus the feedback rotor angle for the prediction of the

electromagnetic torque of the induction motor. Fig. 9 and Fig.

flux vector position (for the alignment of the flux vector along

10 shows the decoupling networks for rotor, stator, and air gap

the d-axis). In the stator and air gap flux decoupling control

flux orientation respectively.

schemes, a step change in command torque will case the generation of delta function due to disq/dt in Fig. 10. In practice, the rate of rise of machine currents are limited by the stator leakage inductance, therefore a very small first-order time delay is incorporated in the torque input path to ensure a smooth function of isq [19]. Then, the stator and air gap flux decoupling control schemes

Fig-9: Decoupling circuit for rotor flux orientation

shown in Fig. 10 require differentiation of isq in calculating command slip frequency. A large command slip frequency is

The decoupling circuit obtained for the indirect field oriented controller can also be used in the case of direct field orientation [20].

required during torque transient for the correct adjustment of the current vector angle φ. In order to avoid differentiation and simplify the complexity of the decoupling control scheme of Fig.11, the command slip frequency is employed instead of command torque. The modified decoupling control scheme shown in Fig. 12 utilizing command slip frequency as input. With this scheme, differentiation is eliminated and the two first order blocks can share an identical subroutine. Hence, the overall complexity is reduced.

Fig-10: Decoupling circuit for stator and air gap flux orientation The decoupling controllers based on stator, air gap, and rotor flux methods with current impressed scheme are shown in Fig. ll.

Fig-12: Modified decoupling controller scheme for stator and air gap flux orientation



11


E-ISSN: 2349-7610

In case of voltage impressed schemes (voltage-fed induction

The machine parameters (Rs, Rr , Lm, and Lr) will vary due to

motor), the voltage vectors Usd and Usq can be generated from

temperature, frequency, and saturation effects. The indirect

the current and flux vector by the following relationships for

vector controller dependent on the rotor resistance Rr, the

stator, air gap and rotor flux orientation respectively:

mutual inductance Lm, and self inductance of the rotor, Lr. In particular, the rotor resistance value changes 100% with

U sdq  Rs I sdq  ( D  js )s

(41)

temperature and frequency. This causes the actual flux and torque deviate from their set values. Also, the inductances

U sdq  Rs I sdq  ( D  js )(Ls1 I sdq  m )

(42)

U sdq  Rs I sdq  ( D  js )(Ls I sdq

(43)

change with the saturation of magnetic material.

8.1. Parameters Sensitivitv with CSI

 Lm r / Lr ) The voltage decoupling circuit (assume stator field orientation) depending on Usd* and Usq* of Eq. (41) is shown in Fig. 13. It incorporate also the current decoupling network to produce i sd* and isq* furthermore, the frequency ω1 of the flux linkage Ψ must be computed. The structure of ω1-calculator depends on the choice of control [8]. If indirect control is used, then ω 1 is calculated from ωs1* produced by the current decoupling circuit added to the rotor speed ω as shown in Fig. 13.

In the system with machine fed by impressed currents, the drive dynamics are independent of the stator supply frequency. The amount of deviation in the machine flux due to parameter variations increases with larger load values. At no-load, the variation in the ratio of actual to command flux is minimal, However, the machine torque is being adversely affected. In practice, an outer loop speed controller can be used to compensate the discrepancy in torque for particular desired speed.

8.2. Parameters Sensitivitv with VSI In the

system

with Voltage

impressed

scheme,

the

performance is speed dependent. At higher speed, the degree of sensitivity is very much lower. With voltage impression at high speeds, the machine air gap voltage is approximately equal to the terminal voltage. Therefore, variations on the rotor Fig-13: Decoupling controller with voltage-fed induction motor

resistance do not affect the machine flux model. However, at low speeds, the stator impedance voltage drop becomes significant when compared to the air gap voltage. Hence, the effect increases with varying the rotor resistance. Finally, the

8. EFFECT OF MACHINE PARAMETERS

parameter sensitivity characteristics are similar in all the three methods of flux control (stator, air gap, and rotor flux). Also it

VARIATION

similar with the current impressed scheme and the voltage

The performance of decoupling control methods based on

impressed scheme at zero speed. However, at higher speeds,

inverting machine models can be influenced by a mismatch

the latter has an edge over the former scheme [19].

between the parameters values being used in the controller and the actual machine parameters [19]. On the other hand, the parameters of the machine may change during the operation of

9. MACHINE PARAMETERS IDENTIFICATION

the drive, causing deviation between the corresponding signals of the model and the machine [21].

From the discussion of the previous section, the parameter sensitivity of the indirect vector controlled induction motor results in steady state errors in torque and flux in the addition



12


of enhanced losses reducing the output of the drive system and

E-ISSN: 2349-7610

11. MICROCONTROLLER BASED FLUX

transient oscillations in flux and torque. Consequently, the

VECTOR CONTROL

efficiency of the motor drive decreases [22]. The incorporation of parameter adaptation schemes is required to off-set these

The microprocessor implementation of the field oriented

effects. These schemes are classified depending on the extent

controllers has the advantages that the hardware can be

of the use of the induction motor model. The Various schemes

simplified and as a result system reliability can be improved

which have been proposed for parameter adaptation in such

[2].

drives can be summarized based on one of the following

microprocessors is that the same standard hardware can be

strategies;

used for many different applications because the function of

The

main

advantage

of

digital

control

with

the processor determined by a flexible custom designed 1- Direct monitoring of the alignment of the flux and

software

program

[15].

Several

types

of

the

fast

microprocessors are used in the implementation of field

torque producing stator current component axes. 2- Continuous real time measurement of the instantaneous

oriented schemes such as INTEL (80386 & 80486), ZILOG 8000 or MOTOROLA 68020 to perform the machine model

rotor resistance. 3- Measurement of modified reactive power with field

and the digital controller [2]. Some of the suggested systems

angle or a combination of the rotor flux and torque

based on the microprocessors 8085 and 8086 are implemented

producing component of the stator current.

in references [12,18]. In the past, the highest performance realtime control applications have employed 8-, 16- and 32-bit

Many systems of field oriented control are implemented with

microprocessors together with interrupt handler chips,

the parameter adaptation schemes are indicated in the previous

programmable timer chips, and ROM and RAM chips, to

works such as [11,23]. A complete review on the parameters

achieve what can now be achieved in a single microcontroller

adaptation of the indirect vector controller is discussed in

chip. The microcontroller such as INTEL 8096, MOTOROLA

details in reference [24]

68HC11and SAB80C166 can be considered as microprocessor with different peripherals integrated internally with them in the

10. DIGITAL SIGNAL PROCESSING The signal processing required for the field oriented control (FOC) of ac motors as well as the flux acquisition (section 5)

same single chip, has simplified many industrial and control application. Consequently, microcontrollers are present on a commercial scale and used extensively in various industrial applications.

using a dynamic model is of considerable complexity and representing the second problem of the field orientation [15]. In the earlier implementation schemes of field oriented control systems, the analog methods had been used to perform the necessary

operations

[14].

In

particular,

the

11. SPEED ESTIMATION FOR SENSORLESS FIELD ORIENTED CONTROL SYSTEMS

various

multipliers, division and function generators needed for the

It is well known that, there are two major techniques for high

coordinate transformation are expensive and difficult to adjust

performance control of induction motor. These two techniques

when they are realized with analogue components, in view of

are, the slip frequency controlled type vector control and the

the necessary high accuracy [15]. These field oriented schemes had complicated hardware, not practical and had gained little industrial interest. The solution of this problems could be solved by converting from analog to digital techniques, using software algorithms implemented on microcomputers and signal processors [13].

field orientation control. The field oriented control system uses an electromechanical speed transducer coupled to the motor shaft to measure the rotor speed. The transducer encumbers the mechanical drive, and spoils the general characteristics of ruggedness and mechanical simplicity of the induction motor drive [25]. These speed sensors restrict the applications of vector control system, For example, if there is no space for the



13


E-ISSN: 2349-7610

sensor connected, the vector control system can not be adapted

an entirely open loop basis. In the first, the angle φ of the rotor

for these system. Also, the problems for practical use, such as

flux vector is defined as follows:

a low speed behavior, have been found with speed sensors [26]. For this regard, a speed sensorless system is preferred. The advantages of sensorless induction motor drives are lower cost, reduced size of the machine set, elimination of the sensor

(48) The derivative of the above equation gives the

synchronous

speed ω1:

cable, and increased reliability. The method of vector control without any rotational transducer (sensorless) has been developed, employing the flux calculated from the stator voltages (Usα & Usβ) and the stator currents (Isα & Isβ). Since,

1  d / dt  [ (d / dt)   ( d / dt)] /[   ] 2

2

(49)

this method required two sensors only for measuring the stator currents and voltages. In order to obtain an accurate dynamic

Substituting for (dΨα/dt) and (dΨβ/dt) from Eq. (46) and Eq.

representation of the motor speed, it is necessary to base the

(47) in Eq. (49), then:

calculation on the coupled circuit equations of the induction motor [28]. Since, the motor voltage (Usα & Usβ) and currents

1    ( Lm / Tr )[is   is  ]

(Isα & Isβ) are measured in a stationary frame of reference, it is

/[   ]

also convenient to express these equations in the stationary frame (i.e. there are two different equations to estimate the rotor flux in the α-β frame).

2

2

(50)

Then, the rotor speed is given by;

  1  ( Lm / Tr )[is   is  ]

d / dt  ( Lr / Lm )U s  Rs is

(44)

 Ls (dis / dt)

(51)

/[   ] 2

2

Equation (51) indicate that the instantaneous rotor speed cd

d / dt  ( Lr / Lm )U s  Rs is

(45)

 Ls ( dis / dt)

can be obtained using rotor flux estimated in Eq. (44) and Eq. (45). This process is illustrated in the block diagram shown in

and

Fig. 14.

d / dt  (1 / Tr )    ( Lm / Tr )is (46) d / dt    (1 / Tr )  ( Lm / Tr )is

(47)

Where: Ls, Lr : are the stator and rotor self inductances. Lm : is the mutual inductance. Rs : is the stator resistance. Tr : is the rotor time constant. ω : is the rotor electrical angular velocity. Fig-14: Block diagram for open loop calculation of rotor

σ; is the motor leakage coefficient.

speed.

Ψ, is, Us, : are the rotor flux, stator current, and stator voltage

This open loop calculation process requires knowledge of four

respectively

constants that depends on the motor parameters. The Equation (44) and Eq. (45) does not contain the rotor speed ω.

parameter sensitivity can be reduced by calculating the slip in

However, Eq. (46) and Eq. (47) does. Thus, by using these two

rotating reference frame (d-q frame) that is locked to the rotor

equations, the instantaneous rotor speed ω can be calculated

flux vector. There are two problems for this speed estimation

directly from the measuring values of voltages and currents on

technique related to the rotor flux estimation in Eq. (44) and

  tan (  /  )

1 VOLUME-2, ISSUE-7,JULY-2015


14


Eq. (45). The first problem is the need of an ideal integral

[7]

E-ISSN: 2349-7610

I. Takahashi & T. Noguchi "A New Quick Response and

where the calculated rotor flux does not work so that it is

High Efficiency Control Strategy of an Induction

unstable in initial operation, as motor speed approaches zero.

Motor", Proc. lEEE/IA Ann. Conf. Rec., pp. 496-502,

The second problem is the dependence of motor parameters,

1985.

such as stator resistance thermal variation and saturation of

[8]

inductance parameters. Particularly, stator resistance variation causes the calculated rotor flux to vary and lead to torque

Ion Boldea, & Syed A. Nasar "Vector Control of AC Drive", CRC press, Florida 1992.

[9]

variation in a low speed range.

G. O. Garcia, R. M. Stephan, & E. H. Watanabe "Comparing the Indirect Field-Oriented Control with a Scalar Method", IEEE Trans. Ind. Elec., Vol. 41, No. 2

12. CONCLUSIONS

[10]

F. Blaschke "The Principle of Field Orientation as

This paper reviews the various aspects in the field oriented

Applied to the New Transvector Oosed Loop Control

control

System for Rotating Field Machine", Siemens Rev., Vol.

of

induction

motor

including

the

principles,

37, No. 5, pp. 217-220, June 1972.

classification (direct and indirect FOC), and the flux vector position determination. The mathematical models of the stator

[11]

K. Hasse "Zur Dynamik Drehzahlgeregegelter Antriebe

flux, the air gap flux, and the rotor flux reference frames was

mit

determined. The decoupling control with current and voltage

Kurzschlusslaufermachinen" ("On the dynamics of speed

fed induction motor for stator, air gap, and rotor flux

control of static ac drives with squirrel-cage induction

orientation was discussed. The field oriented control makes the

machines"), Ph.D. dissertation, TH Darmstadt, 1969.

induction motor is controlled like an armature controlled dc

[12]

stomrich-tergespeisten

Asynchron-

R. Gabriel & W. Leonhard "Microprocessor Control of

motor, with Isq analogous to the armature current and Isd

Induction Motor", IEEE Ind. Appl., Society Annual

analogous to the field excitation. It is expected that the review

Meeting, pp. 385-395, 1982.

will help those interested in the field of efficient and high

[13]

W. Leonhard "30 Years Space Vectors, 20 Years Field Orientation, 10 Years Digital Signal Processing with

performance drives.

Controlled AC Drives, a Review", EPE Journal, Vol. 1, No. 2, pp. 89-102, October 1991.

REFERENCES [1]

M. H. Rashid "Power Electronics Circuits, Devices, and

[14]

Doncker "Universal Field Oriented Controller Based on

Applications", Englewood Cliffs, New Jersey, 1988. [2]

Air Gap Flux Sensing via Third Harmonic Stator

B. K. Bose "Introduction to AC Drives, Speed Control of

Voltage", IEEE Trans., Ind. Appl., Vol. 30, No. 2, pp.

AC Drives", IEEE Collection, pp. 1-21, 1978. [3]

448-455, March/April 1994.

S. B. Dewan, G. R. Slemon, & A. Stranghen "Power Semiconductor Drives", Published in Canada, 1984.

[4]

[15]

[16]

Ind. Appl., Vol. IA-13, No. 2, pp. 139-146, March/April

A. K. Ibrahim "Performance Characteristics of Induction Drives

Under

Field

Oriented

1977.

Control

Techniques", M.Sc. Thesis, Faculty of Engineering,

[17]

using Tapped Stator Windings", IEEE Trans. Power

S.Yamamura & T. Nakagawa "Equivalent Circuit and

Elec., Vol. 5, No. 4, pp. 446-453, October 1990,

Field Acceleration Method of AC Servomotor by Means of Induction Motor", Trans. lEE Japan, Vol. 102-B, No. 7, pp. 433- 439, 1985.


D. S. Zinger, T. A. Lipo, & D. W. Novotny "A Direct Field Oriented Controller for Induction Motor Drives

Cairo University, 1994. [6]

A. B. Plunkett "Direct Flux and Torque Regulation in a PWM Inverter- Induction Motor Drive", IEEE Trans.,

No. 6, pp. 562-575, December 1990.

Motor

W. Leonhard "Control of Electrical Drives", SprigerVerlag, New York, 1985.

P. C. Sen "Electric Motor Drives and Control-Past, Present, and Future", IEEE Trans. Ind. Eiec., Vol. 37,

[5]

G. B. Griva, M. Pastroelh, J. C. Moreira, & R. W. DE

[18]

S. Sathiakumar, S. K. Biswas, & J. Vithayathil "Microprocessor Based Field-Oriented Control of a CSI-


15


[19]

E-ISSN: 2349-7610

Fed Induction Motor Drive", IEEE Trans., Ind. Elec.,

Induction Motor Drives", IEEE Trans. Ind. Appl., Vol.

Vol. IB-33, No. 1, pp. 39-43, February 1986.

IA-21, No. 4, pp. 624-632, May/June 1985.

Edward Y. Y. Ho & P. C. Sen "Decoupling Control of Induction Motor Drives", IEEE Trans., Ind. Elec., Vol. 35, No. 2, pp. 253-262, May 1988.

[20]

R. W. DE Doncker & D. W. Novotny "The Universal Field Oriented Controller", IEEE Trans., Ind. Appl., Vol. 30, No. 1, pp. 92-100, Jan/February 1994.

[21]

J. Holtz & T. Thimm "Identification of the Machine Parameters in a Vector-Controlled Induction Motor Drive", IEEE Trans., Ind. Appl., Vol. 27, No. 6, pp. 1111-1118, November /December 1991,

[22]

J. Holtz "Speed Estimation and Sensorless Control of AC Drives", lECON' 93, International Conference on Ind, Elec. Control and Instrumentation, Vol. 2, pp. 649-654, Hawaii, USA, 1993.

[23]

F. M. H. Khater, R. D. Lorenz, D. W. Novotny, & K. Tang "Selection of Flux Level in Field Oriented Induction Machine Controllers wit Consideration of Magnetic

Saturation

Effects",

IEEE

Trans.,

Ind.

Appl.,Vol. IA-23, No. 2, pp. 276-281, March/April 1987. [24]

R. Krishnan & A. S. Bharadwaj "A Review of Parameter Sensitivity and Adaptation in Indirect Vector Controlled Induction Motor Drive Systems", IEEE Trans., Power Elec., Vol. 6, No. 4, pp. 695-703, October 1991.

[25]

L. Ben-Brahim & A. Kawamura "A Fully Digitized Field-Oriented Controlled Induction Motor Drive using Only Current Sensors", IEEE Trans., Ind. Elec., Vol. 39, No. 3, pp. 241-249, June 1992.

[26]

T. Ohtani, N. Takada, & K. Tanaka "Vector Control of Induction Motor Without Shaft Encoder", IEEE Trans., Ind. Appl., Vol. 28, No. 1, pp. 157-164, Jan/February 1992.

[27]

C. Schauder "Adaptive Speed Identification for Vector Control of Induction Motors Without Rotational Transducers", IEEE Trans., Ind. Appl, Vol. 28, No. 5, pp. 1054-1061 Sept/October 1992.

[28]

C. Schauder "Adaptive Speed Identification for Vector Control of Induction Motors Without Rotational Transducers", IEEE Trans., Ind. Appl, Vol. 28, No. 5, pp. 1054-1061 Sept/October 1992.

[29]

Takayoshi Matsuo & Thomas A. Lipo "A Rotor Parameter Identification Scheme for Vector ControUed



16